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1
All-Metal Monopulse Antenna Array in the
Ka-Band with a comparator network combining
Ridge and Groove Gap Waveguides
Miguel Ferrando-Rocher, Member, IEEE, Jose I. Herranz-Herruzo, Member, IEEE
Alejandro Valero-Nogueira, Senior Member, IEEE and Bernardo Bernardo-Clemente.
Abstract—This letter presents an all-metal three-layer
monopulse antenna based on Gap Waveguide technology oper-
ating in Ka-band. (specifically from 29 to 31 GHz) for direction-
finding applications. This contribution stands out mainly for
the simplicity of its single-layer comparator network (layer 1)
composed of a combination of ridge and groove gap waveguides.
Also, the compactness of the feed network (layer 2) allows for a
low-profile monopulse antenna comprising only three aluminum
pieces. The antenna is bottom-fed through three ports connected,
two of them to the comparator network, and one directly to
the corporate-feed network, to achieve one sum (Σ) and two
difference patterns (∆H,∆E). Experimental results validate
the concept, showing a close agreement with simulations. The
measured null depth is close to −30 dB in both difference
patterns, and the sum pattern achieves 26.9 dBi directivity with
a realized radiation efficiency greater than 80%.
Index Terms—gap waveguide, comparator network, ridge gap
waveguide, groove gap waveguide, monopulse, radar
I. INTRODUCTION
Monopulse antennas for tracking and targeting applications
are a constantly changing field that is still eager to innovate.
They present advantages over alternative techniques, like those
based on sequential lobes. Among the variety of applications
in which this antenna can be used, we find aircraft tracking
in air traffic control [1], flying car-to-car communications [2],
indoor communications [3], detection of cosmic debris [4],
and others. Conventionally, three-dimensional monopulse an-
tennas based on horns [5], lenses [6] and Cassegrain parabolic
antennas [7] are typical candidates to reach narrow beam width
and high-gain. Nevertheless, these antenna types are usually
bulky, costly, and have low integration, limiting their practical
application. Different technologies have been used to find
solutions with the same good performance as those mentioned
above, but with improved lightness and size. These approaches
could be framed in two large categories: non-metallic and all-
metal antennas.
Among the non-metallic ones, the following stand out:
transmitarrays, reflectarrays or radial-line slot arrays (RLSA).
Reconfigurable reflectarray [8]–[10] or transmitarray [11], [12]
antennas can be a low-profile compromise between radiation
performance and bulkiness and could be a proper choice for
large-aperture applications. Printed transmission lines, such as
SIW or microstrip, have been used to design very low-profile
wideband antennas for direction-finding applications [13]–
[15]. As to RLSA antennas, [16] presents an additive-printed
solution, which proved to be effective, low-cost, and low-
profile. Finally, it is particularly appropriate to highlight the
emergence of metasurfaces (MTS) as a particular solution for
monopulse antennas [17], [18]. Nonetheless, the efficiency of
the aperture in this case still stands at 40%.
Within the metallic direction-finding antennas, it is worth
mentioning those based on hollow waveguides. Although they
are inherently high-efficiency antennas, due to the full-metal
characteristic, they suffer from the complexity of the feeding
structures [19]. In addition, it must be taken into account
that, at millimeter-wave frequencies, different metal blocks
are difficult to assemble with each other, which can lead
to spoiling the expected performance. Notwithstanding, good
electrical contact can be achieved by using hardly accessible
fabrication techniques, such as diffusion bonding, but with a
increased fabrication cost [20], [21].
In all this context, a few monopulse antenna designs us-
ing Gap Waveguide (GW) technology, which is a known
solution for all-metal structures in the millimeter-wave band,
have been proposed in the last few years. Particularly, three
works describing monopulse antennas, based on different gap-
waveguide-based structures, have been published in the last
few years [22]–[24]. Of these three contributions, the first two
are RLSAs that integrate GW in some way. The third, to the
best of these authors’ knowledge, is the only one making use
of GW in a fuller fashion. That W-band monopulse slot array
antenna features excellent radiation performance, isolation,
and input impedance matching. However, it still requires three
layers to build the comparator network. In fact, and as stated
in [22], the compactness of the feeding network in monopulse
antennas in GW is still an open issue. This work attempts,
at least, to take a move in that direction by simplifying the
comparator network one step further.
This Ka-band monopulse antenna based on GW technology
consists of only three metallic layers (Fig. 1), which simplifies
the design, cost, and assembly of the system. The following
sections are devoted to the details, dimensions, performance,
and experimental measurements. The antenna description is
done sequentially, from the top cover to the bottom layer. Sec-
tion II details the radiating layer (layer 3) and the corporate-
feed network, as well as the cavities of the intermediate
layer (layer 2). Section III focuses on the piece in which the
comparator network is located (layer 1). Finally, Section IV
presents the experimental validation of the concept and Section
V draws the main conclusions.
2
(a) (b) (c) (d)
Fig. 1. Antenna layers: (a) Top lid (b) 8×8 corporate-fed array antenna, (c) comparator network, and (d) exploded view of the full antenna. The dimensions
of the antenna are 12×12 cm side and 2.6 cm total height.
TABLE I
FEATU RED D IM ENS IO NS OF T HE P ROTOT YPE
- wRGW pRGW 1pRGW 2wGGW sGGW wslit lslit pnail wnail lcavity
Dim. (in mm) 0.8 0.75 0.5 1.25 0.55 0.3 6.1 2.5 1.5 5
Fig. 2. Front view of the intermediate layer. A 4×8 array is inscribed in blue
dashed lines. Waveguide types and ports are indicated.
Fig. 3. Detail of the central part of the distribution network, located in the
intermediate layer. The RGW-GGW transitions are visible, and the phase of
the output ports of each GGW-GGW splitter is also indicated.
II. 8×8 MO NO PU LS E ANT EN NA DESIGN
This section explains the first two layers of the antenna.
The top layer basically consists of 8×8 equispaced square
apertures (Fig. 1(a)). The layer below houses the corporate
distribution network that feeds 8×8 square cavities backing
up the radiating apertures of the top lid (Fig. 1(b)). The
spacing between apertures is 10 mm; λ0at 30 GHz. The
square cavities have side dimensions of 0.5λ0. They function
(a)
(b)
(c)
Fig. 4. Phase variation along the network depending on the feeding port. Port
1 provides the Σ-pattern, port 2 the ∆Eand port 3 the ∆H.
similarly to cylindrical cavities [25], but are more compact.
Since more space is available to house the distribution network
between cavities, the risk of mutual coupling is reduced.
The power distribution network comprises a combination of
groove gap waveguides (GGW) and ridge gap waveguides
3
(RGW), but some particularities should be emphasized. Firstly,
according to how this array is fed, the three desired patterns
(one sum pattern and two difference patterns) are obtained.
The comparator network of the lower layer (Fig. 1(c)) plays
a key role here and will be explained in the next section.
Secondly, the first two power dividers of the corporate-feed
network seen from port 3 are GGW-GGW, and from there
on, all dividers are a combination of RGW and GGW. It
is so to achieve the desired phase inversion resulting from
the characteristic response of the E-plane power dividers. As
it is well known, this kind of divider causes a 180◦phase
difference between outputs. After the two first power dividers,
the imbalance will be maintained until the network’s end,
since the subsequent RGW-GGW dividers keep an even phase
distribution. As shown in Fig. 2, after the first two GGW
splitters, the upper part, an array of 4×8 antennas, will have
a 180◦phase difference from the 4×8 array of the lower part
when port 3 is excited. Fig. 3 indicates the output phases of
the GGW-GGW divider in each case. With this distribution
of phases, the ∆Hpattern will be provided. Note that, for
clarity purposes, the entire antenna is not shown in Fig. 2,
only slightly more than half of the antenna. Fig. 3 corresponds
to a detailed view of the central part of the array to visualize
the network and to indicate some important dimensions of the
prototype. It is worth highlighting that two ridges that access
the distribution network laterally are used to carry the signal
from the bottom layer. This occurs on each side of the network
(see Fig. 3). In short, it is a T-magic to feed the corporate
network with the desired phases, either from ports 1 or 2. The
signal coming through the ridges is evenly distributed through
the lateral arms of the GGW and is isolated from GGW coming
from the front.
III. COMPARATOR NETWORK USING RIDGE AND GROOV E
GAP WAVEGUIDES
The fundamental block for achieving the two remaining
radiation patterns is the comparator network located at the
bottom layer. The key to such a compact structure is to avoid
using hybrids to achieve the different phase shifts needed but
to use a combination of RGW and GGW waveguides. These
hybrid networks composed of both types of GW waveguides
were first proposed in [26]. Later, this scheme has been suc-
cessfully used mainly to achieve antennas with very compact
distribution networks [27], [28]. A different approach is taken
here. Three WR-28 ports feed this comparator block from the
rear part. Port 1 feeds a GGW, and port 2 a RGW. The third
one is connected directly to the intermediate piece, accessing
the antenna’s corporate distribution network, as explained in
the previous section. Feeding through port 1 achieves the Σ
pattern, and feeding through port 2 provides the ∆Eone.
Schematically, Fig. 4 illustrates the phase changes through the
comparator network in the lower layer and then in the upper
layer. The clue here is that the RGW-GGW network keeps
a constant phase distribution, so the needed phase inversions
are produced in previous stages. This figure illustrates how the
GGW-GGW and RGW-GGW dividers, and 90◦bends help to
achieve the desired phase distribution. Note that signals from
Fig. 5. Sketch of the comparator network and detail of the T-junction with
RGW and GGW.
(a) (b)
Fig. 6. Manufactured all-metal low-profile monopulse.
ports 1 and 2 meet in a T-junction (Fig. 5), which allows the
two ports to be decoupled. The third port is connected directly
to the upper layer.
Key design parameters include the dimensions of the bed of
nails in which the comparator network is housed. These pins
have a height of 2 mm, a width of 1 mm, and are equispaced
by 2 mm. It is noted that only 3 rows of pins have been placed
around the network because, as it has been demonstrated very
often in literature [29]–[31], they are enough to avoid field
leakage despite the air gap between the pins and the top layer.
As for the GGW, the most relevant dimensions are the width
(wg=1.5 mm) and depth of the groove (hg=6 mm). In contrast,
the key dimensions of the RGW are the height (hr=1.7 mm),
width (wr=1 mm), and the protrusion of the RGW into the
GGW (lr=0.9 mm). Also, a capacitive coupling window has
been used to improve the impedance matching of this divider
(ws=0.5 mm and ls=1 mm). To sum up, the intermediate layer
is fed at three different points. Port 3 feeds the corporate
network directly and provides an 8×8 array, each vertical half
having a different phase. The bottom layer is responsible for
achieving the appropriate phase to each horizontal half of the
array to obtain the sum or difference pattern, as the case may
be (port 1 or port 2).
IV. EXP ER IM EN TAL RESULTS
The experimental results of the prototype are presented now.
Fig. 6 shows the fabricated prototype. Note that only four
screws are used to assemble the whole structure. The addi-
tional holes are alignment pins, only used for the fabrication
process. Fig. 7 corresponds to the reflection coefficients of
the different ports; it can be seen that the Snn-parameters
(n=1,2,3) remain below −10 dB in the range of interest.
Both simulated and measured curves are shown. The isolation
4
29 29.2 29.4 29.6 29.8 30 30.2 30.4 30.6 30.8 31
−40
−30
−20
−10
0
Frequency (GHz)
Reflection Coeff. (dB)
S11 simulated S22 simulated S33 simulated
S11 measured S22 measured S33 measured
Fig. 7. Measured and simulated reflection coefficient of each port.
29 29.2 29.4 29.6 29.8 30 30.2 30.4 30.6 30.8 31
−60
−40
−20
0
Frequency (GHz)
S-parameters (dB)
S12 simulated S13 simulated S23 simulated
S12 measured S13 measured S23 measured
Fig. 8. Measured and simulated S12 , S13 and S23.
−50 0 50
−40
−20
0
Angle (degrees)
Amplitude (dB)
port 1 port 3
(a)
−50 0 50
−40
−20
0
Angle (degrees)
Amplitude (dB)
port 1 port 2
(b)
Fig. 9. Normalized simulated sum and difference radiation patterns at the
center frequency (30 GHz): (a) XZ-plane (H-plane) and (b) YZ-plane (E-
plane).
between ports, both simulated and measured, is shown in
Fig. 8. An isolation parameter better than −30 dB is achieved
in all cases. Radiation patterns finally provide a clear picture
of the actual performance of the antenna. Fig. 9 shows the
normalized simulated plots at the center frequency (30 GHz)
of the sum and difference patterns. Fig. 10 shows the exper-
imentally measured patterns at different frequencies. Finally,
as seen in summary Table IV, this antenna has a directivity
consistent with its size (26.9 dBi) and exhibits good null depth
(−30 dB). Note that the antenna cannot compete in height
with substrate-based antennas but is very competitive in terms
of radiation efficiency. To conclude that, the most relevant
contribution of this work over previous ones relies on its high
efficiency given by its simple few-layer all-metal architecture
without the need for complex networks.
V. CONCLUSIONS
This letter presents a monopulse antenna based on Gap
Waveguide technology operating in Ka-band. The antenna
is made only in three layers and exhibits a low thickness,
−80 −60 −40 −20 0 20 40 60 80
−40
−30
−20
−10
0
θ(degrees)
Amplitude (dB)
xp 29
xp 30
xp 31 [GHz]
co 29
co 30
co 31 [GHz]
(a)
−80 −60 −40 −20 0 20 40 60 80
−40
−30
−20
−10
0
θ(degrees)
Amplitude (dB)
xp 29
xp 30
xp 31 [GHz]
co 29
co 30
co 31 [GHz]
(b)
−80 −60 −40 −20 0 20 40 60 80
−40
−30
−20
−10
0
θ(degrees)
Amplitude (dB)
xp 29
xp 30
xp 31 [GHz]
co 29
co 30
co 31 [GHz]
(c)
−80 −60 −40 −20 0 20 40 60 80
−40
−30
−20
−10
0
θ(degrees)
Amplitude (dB)
xp 29
xp 30
xp 31 [GHz]
co 29
co 30
co 31 [GHz]
(d)
Fig. 10. Normalized measured radiation patterns according to the excited
port. Port 1→Σ: (a) XZ plane and (b) YZ plane. (c) Port 2 →∆Eand (d)
port 3 →∆H. Note that in the difference patterns (subfigs. (c) and (d)), the
Σpattern has been displayed at 30 GHz for reference.
TABLE II
PER FOR MA NCE O F SOM E RE CEN TLY PR OPO SED L OW-PR OFIL E MO NOP ULS E
AN TEN NAS .
Ref. Type Band Dir. N. RE. Lay. FM.
[16] GW RLSA W 32.3 -40 70%4 No
[18] Metasurface Ka 31.2 -22 42%2 No
[24] GW Array W 30.5 -40 55%4 Yes
[32] Radial LWA K 26 -22 n/a 2 No
This work GW Array Ka 27 -30 80%3 Yes
Dir: Directivity [dBi]; N: Null Depth [dB]; RE: Realized Radiation Effi-
ciency (IEEE) (%) [33]; Lay: Number of layers; FM: Full-Metal structure.
considering it is fully metallic. The experimental results agree
well with the simulated results, even more so considering how
sensitive the phase control is in these arrays. The antenna
presents a directivity close to 27 dBi, a radiation efficiency
above 80%, and stable patterns for a frequency range from 29
GHz to 31 GHz. The prototype could be a suitable solution
for compact millimeter-wave tracking systems.
VI. ACK NOWLEDGEMENTS
Work funded by the Agencia Estatal de Investigacion of
Spain, project PID2019-107688RB-C22; and the University of
Alicante of Spain, project GRE20-06-A.
5
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