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# Compact Integrated Full-Duplex Gap Waveguide-Based Radio Front End For Multi-Gbit/s Point-to-Point Backhaul Links at E-Band

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## Abstract and Figures

This paper presents the design and realization of a high data rate radio front-end module for point-to-point backhaul links at E-band. The design module consists of four vertically stacked unconnected metal layers without any galvanic and electrical contact requirements among the building blocks, by using gap waveguide technology. The module components are a high-gain array antenna, diplexer, and circuitry consisting of a transmitter (Tx) and a receiver (Rx) monolithic microwave integrated circuits (MMICs) on a carrier board, which is successfully integrated into one package with a novel architecture and a compact form. The diplexer consists of two direct-coupled cavity bandpass filters with channels at 71–76 GHz and 81–86 GHz with a measured return loss of 15 dB and an isolation greater than 50 dB. A wideband $16\times 16$ slot array antenna with a measured gain of more than 31 dBi is used to provide high directivity. The measured results show that the packaged transmitter provides a conversion gain of 22 and 20 dB at 76 and 86 GHz, respectively, with an output power of 14 and 16 dBm at 1-dB gain compression point, at the same frequencies. The packaged receiver shows an average conversion gain of 20 dB at 71–76-GHz and 24 dB at 81–86-GHz bands. A real-time wireless data transmission is successfully demonstrated with a data rate of 8 Gbit/s using 32-quadrature amplitude modulated signal over 1.8-GHz channel bandwidth with spectral efficiency of 4.44 bit/s/Hz. The proposed radio front end provides the advantages of low loss, high efficiency, compact integration, and a simple mechanical assembly, which makes it a suitable solution for small-cell backhaul links.
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1
A Compact Integrated Full Duplex Radio Front-end
For Multi-Gbit/s Point-to-Point Backhaul Links at
E-band
Abbas Vosoogh, Milad Shariﬁ Sorkherizi, Vessen Vassilev, Ashraf Uz Zaman, Zhongxia Simon He, Jian Yang,
Ahmed A. Kishk, Fellow, IEEE, and Herbert Zirath, Fellow, IEEE
Abstract—This paper presents the design of a high data rate
band. The design module consists of four vertically stacked
unconnected layers without any galvanic and electrical contact
requirements among the building blocks, by using gap waveguide
technology. A high gain array antenna, a diplexer, and a circuitry
consisting of a transmitter (Tx) and a receiver (Rx) monolithic
microwave integrated circuits (MMICs) on a carrier board are
successfully integrated in one package with a novel architecture
and a compact form. The diplexer consists of two direct-coupled
cavity bandpass ﬁlters with channels at 71-76 GHz and 81-86 GHz
with a measured return loss of 15 dB and an isolation greater
than 50 dB. A wideband 16×16 slot array antenna with measured
gain of more than 31 dBi is used to provide high directivity. The
measured results show that the packaged transmitter provides a
conversion gain of 22 dB and 20 dB at 76 GHz and 86 GHz,
respectively, with an output power of 14 dBm and 16 dBm
at 1-dB gain compression point, at the same frequencies. The
packaged receiver shows an average conversion gain of 20 dB
at 71-76 GHz, and 24 dB at 81-86 GHz bands. A wireless data
transmission is successfully demonstrated with a data rate of
6 Gbit/s using 64 quadrature amplitude modulated signal over
1.35 GHz channel bandwidth with spectral efﬁciency of 4.4
of low loss, high efﬁciency, compact integration, and a simple
mechanical assembly, which makes it a suitable solution for small
Index Terms—integration, frequency division duplex (FDD),
millimeter-wave, gap waveguide, slot array antenna, wireless
communication.
I. INTRODUCTION
THE increasing demand for higher data trafﬁc and speed
in cellular mobile networks leads to a need for higher
bandwidth and data rate in the backhaul side of the network.
Nowadays, smart phones are not only used for voice call,
but also employed for applications that involve an extensive
data consumption. For example, for a typical fourth-generation
(4G) cellular network the required backhaul data exceeds
250 Mb/s [1]. There is a demand for ultra-high data rate
backhaul point-to-point wireless links as a ﬂexible and cost-
effective alternative to ﬁber optic networks to provide multi
A. Vosoogh, A. Uz Zaman, and J. Yang are with the Electrical Engineering
Department at Chalmers University of Technology, Gothenburg, Sweden (e-
mails: abbas.vosoogh@chalmers.se).
M. Shariﬁ Sorkherizi and A. A. Kishk are with the Electrical and Computer
Engineering Department, Concordia University, Montreal, Canada.
V. Vassilev, Z. S. He and H. Zirath are with the Department of Microtech-
nology and Nanoscience at Chalmers University of Technology, Gothenburg,
Sweden.
Gbit/s speed for the ﬁfth-generation (5G) wireless cellular
networks. The millimeter-wave frequency band (30-300 GHz),
has recently got a lot of attention to provide high capacity
backhaul wireless links with high speed and low latency.
Millimeter-wave wireless communication systems suffer
from high path loss, atmospheric absorption and they are
vulnerable to weather conditions and precipitation [1]. All this
limits the hop length to a couple of kilometers at millimeter-
wave frequencies. However, an atmospheric window exists at
E-band (71-76 GHz and 81-86 GHz) with low atmospheric
attenuation of around 0.4 dB/km [1]. This makes the E-band
a potential candidate to provide multi-Gbit/s data transfer and
complements the conventional wireless links that operate in
the microwave frequency band. In [2] path loss measurements
during ten months versus weather conditions for a 1 km link at
71-76 GHz is presented. The measured attenuation for the rain
rate of 20 mm/h and 40 mm/h in one kilometer is estimated to
be 10 dB and 16.4 dB at 86 GHz, respectively, which are with
good agreement with the ITU-R attenuation model in [3].
Lot of effort has been devoted for developing a complete
transmitter (Tx) and receiver (Rx) chipsets with different
technologies at E-band during the past few years. In [4] an
E-band Tx in SiGe BiCMOS technology is presented with an
average conversion gain of 23 dB over 71-86 GHz frequency
band and maximum 1-dB compression output power of 15
dB at 72 GHz. A wireless data transmission of 10.12 Gbit/s
with 64-QAM modulation over 2 GHz channel is demon-
strated using bench-test measurement setup. In [5] 6 Gbit/s
data transmission by aggregating 4 channels with 625 MHz
bandwidth has been demonstrated by using 8PSK modulation
with bench-test measurement setup. A 2.4 bit/s/Hz spectral
efﬁciency has been achieved in the 81-86 GHz frequency band.
A 10 Gbit/s data transmission of 16-QAM modulated signal
at 70/80 GHz band is shown in [6] with spectral efﬁciency
of 2 bit/s/Hz. Packaged eWLB SiGe transceiver chipsets are
presented in [7] for wireless backhaul application in V- and E-
band. A maximum data transmission of 20 Gbit/s with spectral
efﬁciency smaller than 5 bit/s/Hz using 32-QAM modulation
is reported in that work.
This paper presents an E-band full duplex radio front-
end module for multi-Gbit/s point-to-point wireless link ap-
plications. We present a compact integration of passive and
active components, as well as a system packaging solution
based on gap waveguide technology with a ﬂexible mechanical
assembly. An E-band radio transceiver module consisting of
2
Feed-network
(backside)
Diplexer
(backside)
Carrier board
Cavities
#1
#2
#3
#4
Fig. 1. Conﬁguration of the proposed compact radio front-end.
a high gain array antenna, a diplexer, and Tx/Rx chipsets
(all in one package) is designed with capability of sending
and receiving data simultaneously at the 71-76 GHz and the
81-86 GHz bands with a frequency division duplex (FDD)
transmission scheme. The design procedure and performance
veriﬁcation of each building block, i.e., diplexer, 16×16 slot
array antenna, circuitry, and packaging, are here presented.
In this paper we have shown the fascinating packaging and
integration features that gap waveguide technology offers to
build up a complex system with simple mechanical assembly.
Section II presents the overall conﬁguration and charac-
teristics of the proposed E-band integrated radio front-end
module by using gap waveguide technology. The design and
performance evaluation of a special hybrid diplexer-splitter is
presented in Section III. In Section IV the design and experi-
mental evaluation of a 16×16 array antenna and its integration
with the diplexer are described. Section V deals with the
carrier board design and the Tx and Rx MMICs packaging
and performance evaluation. The implementation of a point-
to-point wireless link with the proposed compact integrated
radio modules is demonstrated in Section VI. Finally, some
concluding remarks are given in Section VII.
II. IN TE GR ATED RA DI O FRON T-END MODULE
The conﬁguration of the proposed E-band integrated radio
front-end is shown in Fig. 1. The module consists of a tightly
integrated high gain array antenna, a 5th order diplexer, and
Tx and Rx chipsets on a carrier board in four distinct layers,
where each layer has different functionality. The layers are
vertically stacked up and integrated in a compact form. The
proposed module has total dimensions of 110 mm×90 mm
×8.5 mm.
The bottom layer acts as a carrier printed circuit board
(PCB) for two complete highly integrated GaAs transmitter
and receiver MMICs. We have used commercially available
Tx MMIC from Gotmic AB (part No. gTSC0023) with on-
wafer 25 dB gain and 16 dBm 1-dB compression gain output
(a) (b)
(c) (d)
Fig. 2. Detail illustration of each layer. (a) Radiating slots. (b) Cavities to
feed the slots on topside of layer 3. (c) View of the bottom side of layer 3
containing the corporate feed-network. (d) Diplexer on the bottom side of
layer 2.
power. The Tx has a direct conversion architecture with a ×6
frequency multiplier. It covers 71-86 GHz frequency band with
IF frequency bandwidth of DC-12 GHz in input LO frequency
band of 11.8-14.3 GHz. Two receiver MMICs are used to cover
the frequency band of 71-76 GHz (part No. gRSC0012) and
81-86 GHz (part NO. gRSC0013) form the same manufacturer.
The Rx MMICs are highly integrated chipsets with a ×6
frequency multiplier, an in-phase and quadrature (IQ) mixer, IF
and low noise ampliﬁers with around 25 dB conversion gain
and 6 dB noise ﬁgure. The Tx and Rx MMICs are placed
on the carrier board that consists of 100 µm Liquid Crystal
Polymer (LCP) substrate on a 1 mm copper plate. More details
of the design and layout of the carrier board are presented in
Section V.
A 5th order hybrid diplexer-splitter is designed on the
bottom side of layer 2 in groove gap waveguide (GGW). The
Tx and Rx MMICs are wire-bonded to microstrip lines and
the RF signals are coupled to the diplexer via a microstrip
to GGW transition. We have tried to minimized the length
where the RF signal propagates within the microstrip lines to
reduced the high dielectric and conductive losses of microstrip
transmission lines. The designed diplexer has two channels (at
71-76 GHz and 81-86 GHz) with 5 GHz bandwidth. In order
to achieve a compact design, the output of the diplexer is
directly connected to a power divider, so that the diplexer acts
3
1
k12
k13
3
k13
4 5 06 7 8 9 10
k7
Port 1
Port 4
Port 3
Port 2
k2k3k4k5k6k8k9k10 k11 k12
11
12
13
14
k14
k15
k14
k15
Fig. 3. Coupling diagram of a hybrid diplexer-splitter with a 5thorder divider
network and 5thorder channel ﬁlters.
TABLE I
OPTIMIZED PARAMETERS OF THE COUPLING MATRI X OF TH E HYBRID
DIPLEXER-SPLI TT ER WI TH A DIVIDER NET WO RK (RE FE RS TO FI G. 3)
Parameter value Parameter value (GHz)
k10.0932 f080.057
k20.0626 f173.051
k30.0434 f273.252
k40.0415 f373.367
k50.0482 f473.583
k60.0959 f574.806
k70.0801 f683.295
k80.0461 f783.366
k90.0398 f883.289
k10 0.040 f983.277
k11 0.0577 f10 83.264
k12 0.0886 f11 79.57
k13 0.0106 f12 80.241
k14 0.0201 f13 79.587
k15 0.0203 f14 80.241
as a ﬁrst stage of a power division for corporate feeding of
the array of slots on the top layer. Fig. 2(d) shows the hybrid
diplexer-splitter on the backside of layer 2.
A distribution feed network is designed on the backside
of layer 3.The outputs of the diplexer are coupled to the
next layer, i.e., the feed network, via a right-angle GGW to
rectangular waveguide transition. The feed network is deigned
in ridge gap waveguide (RGW) in order to accommodate the
corporate feeding of radiating elements of the array antenna
in a limited available space. We have used a 2×2 cavity-
backed slot subarray as radiating element of the designed array
antenna. The radiating part consists of 8×8 subarrays (16×16
slots in total), where they are fed by the same phase and
amplitude. The subarrays are formed in two distinct layers, i.e.
the cavities (placed at top side of layer 3) and the radiating
layer (on layer 4). The topside of layer 3 contains 8×8 cavities,
that feed the 16×16 slots of layer 4 uniformly. The geometry
of the feed network and the radiating elements are shown in
more detail in Fig. 2(a), (b), and (c).
In the proposed module shown in Fig. 1, the different layers
do not need to have any electrical and galvanic contact, and a
small gap between each layer would not affect the electrical
performance of the module. All ﬁeld leakages and unwanted
modes have been suppressed by using a stopband produced
by an electromagnetic bandgap (EBG) structure realized in
gap waveguide technology for the ﬁrst time in [8], [9].
Therefore, despite of the H-plane split-block conﬁguration of
the proposed module, electrical and galvanic contacts among
the layers are not required.
In gap waveguide technology a guiding structure, such
as groove [10], ridge [9], or inverted microstrip line [11],
is created between two parallel plates by using an EBG
structure, i.e., a periodic pin texture, to control the direction of
propagation and preventing any ﬁeld leakage without the need
of electrical contact between the building blocks. This provides
a ﬂexible and cost-effective fabrication, and mechanical as-
sembly. The advantages of this technology to design high gain
millimeter-wave array antennas [12]–[16], bandpass ﬁlters
[17]–[24], array antenna and passive components integration
[25]–[27] have been shown in the past few years.
In the present work, we try to show the advantages of
and packaging point of views. Due to the complexity of the
proposed integrated radio front-end module, we have initially
designed and veriﬁed the performance of each building block
separately. The following sections present the detailed design
and measured performance for each different part.
III. HYBRID DIPLEXER-SPL IT TE R DESIGN
In this section, the design and optimization of the hybrid
diplexer-splitter that is used in the module is explained. We try
to keep this section brief and refer the reader to the references
for more detailed technical discussion. The proposed design is
based on the prototype developed in [28] and [25]. However,
due to the required wideband performance of the diplexer
for the current work, we had to expand the design to a
more complicated geometry, and as a result, some minor
modiﬁcations were required to the original methodology. The
diplexer channels are required to be allocated at 71-76 GHz
and 81-86 GHz with 20 dB return loss and provide at least
50 dB isolation. This translates to 19.2% of overall fractional
bandwidth for the diplexer. Such a wide bandwidth requires
strong couplings especially in the divider section of the device.
After calculating the required polynomial degrees of the ﬁlters
to achieve the isolation, we tried to use a similar geometry
as the one shown in [28], with a single resonator node as a
divider to implement the required electrical speciﬁcations. By
doing like this, we understood that this geometry is unable to
provide the required bandwidth for the common couplings. To
overcome this issue, we expanded the divider network to be
constituted of ﬁve resonators as shown in Fig. 3 to decrease
the couplings for each element. Port 1 and 2 are the outputs
of the channel ﬁlters while port 3 and 4 are the inputs of the
divider section.
The next step is to calculate the lumped element network
of the hybrid diplexer-splitter. In [28] we proposed a goal
function that by applying an optimization routine we can
extract the Coupling Matrix (CM) of the diplexer-splitter,
while eliminating the spurious pole between ports 3 and 4.
Since here we need a divider with an equal 3 dB power
division, the divider section is symmetric with respect to the
center. It is possible to design the hybrid diplexer-splitter with
any desired power division value by removing this symmetry
4
Port 4
Port 3
Port 1
Port 2
Upper lid
Coupling Pins
Divider network
12345
12
78910
6
14
13
11
0
Fig. 4. Geometry of the proposed hybrid diplexer-splitter.
Frequency (GHz)
0 20 40 60 80 100 120 140
0
500
1000
1500
2000 Mode 1
Mode 2
stopband
df
ar
gr
pr
Fig. 5. Dispersion diagram of the inﬁnite periodic pin unit cell (df= 0.7 mm,
ar= 0.64 mm, gr= 0.04 mm, and pr= 1.32 mm).
and modifying the goal function to represent the division
characteristic. The instruction to choose the correct initial
values in order to achieve a fast convergence of the routine
is already available in [28] and [25]. The calculated CM is
shown in TABLE I. To avoid the ambiguity about the coupling
bandwidth, the de-normalized k coupling is presented along
with the resonant frequencies.
The calculated CM is meant to be implemented by using
the geometry shown in Fig. 4. The structure is based on
groove gap waveguide that consists of a semi-periodic array of
metallic pins along with a metallic top plate. The top plate is
transparent in Fig. 4 for a better visualization of the inner part
of the design. Metallic pin dimensions are designed to create
a bandgap that can conﬁne the ﬁeld in the cavities without
the need of galvanic contact among the top and lower plate.
Fig. 4 shows the dispersion diagram of a pin unit cell with
inﬁnite periodic boundary conditions, by using the Eigenmode
solver CST Microwave Studio EM simulation software. The
given periodic pin unit cell provides a stopband that covers
the whole frequency band of interest.
The resonators are created by removing two adjacent pins,
and the resonant frequency is controlled by moving the
surrounding pins closer or away to the center of the cavity,
thereby by changing the volume of the cavities. The height
of the pins remains constant except for the pins that are
11
14
13
12
0
Shorting pins
De-tuned resonators
Active resonators
Port 4
Port 3
4 5 6 7
(a)
(b)
Fig. 6. Sub-circuits that are used to implement the CM to the physical model.
(a) Sub-circuit 1 for calculating the divider network cavities dimensions. (b)
Sub-circuits 2 and 3 for calculating the channel ﬁlter cavities dimensions.
Frequency (GHz)
68 70 72 74 76 78 80 82 84 86 88
(dB)
-60
-50
-40
-30
-20
-10
0
|S11|CM
|S22|CM
|S31|,|S41|CM
|S32|,|S42|CM
|S11|CST
|S22|CST
|S31|,|S41|CST
|S32|,|S42|CST
Fig. 7. Response of the coupling diagram in Fig. 3 with the parameters of
TABLE I, along with the EM model response.
responsible for controlling the couplings. The coupling value
between each cavity is controlled by the height of a short pin
between the resonators.
To translate the lumped model of Fig. 3 to the EM model in
Fig. 4, an efﬁcient technique is required since gap waveguide
technology has a complex structure and therefore optimizing
with full-wave simulations is time consuming. The method
proposed in [29] is based on dividing the structure in many
sub-circuits and using the delay response of each sub-circuit
to extract the CM of them. By using the Space Mapping of
[30], each sub-circuit is optimized in an iterative procedure
that can converge in a few iterations. Later, dimensions of the
cavities and coupling pins from all the sub-circuits are used to
generate the complete model. Doing so, without performing
5
Fig. 8. Fabricated diplexer.
(a) (b)
Fig. 9. Simulated E-ﬁeld within the pin texture of the proposed diplexer at
(a) 73.5 GHz, and (b) 83.5 GHz.
any full-wave EM optimization, the geometry yields a very
close response compared to the designated CM.
We have used three sub-circuits to calculate all the dimen-
sions of the proposed hybrid diplexer-splitter in Fig. 4. Sub-
circuit 1 consists of resonators 0, 4-7 and 11-14 which include
the divider section and two cavities from each channel as
shown in Fig. 6 (a). The rest of the resonators are detuned by
having shorting pins at their centers. In the sub-circuits 2 and 3
(Fig. 6 (b)), the remaining cavities for each ﬁlter are contained
in the circuits. It is important to note that sub-circuit 2 and 3
can be solved at a same time, because they are isolated from
each other by the detuned cavities between them. Sub-circuits
are chosen in a way that they have an overlap between each
on the last resonator of each. This is further explained in [29].
After calculating all the dimensions by following the procedure
in [25] and [29], the electric response of the diplexer-splitter is
simulated which is shown in Fig. 7. We used CST Microwave
Studio to solve the EM model. Fig. 7 shows the response of
the CM compared to the optimized EM model performance.
In order to verify the design procedure and simulations, a
prototype is fabricated by Computer Numerical Control (CNC)
milling in aluminum, as shown in Fig. 8. All the inputs of the
fabricated diplexer-splitter are connected to standard WR-12
waveguide ﬂanges. Since the height of the pins (1.28 mm) are
smaller than the height of the standard rectangular waveguide
(1.55 mm), we have tapered the inputs in order to reach
the same height, as shown in Fig. 8. The simulated E-ﬁeld
distribution within the pin texture of the designed diplexer at
73.5 GHz and 83.5 GHz are shown in Fig. 9. It can be seen
that the energy is well conﬁned within the resonators and there
is no ﬁeld leakage, in spite of no electrical contact between
the pin texture and the top lid.
Fig. 10 shows the measured an simulated performance of
Frequency (GHz)
66 70 74 78 82 86 90 92
(dB)
-70
-60
-50
-40
-30
-20
-10
0
Sim. Meas.
|S32|,|S42|
|S31|,|S41|
|S11||S22|
|S12|
(a)
Frequency (GHz)
70 72 74 76 78 80 82 84 86 88
Transmission Coecient (dB)
-6
-5
-4
-3
-2
-1
0
Sim. Ch. 1
Sim. Ch. 2
Meas. Ch. 1
Meas. Ch. 2
(b)
Fig. 10. (a) Measured and simulated performance of the designed diplexer.
(b) Zoomed view of the simulated and measured transmission coefﬁcient of
the two channels.
the fabricated diplexer. The measured results are in a very
good agreement with the simulated ones. The measured input
reﬂection coefﬁcients of the two channels are below -17 dB.
The measured isolation between the two input ports (|S12|)
is smaller than -50 dB. The measured isolation between
the two channels (|Sj1|,|Sj2|) is also better than -50 dB.
Fig. 10(b) shows that the designed diplexer has a low loss
performance. The measured transmission coefﬁcients for both
of the channels are better than -3.5 dB, which in an ideal
lossless situation should be -3 dB. Therefore, the fabricated
prototype shows a maximum 0.5 dB insertion loss. It is worthy
to mention that in the simulated results aluminum is considered
as the constructed material.
IV. ANT EN NA DESIGN
We have designed a planar array antenna to provide a
compact structure and high directivity. We have previously
presented several gap waveguide based slot array antennas
in different frequency bands in [12], [14], [16], [25]. The
radiating element of the array and the design procedure are
similar to the mentioned references. In this section we give an
overview of the designed slot array and diplexer and antenna
integration, followed by experimental veriﬁcations.
Fig. 11 shows the conﬁguration of the designed array
antenna. It consists of 16 ×16 uniformly excited slots which
provide a gain higher than 31 dBi in the 71-86 GHz frequency
band. The array antenna shown in Fig. 11 (a) is formed by four
6
(c)(a) (d)
# 2
# 1
Cavity-feed layer
Slots
# 4
# 3
RGW feed network
RGW to RW
transition
GGW power divider
Cavity-backed unit cell
ridge line
# 3 # 1
ls
ws
hsc
Slots
l
w
L
W
lc
wc
wr
GGW to RW
transition
(b)
70 mm
lsc
wsc
hc
Fig. 11. The designed multi-layer 16×16 slot array antenna. (a) Different layers conﬁguration. (b) 2×2 cavity-backed slot subarray. (c) Distribution ridge
gap waveguide feed network. (d) Bottom layer power divider for experimental evaluation of the designed array.
Port2
Port1
wg
Top lid
dg
pg
aglst
hst1
hst2
wst1
wst2
stoff
Port2
Port1
Top lid
pr
wr
df
lb
wb
hb
dr
ar
(a) (b)
Fig. 12. Conﬁguration of the proposed, (a) Groove gap waveguide (GGW) to
rectangular waveguide, and (b) Ridge gap waveguide (RGW) to rectangular
waveguide transitions.
Frequency (GHz)
65 70 75 80 85 90
S-parameters (dB)
-30
-20
-10
0
S21
S11
Frequency (GHz)
65 70 75 80 85 90
S-parameters (dB)
-30
-20
-10
0
S21
S11
(a) (b)
Fig. 13. Simulated performance of the designed transitions. (a) GGW to RW
transition. (b) RGW to RW transition.
stacked up layers, where the layers do not need any electrical
contact among them. A 2×2 cavity-backed slot element is used
as a subarray, to provide enough space to design a corporate
feed network, as shown in Fig. 11 (b). The designed array
contains 8×8 subarrays in total. The slots placed on layer 4,
i.e. the radiating layer, are fed by cavities on layer 3, as shown
in Fig. 11 (a). A ridge gap waveguide (RGW) feed network
on the backside of layer 3 feeds the cavities via coupling
apertures. The conﬁguration of the designed feed network is
shown in Fig. 11 (c). The feed network is divided into two
parts, where each of them excites half of the radiating slots.
The reason for this is that the designed array is going to
TABLE II
DIMENSIONS OF THE DESIGNED AR RAY ANTE NNA (R EF ERS T O FIG. 1 1
AN D FIG. 1 2)
Parameter Value (mm)
Subarray
Subarray width in x direction (L) 6.48
Subarray width in y direction (W) 6.23
Width of the slot (ws) 1.2
Length of the slot (ls) 2.2
Width of the slot cavity (wsc) 2.5
Length of the slot cavity (lsc) 2.63
Height of the slot cavity (hsc) 0.98
Thickness of the slot layer (ts) 0.35
Length of the cavity (l) 5.65
Width of the cavity (w) 4.82
Height of the cavity (hc) 0.75
Width of the coupling slot (wc) 0.82
Length of the coupling the slot (lc) 2.1
RGW Feed network
Width of the ridge (wr) 0.8
Height of the ridge (dr) 0.47
Air gap between the pins and the upper lid (gr) 0.04
Pins period (pr) 1.32
Width of the pins (ar) 0.64
Height of the pins in RGW (df) 0.7
GGW to RW transition
Width of the pins in GGW (ag) 0.83
Height of the pins in GGW (dg) 1
Air gap in GGW (gg) 0.1
Pins period in GGW (pg) 1.7
Length of the step (lst) 1.6
Height of the step1 (hst1) 0.41
Height of the step2 (hst2) 0.86
Width of the step1 (wst1) 0.55
Width of the step2 (wst1) 0.7
Step offset (stoff ) 0.35
RGW to RW transition
Length of the metal block (lb) 1.77
Width of the metal block (wb) 0.67
Width of the metal block (hb) 0.81
be integrated with a diplexer which provides the ﬁrst power
division stage in order to achieve a compact integration. With
7
Flange layer
#1
Diplexer layer
#2
Cavity-feed layer
#3
#4 Input ports
Fig. 14. Fabricated antenna-diplexer prototype.
Frequency (GHz)
65 70 75 80 85 90 95
Reection Coecient (dB)
-25
-20
-15
-10
-5
0
Meas.
Sim.
Fig. 15. Comparison of measured and simulated input reﬂection coefﬁcient
of the designed 16×16 slot array.
rge purpose of testing the array antenna separately, we have
designed a groove gap waveguide power divider as shown
in Fig. 11 (c). This is made just for performance evaluation
and this layer will be later replaced by the designed diplexer-
splitter.
Since the presented antenna has a multi-layer architecture,
low loss interconnection and transitions are needed. Fig. 12
shows the two designed transitions used as interconnections
of the different layers. The geometry of the proposed vertical
transition between a rectangular waveguide (RW) and a groove
gap waveguide is shown in Fig. 12 (a). A 90E-Plane
bend is formed by a metal step at the end of the GGW
line. This transition is used to couple the energy from the
diplexer-splitter to the upper layer, which is the feed network
of the array antenna. Fig. 12 (b) illustrates the rectangular
waveguide (RW) to ridge gap waveguide (RGW) to match
the TE10 dominant mode of rectangular waveguide to the
quasi-TEM mode of the RGW. The simulated S-parameters
of the proposed transitions after optimization are presented
in Fig. 13. Both transitions show a wideband impedance
matching performance with reﬂection coefﬁcients bellow -20
dB over the frequency band 65-90 GHz. The dimensions of
the designed array antenna are presented in Table II.
Fig. 14 shows a fabricated prototype of the integrated
antenna-diplexer manufactured by CNC milling in aluminum.
The layers are assembled and kept in place by using few
screws. The antenna-diplexer prototype has two WR-12 stan-
dard ﬂanges at the backside. The layer 2 (diplexer layer) in
Fig. 14 is replaced by the fabricated GGW power divider layer
that is shown in Fig. 11 (d), in order to verify the performance
of the antenna alone as well.
The simulated and measured input reﬂection coefﬁcients
of the designed 16×16 slot array antenna are presented in
Fig. 15, showing a good agreement between simulations and
Frequency (GHz)
65 70 75 80 85 90 95
Reection Coecient (dB)
-25
-20
-15
-10
-5
0
Meas. Ch. 1
Meas. Ch. 2
Sim. Ch. 1
Sim. Ch. 2
Fig. 16. Measured and simulated performance of the integrated antenna-
diplexer.
Angle (deg)
-90 -60 -30 0 30 60 90
Relative Amplitude (dB)
-60
-40
-20
0Sim. Co-pol
Meas. Co-pol
Meas. X-pol
Angle (deg)
-90 -60 -30 0 30 60 90
Relative Amplitude (dB)
-60
-40
-20
0Sim. Co-pol
Meas. Co-pol
Meas. X-pol
(a) (b)
Fig. 17. Simulated and measured radiation patterns of the fabricated antenna-
diplexer at center of channel 1 at 73.5 GHz. (a) E-plane, (b) H-plane.
Angle (deg)
-90 -60 -30 0 30 60 90
Relative Amplitude (dB)
-60
-40
-20
0Sim. Co-pol
Meas. Co-pol
Meas. X-pol
Angle (deg)
-90 -60 -30 0 30 60 90
Relative Amplitude (dB)
-60
-40
-20
0Sim. Co-pol
Meas. Co-pol
Meas. X-pol
(a) (b)
Fig. 18. Simulated and measured radiation patterns of the fabricated antenna-
diplexer at center of channel 2 at 83.5 GHz. (a) E-plane, (b) H-plane.
measurements. The measured refection coefﬁcient is below -
11.5 dB in a frequency band of 70-86.5 GHz, which shows a
wide impedance bandwidth of 21%.
Fig. 16 shows the simulated and measured reﬂection coef-
ﬁcient of the manufactured integrated antenna-diplexer. There
is a certain discrepancy in the measured results with respect
to the simulations. The measured input reﬂection coefﬁcient
of the upper channel (81-86 GHz) shows a wider passband
bandwidth. This could be due to assembly errors, since the
fabricated diplexer layer prototype was slightly bent because
of mechanical tensions in the fabrication process. Due to the
large surface of this layer, the ﬂatness of the layers cannot be
guaranteed by few screws. However, this is not a problem
in smaller work pieces, such as the fabricated diplexer of
Section III.
The far-ﬁeld radiation performance of the fabricated
antenna-diplexer is measured in a far-ﬁeld anechoic chamber.
The simulated and measured radiation patterns of the fabri-
cated antenna-diplexer are presented in Fig. 17 and Fig. 18 at
the center frequencies of the two channels. As can be seen, the
simulated and experimental results are in a good agreement.
8
Frequency (GHz)
65 70 75 80 85 90 95
Gain (dBi)
-55
-45
-35
-25
-15
-5
5
15
25
35
Sim.
Ch. 1
Ch. 2
Meas. V-band
Meas. W-band
(a)
Frequency (GHz)
65 70 75 80 85 90 95
Gain (dBi)
26
27
28
29
30
31
32
33
34
80%
70%
60%
w/o diplexer
(b)
Fig. 19. (a) Simulated and measured gains of the fabricated antenna-diplexer.
(b) Zoomed view of simulated and measured gains. The dashed lines are the
directivities with 80%, 70%, and 60% aperture efﬁciencies.
The fabricated prototype shows good co-polar radiation pat-
terns in E- and H-planes, with a measured cross-polarization
better than -35 dB. Fig. 17 and Fig. 18 demonstrates that
although the designed antenna has a multi-layer architecture
with several interconnections and transitions, the radiating
slots are excited correctly.
Fig. 19 shows the simulated and measured gains of the two
inputs of the manufactured antenna-diplexer. The measurement
is done with two V-band and W-band measurement setup to
cover the frequency band of interest. The measured results
show 55 dB and 50 dB isolation in gain in the 71-76 GHz
and 81-86 GHz frequency bands, respectively. A more detailed
measured and simulated gain performance of the fabricated
antenna-diplexer and the simulated gain of the array antenna
without the diplexer is illustrated in Fig. 19 (b). The maximum
directivity from an aperture with the same aperture size of the
designed antenna with different efﬁciencies is also presented in
Fig. 19 (b). The simulated antenna without the diplexer shows
80% antenna efﬁciency. The measured integrated antenna-
diplexer shows average 31 dB and 32 dB gains in channel
1 and channel 2 at 71-76 GHz na d 81-86 GHz, respectively,
with an antenna efﬁciency of around 65%. It is expected that
0.5 dB (as shown in Fig. 10 (b)) drops in the gain and having
lower antenna efﬁciency is due to the extra loss of the diplexer.
V. PCB CARRIER BOAR D DESIGN
Fig. 20 shows the designed PCB board to mount the Tx and
Rx MMICs and supplies the required DC bias voltages. The
Tx and Rx dies are attached and wire bonded to the board,
Tx MMIC
IF for Tx
Wire-bond Compensation
LO for Rx
IF for Rx
Rx MMIC
LO for Tx
diplexer footprint
DC
DC
DC
DC
Metal step of the transition
Fig. 20. Carrier board consisting of Tx and Rx MMICs, wire-bond compen-
sation, IF and LO coaxial ports, and microstrip to GGW transitions.
as shown in Fig. 20. To minimize the effect of the wire-bond
and reduce the loss, a matching section is used at the RF
input/output of Rx/Tx MMICs to compensate the inductance
of the wire-bond. We have used Liquid Crystal Polymer (LCP)
substrate that provides a stable dielectric constant and low loss
tangent at millimeter wave frequency. The substrate consists of
50 µm Rogers Ultralam 3850HT and 50 µm Rogers Ultalam
3908 bonding ﬁlm on a 1 mm copper plate. The PCB is gold
plated afterwards to be able to apply wire-bonding. In [31],
[32] the electrical properties of LCP substrate is measured and
estimated, with dielectric constant of 3.2 and loss tangent of
around 0.005 at 80 GHz. We have used these values in the
modeling and all simulations.
The Tx and Rx MMICs have direct conversion architecture
with differential IQ mixer based on GaAs technology. Dif-
ferential IF (0-12 GHz) inputs/outputs and LO inputs (11.8-
14.3 GHz) are provided with coaxial interfaces. The length
of the microstrip lines that are connecting the IF coaxial
connectors to the MMICs are equal to 90 mm and 42 mm
for Tx and Rx, respectively. Furthermore, the LO path for the
Tx and Rx MMICs are 35 mm and 47 mm. The measured
insertion loss of a 50 microstrip line at 3 GHz and 15
GHz in [31] are estimated to be around 0.01 dB/mm and 0.04
dB/mm, respectively. Therefore, the maximum loss of 1.88
dB is expected for the Rx LO line (largest microstrip line at
higher frequency) at 15 GHz.
The substrate is removed around the footprint of the
diplexer, which is on the next layer of the module as explained
before. The carrier board also includes the two metal steps of
the GGW to rectangular transitions (Fig. 12). The arrangement
of the different parts of the designed carrier board is illustrated
in Fig. 20. In this section, the detailed design of the main
building blocks of the carrier board, such as a transition
from microstrip line to groove gap waveguide (GGW) and
a compensation network for wire-bond interconnection, are
presented. Moreover, the performance of the packaged Tx and
Rx MMICs is also evaluated.
9
wtm wg
ltc
wtc
dtc
ltm
a
p
d
Port 1
Port 2
cavity
wp
lp
wms
Matching
pins
Fig. 21. Microstrip to groove gap waveguide transition geometry. (ltc =
2.57 mm, wtc = 0.62 mm, dtc = 0.76 mm, wg= 3.1mm, wtm =
2.34 mm, wp= 1 mm, lp= 0.33 mm, ltm = 1.2mm, a= 0.65 mm,
d= 1 mm, and p= 1.26 mm)
Frequency (GHz)
70 75 80 85 90
S-parameters (dB)
-25
-20
-15
-10
-5
0
S21
S11
Fig. 22. Simulated S-parameters of a single microstrip to GGW transition.
A. Microstrip to GGW transition design
Fig. 21 shows the designed transition between a microstrip
line to groove gap waveguide (GGW). Similar transitions with
different arrangement are presented in [33] and [34] at V-
and W-band, respectively. The proposed transition is formed
in two distinct parts, where the energy from the microstrip
line couples to the GGW via a resonant cavity. The GGW
line is created by a pin texture on the upper layer and a lid
on the lower layer. The substrate is LCP (r=3.2) with the
thickness of 100 µm. An E-plane probe is extended into the
cavity and the quasi-TEM mode of the microstrip is matched
to TE10 mode of GGW by using the resonant characteristic of
the cavity. The length of the cavity is around λg/2at the
lower operating frequency (70 GHz). The cavity has been
made within the 1 mm thick copper plate. The copper plate
acts as the ground plane of the microstrip line and the top
lid of GGW. Two pins in the upper layer with a distance
of around λg/4at the lower operating frequency from the
E-plane probe are extended into the waveguide channel in
order to add an extra reﬂection and improve the impedance
matching of the transition. The dimensions of the cavity, the
position of the E-plane probe, and the position and length
of the matching pins are optimized to achieve a broadband
impedance matching. The detailed geometry of the transition
Side view cut
Brass pieces
8 mm
(a)
(b)
Fig. 23. (a) Conﬁguration of back-to-back transition between microstrip
line and GGW. (b) Fabricated back-to-back microstrip to GGW transition
prototype.
with optimized parameter values are illustrated in Fig. 21.
The simulated performance of the single transition is shown
in Fig. 22. The proposed transition covers sufﬁciently the
frequency band of interest (71-86 GHz) with a reﬂection
coefﬁcient of below -18 dB. A back-to-back transition is
fabricated in order to verify the performance of the designed
transition. Fig. 23 shows the geometry of the back-to-back
transition and the fabricated prototype. The pin texture in
the upper layer, that forms the groove gap waveguide line,
is fabricated in aluminum with CNC machining. A 8 mm
long microstrip line is fabricated on 100 µm LCP substrate
with 1mm thick copper base. To simplify the fabrication, the
cavities are realized by ﬁrst making through holes with the
required dimensions in the 1 mm copper plate, and afterwards
the holes are closed from the backside with small metal pieces.
We have used brass pieces and conductive silver paste to
enclose the holes and from the cavities, as shown in Fig. 23(a).
The simulated and measured performance of the fabricated
back-to-back transition is shown in Fig. 24. The measured
reﬂection coefﬁcient is below -12 dB over the frequency band
69-87 GHz. The fabricated prototype shows higher insertion
loss than the simulation. A measured average insertion loss
of -1.5 dB has been achieved over the frequency band 71-86
GHz, where the simulated insertion loss is -1 dB over the same
frequency band. The extra loss could be due to the enclosure
of the cavities with conductive silver paste and extra substrate
losses.
The measured results in [31] shows that the insertion loss of
a 50 microstrip line on a 100 µm LCP substrate is around
0.11 dB/mm at 80 GHz. Therefore, by subtracting the loss
10
Frequency (GHz)
67 71 75 79 83 87 90
S11 (dB)
-25
-20
-15
-10
-5
0
Meas.
Sim.
(a)
Frequency (GHz)
67 71 75 79 83 87 90
S21 (dB)
-5
-4
-3
-2
-1
0
Meas.
Sim.
(b)
Fig. 24. Simulated and measured performance of back-to-back microstrip to
GGW transition. (a) Reﬂection coefﬁcient. (b) Transmission coefﬁcient.
lcop1
lcop2
wcop1
wcop2
50 mm
Port 2
Port 1
wire-bond
100 mm
50 mm
Fig. 25. Conﬁguration of compensation matching network for wire-bond
interconnection. (lcop1= 0.48 mm, wcop1= 0.09 mm, lcop2= 0.49 mm,
and wcop2= 0.1mm)
of the 8 mm microstrip line of the fabricated back-to-back
transition, the measured insertion loss of a single transition is
estimated to be around -0.31 dB.
B. Wire-bond compensation network
Wire bonding is a conventional interconnection approach
to electrically connect MMIC to microstrip or coplanar wave-
guide (CPW) transmission lines. However, using this approach
becomes tricky by increasing the operating frequency. The
wire-bond introduces an extra inductance that needs to be
compensated in order to achieve a good impedance matching,
especially at high frequencies. Fig. 25 shows the matching
section that we have used to compensate the wire-bond effect.
Frequency (GHz)
67 72 77 82 87 90
S11 (dB)
-20
-15
-10
-5
0with compensation
without compensation
Frequency (GHz)
67 72 77 82 87 90
S21 (dB)
-2
-1.5
-1
-0.5
0
with compensation
without compensation
(a) (b)
Fig. 26. Simulated S-parameters of wire-bonded 50 µm microstrip lines
in GaAs and LCP substrates with and without compensation. (a) Reﬂection
coefﬁcient. (b) Transmission coefﬁcient.
DC board
180o coupler
90o couplers
Fig. 27. Tx and Rx MMICs performance evaluation measurement setup.
A T-shape microstrip section with a capacitive impedance is
used to compensate the inductance that is introduced by wire-
bonding. We have assumed that a 50 microstrip line on a
50 µm GaAs substrate (εr= 12.5) is connected to a microstrip
line on a 100 µm substrate with a dielectric constant of 3.25 by
a low proﬁle wire-bond. The simulated S-parameters with and
without compensation are presented in Fig. 26. The materials
are considered loss less, and the losses in Fig. 26 (b) are
only due to mismatch and radiation losses. Fig. 26 (a) shows
that the reﬂection coefﬁcient improves from around -8 dB to
below -15 dB over the frequency band 70-88 GHz by using
the compensation matching network.
C. Tx and Rx modules performance
The performance of the Tx and Rx MMICs are measured
to characterize and evaluate the packaging and assembly. The
measurement setup is shown in Fig. 27. The IF and LO are
provided by a Keysight N5241A PNA-X network analyzer
and RF input/output is connected to a VDI WR-12 extender.
The IF port from the network analyzer is connected to a
180hybrid coupler which provides two out-of-phase outputs
with equal amplitude. The outputs of the 180hybrid are
connected to two 90hybrid couplers to provide differential
IQ for input/output of Tx/Rx modules. An extra part with a
GGW line is also fabricated for measurement purposes. The
RF output/input of the Tx/Rx module is coupled to the GGW
line via the microstrip to GGW transition, and afterwards to
a WR-12 rectangular waveguide, as shown in Fig. 27. In the
presented measurement results of this section, the losses due to
the IF an LO microstrip lines on the PCB board, the microstrip
to GGW transition, wire-bond, and the GGW to rectangular
waveguide transition are included.
11
Frequency (GHz)
71 74 77 80 83 86
S11 (dB)
-30
-25
-20
-15
-10
-5
0
Fig. 28. Measured small signal reﬂection coefﬁcient at output of Tx.
Frequency (GHz)
71 74 77 80 83 86
CG (dB)
-30
-20
-10
0
10
20
30
40
USB
LSB
Fig. 29. Measured conversion gain of the signal (upper sideband) and the
image (lower sideband) versus frequency of Tx for IF (3 GHz, -15 dBm) and
LO (8 dBm).
Pin (dBm)
-30 -25 -20 -15 -10 -5
Pout (dBm); Gain (dB)
-10
0
10
20
30 Gain
Pout
CP1dB
83.5 GHz
73.5 GHz
83.5 GHz
73.5 GHz
Fig. 30. Measured conversion gain and output power of Tx versus input
power at 73.5 GHz and 83.5 GHz.
Frequency (GHz)
71 74 77 80 83 86
P1dB (dBm)
0
5
10
15
20
Fig. 31. Measured output power of Tx at 1-dB gain compression point versus
frequency.
Fig. 28 shows small signal output matching of the Tx. The
measured conversion gains of upper sideband (USB) and lower
sideband (LSB) of the Tx are shown in Fig. 29. We have
used a ﬁxed IF of 3 GHz with output power of -15 dBm
Frequency (GHz)
68 74 80 86 90
S11 (dB)
-20
-15
-10
-5
0
Frequency (GHz)
68 74 80 86 90
S11 (dB)
-20
-15
-10
-5
0
(a) (b)
Fig. 32. Measured small signal reﬂection coefﬁcient at the input of Rx. (a)
Frequency (GHz)
68 74 80 86 90
CG (dB)
-20
-10
0
10
20
30
USB
LSB
Frequency (GHz)
68 74 80 86 90
CG (dB)
-20
-10
0
10
20
30
USB
LSB
(a) (b)
Fig. 33. Measured conversion gain of the signal (upper sideband) and the
image (lower sideband) versus frequency of Rx for IF at 3 GHz, RF input
power of -40 dBm, and LO power of 8 dBm. (a) Lower band (71-76 GHz)
TABLE III
SUMMARY OF THE MEASURED PER FOR MAN CE O F THE IN TE GRAT ED
Parameter 71-76 GHz 81-86 GHz
Diplexer insertion loss (dB) 0.5 0.5
Antennas+diplexer gain (dBi) 31 31.5
Microstrip to GGW transition loss (dB) 0.31 0.31
Transmitter gain (dB) 24 21
Transmitter P1dB (dBm) 14 16
Receiver noise ﬁgure (dB) 5.5 5.5
on-wafer measurements
and swept the LO frequency with LO output power of 8
dBm in these measurements. The measurement results show
an average 23 dB conversion gain with good ﬂatness and
a measured image rejection of more than 30 dB over the
frequency band of 71-86 GHz.
The measured output power and gain of the Tx module
versus input power at 73.5 GHz and 83.5 GHz are presented
in Fig. 30. The results show gains of 24 dB and 21 dB at 73.5
GHz and 83.5 GHz, respectively. The 1-dB gain compression
points are also illustrated, with input power at -9 dBm and -4
dBm at 73.5 GHz and 83.5 GHz respectively. The measured
output power of the Tx at 1-dB gain compression versus
frequency is presented in Fig. 31. The Tx module provides
a high P1dB output power around 14 dBm at 71-76 GHz, and
16 dBm at 81-86 GHz frequency bands.
The small signal RF input reﬂection coefﬁcients of the two
Rx MMICs on two carrier boards are shown in Fig. 32. The
measured conversion gains of the Rx modules for IF frequency
of 3 GHz and LO power of 8 dBm are shown in Fig. 33.
Fig. 33(a) shows a measured conversion gain of 20 dB with
an image rejection of more than 20 dB over the frequency
12
PRBS
2.5 GHz
Gray
coding
16/32/64 QAM
maping
RRC pulse
shaping
I/I
Q/Q
IF
I I Q Q
Tektronix AWG7102
Agilent E8257D Agilent E8257D
Teledyne Lecroy Osc.
IF
IF
AGC LPF &
Down-sampling
STR &
Down-sampling
Match
Filter
EQ
Carrier & PahseRcovery
EVM calc.
RF Front-end Modules
6 m
Fig. 34. Sketch of the data transmission test setup.
band of 71-76 GHz. Similarly, the measured conversion gain
of 24 dB and image rejection of 30 dB have been obtained for
the other Rx MMIC over the frequency band of 81-86 GHz.
The designed module has the ability of full duplex wireless
communication, i.e. to transmit in 71-76 GHz or 81-86 GHz
bands, and receive in one of the lower or upper bands
depending on which Rx MMIC is used in the carrier board. We
have fabricated two integrated radio front-end modules, one
with gRSC0012 MMIC to receive in the 71-76 GHz frequency
band, and another one with gRSC0013 chipset to receive in the
81-86 GHz band. A summary of the measured performance
of the fabricated prototypes is presented in Table III.
Based on the characterized performance in Table III, the
signal power given by (1) and the noise power given by (2)
Psig =PT x + 2 ×GAnt LF S LRain LAtm LM(1)
Pnoise = 10log10(K T B) + N FRx (76.7@B=1.5GH z )
(2)
where PT x is the transmitter output power, GAnt is the Tx and
Rx antenna gain, LF S is the free space path loss given by (3),
LRain is the loss due to rain (16.4 dB for 40 mm/h), LAtm
is the atmospheric attenuation (0.4 dB/km), LMis margin
(4 dB as suggested in [35] for the 71-86 GHz band), K
is the Boltzmann’s constant (1.38×1023 W/(K.Hz)), Tis
the receiver temperature (290 K), Bis the transmitted signal
bandwidth, and NFRx is the receiver’s noise ﬁgure.
LF S = 20log10(d[km]) + 20log10 (f[GHz ]) + 92.45 (3)
For a transmitting signal with 2 GHz bandwidth, a maxi-
mum hop-length of around 800 m is expected for 16-QAM
modulation with theoretical Signal-to-Noise Ratio (SNR) of
20.5 dB at a bit error rate (BER) of 106[35]. The spec-
tral efﬁciency can be improved by using higher modulation
Fig. 35. Photograph of data transmission test setup.
schemes. However, the hop-length (d) needs to be reduced
in order to achieve the required higher SNR. For a 64-QAM
modulation, to obtain a BER of 106an SNR of 26.5 dB
is required [35]. This decreases the expected hop-length to
200 by considering 6 dB back-off in P1dB output power of
the Tx to have the required linearity.
A wireless data transmission has been performed by sending
SSB IF signal. The measurement setup is shown in Fig. 34.
QAM modulated IQ signals at 2.5 GHz center IF frequency are
produced by a Tektronix AWG7102 arbitrary waveform gen-
erator (AWG). A Pseudo-Random Binary Sequence (PRBS-9)
pattern is mapped to gray-coded QAM modulated symbols.
We have used root-raised cosine (RRC) ﬁlter for pulse shap-
ing with a roll-off factor of 0.35. The AWG’s outputs are
connected to the Tx IF ports of the integrated radio front-
end module, where it up-converts and sends the data at E-
band. The LO frequency is provided by an Agilent E8257D
signal generator. Another module receives and down-converts
the signal to IF, where it is connected to a real-time Teledyne
Lecroy oscilloscope with 100 GHz bandwidth. We have used
the build-in Teledyne VGA software to demodulate and ana-
lyze the received data. A separate signal generator is used to
provide LO for the RX module. The two modules are separated
only 6 m due to the limited space in lab measurements. The
gain and output power of the Tx MMIC can be controlled by
a VGA voltage. Therefore, to avoid saturating the Rx at the
other side of the link, we have decreased the Tx output power
and as well as slightly misaligned the two antennas.
The constellation diagrams for different modulations are
presented in Fig. 36. A maximum of 6 Gbit/s data transmission
with 16-QAM modulation is achieved with an error-vector-
magnitude (EVM) of 9.5%.
A summarized measured link performance for different
modulation schemes is presented in Table IV. A maximum
data rate of 8 Gbit/s is achieved by using 16-QAM modulation
with spectral efﬁciency 2.96 b/s/Hz. A compression with
some published articles has been presented in Table V. In all
published works in Table V a wireless link and data transmis-
sion are demonstrated by either using bench-test measurement
setup, or by designing the circuitry, packaging, diplexer, and
13
7.6 mV/div
6.2 mV/div
(a) (b)
10 mV/div
9.7 mV/div
(c) (d)
Fig. 36. Constellation diagrams of the received signal. (a) 16-QAM with 4
Gbit/s at 72.1 GHz, (b) 16-QAM with 8 Gbit/s at 83 GHz, (b) 32-QAM with
5 Gbit/s at 74.6 GHz, and (b) 64-QAM with 6 Gbit/s at 84.6 GHz.
TABLE IV
SUMMARY OF OVER -TH E-AIR DATA TRANSMISSION TES T.
Center Freq.
(GHz) Modulation Symbol (Baud)/
Data rate (Gbit/s)
Spectral eff.
(bit/s/Hz)
EVM
(%)
72.1
16
32
64
1/4
1/5
1/6
2.96
3.7
4.44
5.3
5.9
4.76
73 16 2/8 2.96 9.5
74.6
16
32
64
1/4
1/5
1/6
2.96
3.7
4.44
5.25
5.8
4.8
82.1
16
32
64
1/4
1/5
1/6
2.96
3.7
4.44
5.1
5.65
4.7
83 16 2/8 2.96 9.3
84.6
16
32
64
1/4
1/5
1/6
2.96
3.7
4.44
5.25
5.6
4.73
antenna in separate modules, which increase the cost and size
of the system.
VII. CONCLUSION
A compact integrated solution for multi-Gbit/s data trans-
mission for point-to-point wireless link applications at E-band
has been presented. A full duplex FDD radio front-end module
has been designed by integrating a high gain array antenna,
a diplexer, and RF circuitry consisting of Tx/Rx MMICs in
one package. The proposed solution has a novel architecture,
consisting of four vertically stacked layers with a simple
mechanical assembly. This is due to the use of gap waveguide
technology, which eliminates electrical and galvanic contact
requirement in waveguide structures and provides an effective
system packaging solution.
TABLE V
COMPARISON BETWE EN STATE-O F-T HE-A RT E-BAN D MULTI -GBIT/S
DATA TRA NSM IS SIO N.
Ref Test setup Data rate-
Modulation
Spectral eff.
(bit/s/Hz
[4] (SiGe) bench test 10.12 (Gbit/s)
64-QAM 5.06
[5] (GaAs) bench test 6 (Gbit/s)
8PSK 2.4
[6] (n. a.) bench test 10 (Gbit/s)
16-QAM 2
[7] (SiGe) bench test 20 (Gbit/s)
32-QAM <5
16-QAM 3.2
This work
8 (Gbit/s)
64-QAM
8 (Gbit/s)
16-QAM
4.4
2.96
The performance of each building block of the designed
module is initially evaluated separately and then two integrated
modules are used to demonstrate a multi-gigabit data trans-
mission. The measurement results show that the integrated
antenna-diplexer prototype has a gain of more than 31 dBi
with an antenna efﬁciency better than 65%, where 0.5 dB
loss is due to the diplexer. A maximum data rate of 8 Gbit/s
was achieved by sending a 16-QAM modulated signal over a
distance of 6 m, with spectral efﬁciency of 2.96 bit/s/Hz. Based
on the summarized measured performance of the fabricated
modules given in Table III, for different modulations of 16-
QAM and 64-QAM the hop-length of around 450 m and 200 m
is expected with 1.5 GHz channel bandwidth. The proposed
integrated module has the ability of sending and receiving
data simultaneously by using an FDD transmission scheme.
A maximum data rate of 24 Gbit/s can be achieved by using
the full potential of the designed module.
The proposed integrated module shows the ﬂexibility and
great potential that gap waveguide technology can offer in
system integration and packaging with high complexity.
ACKNOWLEDGMENT
The authors would like to thank Gapwaves AB for making
the mechanical structure, and Gotmic AB for providing the
Tx/Rx MMICs.
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