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Modular coherent photonic-aided payload receiver for communications satellites


Abstract and Figures

Ubiquitous satellite communications are in a leading position for bridging the digital divide. Fulfilling such a mission will require satellite services on par with fibre services, both in bandwidth and cost. Achieving such a performance requires a new generation of communications payloads powered by large-scale processors, enabling a dynamic allocation of hundreds of beams with a total capacity beyond 1 Tbit s−1. The fact that the scale of the processor is proportional to the wavelength of its signals has made photonics a key technology for its implementation. However, one last challenge hinders the introduction of photonics: while large-scale processors demand a modular implementation, coherency among signals must be preserved using simple methods. Here, we demonstrate a coherent photonic-aided receiver meeting such demands. This work shows that a modular and coherent photonic-aided payload is feasible, making way to an extensive introduction of photonics in next generation communications satellites.
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Modular coherent photonic-aided payload receiver
for communications satellites
Vanessa C. Duarte 1,2, João G. Prata1, Carlos F. Ribeiro1, Rogério N. Nogueira1,3, Georg Winzer2,
Lars Zimmermann2, Rob Walker4, Stephen Clements4, Marta Filipowicz5, Marek Napierała5,
Tomasz Nasiłowski5, Jonathan Crabb6, Marios Kechagias6, Leontios Stampoulidis6, Javad Anzalchi7&
Miguel V. Drummond 1
Ubiquitous satellite communications are in a leading position for bridging the digital divide.
Fullling such a mission will require satellite services on par with bre services, both in
bandwidth and cost. Achieving such a performance requires a new generation of commu-
nications payloads powered by large-scale processors, enabling a dynamic allocation of
hundreds of beams with a total capacity beyond 1 Tbit s1. The fact that the scale of the
processor is proportional to the wavelength of its signals has made photonics a key tech-
nology for its implementation. However, one last challenge hinders the introduction of
photonics: while large-scale processors demand a modular implementation, coherency
among signals must be preserved using simple methods. Here, we demonstrate a coherent
photonic-aided receiver meeting such demands. This work shows that a modular and
coherent photonic-aided payload is feasible, making way to an extensive introduction of
photonics in next generation communications satellites. OPEN
1Instituto de Telecomunicações, Campus Universitário de Santiago, 3810-193 Aveiro, Portugal. 2IHP, Im Technologiepark 25, 15236 Frankfurt (Oder),
Germany. 3Watgrid Lda., Via do Conhecimento, 3830-352 Ílhavo, Portugal. 4aXenic Ltd., Thomas Wright Way, Sedgeeld TS21 3FD, UK. 5InPhoTech
Sp. z o.o., Meksykańska 6 lok. 102, Warsaw 03-948, Poland. 6Gooch & Housego, Broomhill Way, Torquay TQ2 7QL, UK. 7Airbus Defence & Space, Gunnels
Wood Rd, Stevenage SG1 2AS, UK. Correspondence and requests for materials should be addressed to V.C.D. (email:
NATURE COMMUNICATIONS | (2019) 10:1984 | | 1
Content courtesy of Springer Nature, terms of use apply. Rights reserved
The evolution of high-throughput communication satellites
(HTCSs) has been paced by a raw increase in capacity,
obtained by packing more beams within the coverage zone
of the satellite1. So far, adding a new beam has been typically
achieved by introducing at least one more feed to the payload,
placing it according to the position of the beam2. While such a
design paradigm stood for more than half a century, it has now
become obsolete for two main reasons. First, it scales aggressively.
The number of feeds increases not only in proportion to the
number of beams but also inversely to their width, as thinner
beams require larger apertures encompassing multiple feeds3.
Second, user needs are changing at an accelerating rate. Having a
static beam conguration matching user needs at all times during
the entire lifetime of the satellite, typically spanning 15 years,
is nowadays impossible. Consequently, satellite operators
recently started asking for exible satellite payloads1,4, with
satellite QUANTUM being the rst example of a communications
satellite designed with a programmable digital signal processor.
QUANTUM offers a total bandwidth of 3.5 GHz which can be
exibly allocated to eight steerable spot beams. The correspond-
ing capacity is almost 30 times lower than already launched
HTCSs, presently beyond 100 Gbit s1, evidencing that capacity
and exibility could not be simultaneously increased with either
radio frequency (RF) or digital technologies.
A scalable capacity increase and a exible beam conguration
can be both achieved by operating the feeds as a phased array
antenna (PAA), as illustrated in Fig. 1. Without loss of generality,
let us consider a PAA comprising Nfeeds and receiving a set
of N
beams. Each feed now receives all beams, each beam
arriving with a unique delay with respect to other feeds. Such a
property enables separating all N
beams from the Nsignals
produced by the PAA by multiplying these by a matrix with N
Ncoefcients. In practice, such a matrix is implemented by a
processor denominated by beamforming network (BFN), and
each coefcient is implemented by an amplier or attenuator
combined with a phase shifter. In terms of scale, the required
number of feeds increases only inversely to the minimum beam
width, resulting in the lowest possible number of feeds3,5. As for
exibility, a full-scale recongurable BFN is able to adapt to any
set of beams.
While implementing the feeds to operate as a PAA is trivial, a full-
scale implementation of a BFN comprising all N
has so far been too cumbersome to implement either with digital or
analogue signal processors. On the one hand, digitally processing all
signals provided by the PAA requires an unrealistic processing
power of at least 1 Gsa s1per input signal and per beam, over
10 Tsa s1for already launched HTCSs1,3.Ontheotherhand,the
size of an analogue BFN depends on the length of each phase shifter,
which is proportional to the wavelength of the RF signals. This
results in a deadlock: while a BFN processing high-frequency RF
signals is inherently compact, a low-loss implementation is chal-
lenging due to the high frequency of the RF signals. Photonics allows
overcoming such a trade-off. The wavelength of optical signals is
more than 5000 times shorter than a typical Ka-band RF signal,
enabling signicant miniaturization of the BFN. Yet optical wave-
guides, either bre or within a photonic integrated circuit (PIC), are
well known for being low-loss6. Such unique advantage puts pho-
tonics in a leading position for implementing a BFN.
A photonic-aided payload implements the BFN with a pro-
network (OBFN)5,7,8. A miniaturizable OBFN must be identical
to an RF BFN, with an optical phase shifter being equivalent to an
RF phase shifter. Consequently, a miniaturizable OBFN relies on
coherent optical signal processing, which in turn enables coherent
detection3,5,9,10. Coherent detection can also provide heterodyning,
thus enabling RF frequency conversion3,5,11. As a result, RF hard-
ware is assigned only to basic tasks such as amplication and
inverse/output multiplexing3. The main advantage of such an
approach is that it does not change the payload architecture,
allowing to keep mandatory function modularity and redundancy
mechanisms3. However, the sheer scale of the photonic-aided
payload unavoidably results in long optical paths, more than 5000
times longer than RF paths when normalized to the wavelength. As
a result, unavoidable thermal and mechanical gradients imposed to
optical paths produce a slow but random phase drift to the pro-
pagating signals, obliterating beamforming if no action is taken.
Given that there is no such problem in an RF BFN, its conguration
may be performed by a fairly static monitoring and control loop
(MCL). Conversely, an OBFN requires a dynamic MCL, which
nonetheless must be kept simple, scalable and fast just enough to
RF stage
Modulation &
LO generation
RF stage
Rx feeds Tx feeds
Beam 1
Beam 2
Beam 3
Feed 1
Feed 2
Feed N
Fig. 1 Photonic-aided payload receiver. User beams are received by the satellite, aggregated by the payload and re-transmitted to a ground station
connected to the World Wide Web. Inset: the photonic-aided payload receiver up-converts the signals provided by the receiving feeds to optical signals,
separates all beams via beamforming, and down-converts the separated beams back to radio frequency (RF) signals at the right frequency to be re-
transmitted to a ground station by means of heterodyning. A monitoring and control loop (MCL) is used to operate, optimize and stabilize the optical
beamforming network (OBFN)
2NATURE COMMUNICATIONS | (2019) 10:1984 | |
Content courtesy of Springer Nature, terms of use apply. Rights reserved
track phase drifting, in order to be suitable for a large-scale OBFN.
While all stages of a photonic-aided payload have been individually
9,10,1721, frequency
conversion22,23 and coherent detection6,9,24such an MCL has
never been demonstrated, preventing the demonstration of a
scalable, modular and coherent photonic-aided payload.
In this paper, we demonstrate a complete scalable, modular
and coherent photonic-aided payload receiver comprising four
custom-made modules, all aiming for miniaturization and mod-
ularity: two arrays of two GaAs MachZehnder modulators
(MZMs) each, a radiation-hardened erbium-doped multicore ber
amplier (EDMCFA) with seven cores, a 4-by-1 integrated OBFN,
and an MCL for dening and stabilizing the amplitude, delay and
phase of each signal. The receiver was fed with two beams arriving
from different directions, each carrying a 1 Gbit s1quadrature
phase shift keying (QPSK) signal at 28 GHz, being able to separate
one from the other in real time. To the best of our knowledge, it is
the rst demonstration of real-time beam separation ever achieved
by a complete photonic-aided payload receiver.
Self-coherent photonic processor. The main task of a payload
receiver is to down-convert the frequency of an input beam
such that it can be processed by an inverse multiplexer3. Fre-
quency down-conversion is accomplished by heterodyning the
beam with a local oscillator (LO), which is typically derived from
a single reference oscillator3. A photonic-assisted payload receiver
should likewise resort to heterodyne detection, with optical
local oscillators (OLOs) taken from a single reference OLO, thus
resulting in self-heterodyne detection5,20. Self-heterodyne detec-
tion has the important advantage of cancelling laser phase
noise, which relaxes the required laser linewidth, and makes the
proposed MCL possible.
The self-coherent photonic processor at the core of the
proposed photonic-assisted payload receiver complies with such
principle by using a single laser source, as depicted in Fig. 2a. The
processor rst converts input RF signals to the optical domain.
The converted signals are individually amplied, power equalized,
phase shifted, delayed and coherently added into one output
RC Filter
Upconv. to 28 GHz
fref = 10 MHz
26 GHz
2 MZM arrays
f [GHz]
f [GHz]
f [GHz]
f2,LO = 2.6 GHz – 5 kHz
f1,LO = 1.4 GHz – 2 kHz
~ch1 ch1
Test equipment
coherent processor
Monitoring and
control loop
f [GHz]
ADC bandwidth
5 kHz
2 kHz
Amp., EVM, SER
Low-pass filtering
Freq. downconvertion
Arbitrary waveform
Offline processing
f2 = 28.6 GHz
f [GHz]
28 f [GHz]
Digital signal
23 DACs
Extra attenuation [dB]
Extra attenuation [dB]
Extra attenuation [dB]
1 path
2 paths
4 paths
EVM [dB]
4~5 dB
4~5 dB
1 path
4 paths
2 paths
f1 = 27.4 GHz
Fig. 2 Set up and characterization results of the proposed photonic processor. aExperimental set-up. bThe photonic integrated circuit (PIC) is bond-wired
to a printed circuit board (PCB), which provides lateral direct current (DC) access to the phase shifters and to a thermistor, and two radio frequency (RF)
connectors for accessing the differential outputs of the transimpedance ampliers (TIA). The pads of the germanium photodiode (Ge-PD) are bond-wired
to a bias tee, which interfaces the photodiode with the transimpedance amplier (TIA). cAmplitude, error vector magnitude (EVM) and symbol error rate
(SER) of the output signal obtained when adding one, two and all four signals. dRepresentative constellation diagrams of the output signal obtained
without extra attenuation
NATURE COMMUNICATIONS | (2019) 10:1984 | | 3
Content courtesy of Springer Nature, terms of use apply. Rights reserved
optical signal. The resulting signal is coherently detected through
heterodyning, therefore resulting in down-conversion of the RF
frequency20,25. A low-complexity pilot-aided MCL is used to
control and stabilize the system in real time. The experimental
set-up is detailed as follows.
Light from a TLS (tunable laser source) is split by three paths.
In the two upper paths, RF-to-optical conversion is achieved by
means of amplitude modulation done by two arrays of GaAs
MZMs, each with two MZMs13,26. Each array outputs the two
modulated signal to a single bre owingthanks to polarization
multiplexing. The modulated signals are then pre-amplied, noise
ltered and polarization demultiplexed. For each of the four
resulting optical signals there is a dedicated free-space optical
delay line to equalize the relative delays of the modulated signals
within the tuning range of the tunable optical delay lines (TODLs)
of the OBFN. The delay-equalized signals are then boosted by four
of the seven cores of a saturated EDMCFA27. The amplied
signals are equalized in power by variable optical attenuators
(VOAs), and then fed to an integrated OBFN embedded in a
printed circuit board (PCB), as shown in Fig. 2b. Details of the
silicon PIC implementing the integrated OBFN are given in the
Supplementary Note 4. The integrated OBFN comprises four
identical paths, each with an input phase shifter, for adjusting
the phase of each input signal, and a TODL based on a
MachZehnder delay interferometer (MZDI) with variable
coupling ratio and with a tuning range of τ=50 ps20. The four
delayed signals are combined into a processed output signal that,
in turn, is split in two copies. One is routed to an external power
metre, which is connected to the MCL. The other is combined
with an OLO and fed to a Germanium photodiode (Ge-PD),
thus being coherently detected. The resulting electrical signal
is amplied by a transimpedance amplier (TIA) with two
differential outputs, one of which is the electrical output signal of
the photonic processor. The OLO is generated in the purple-
coloured optical path as follows. An I/Q modulator driven by
two RF tones with a frequency of f
and dephased by π/2
produces a frequency-shifted version of the laser signal. The
modulated signal is amplied and combined with the processed
output signal, thus serving as a frequency-shifted optical local
oscillator (FSOLO). Consequently, the processed signal is
frequency down-converted by f
. Further details about the
experimental set-up can be found in the Supplementary Note 3.
An MCL is required for setting up and stabilizing the
amplitude, phase and delay of each input optical signal. A
simple, scalable and low-power MCL should rely only on the
output signals of the processor, and handle as few RF signals as
possible25. The proposed MCL relies on two out-of-band pilot
tones with frequencies f
and f
added to the input RF signals,
thus being transparent to the input RF signals and associated
propagation impairments. As explained in ref. 28, using a pair of
pilot tones enables a simple and precise estimation of the time
delay of a given path, as the time delay is proportional to the
phase difference between pilot tones. Amplitude and phase can be
directly estimated from one of the pilot tones. Given that identical
pilot tones are fed to all input RF signals, and that the MCL takes
as input the output electrical signal, a method for distinguishing
the pilot tones associated with different signals must be
enforced28. A simple method is here proposed, in which weak
dithering tones with different low frequencies (<2 kHz) are
digitally generated and fed to the phase shifters β. As a result, the
output signal comprises the RF signal, pilot tones f
and f
, and
weak dithering tones f
and f
, where f
is the
frequency of the dithering tone fed to the phase shifter βof the
path k. The MCL thus uses both dithering tones f
and f
to estimate the delay of path k, and at least one of such tones
to estimate the amplitude and phase of the corresponding signal.
While the amplitude and time delay of the optical signals are
fairly static, the phase wanders over time as a result of
temperature and mechanical gradients affecting optical bres.
Consequently, the MCL refresh rate should be high enough to
counteract phase wandering, thus providing fundamental stability
when coherently combining optical signals in the OBFN. Such is
the case of the presented set-up, as explained in the Supplemen-
tary Note 1. The impact caused by parasitic phase modulation on
the signal is also discussed in the Supplementary Note 2.
The implementation of the MCL is depicted at the bottom of
Fig. 2a. An RF I/Q demodulator is used to simultaneously down-
convert the spectral content around f
to 2 kHz and around f
5 kHz. The resulting signal, containing all pilot and dithering
tones at low frequency, is digitized by an analog-to-digital
converter (ADC) with a low sampling rate. A digital signal
controller (DSC) is used to control and synchronize the
acquisition of samples by providing a time stamp, allowing
accurate digital down-conversion of all dithering tones to
baseband, and thus phase and delay estimation. Frequency
down-conversion, parameter estimation and control are split
between the DSC and a computer. The control algorithms output
new voltages to be applied to the phase shifters and VOAs, which
are programmed in the respective digital-to-analog converters
(DACs) via the DSC. All components used for implementing the
MCL are low-end commercial off-the-shelf.
We validated the operation of the proposed photonic processor
by setting it up to coherently add four identical RF signals
with equalized power. As shown in Fig. 2a, we programmed an
arbitrary waveform generator (AWG) to generate a QPSK signal
at 1 Gbit s1comprising 640 symbols, pulse-shaped by a raised
cosine (RC) lter with a roll-off factor of 0.25 GHz, with a carrier
frequency of 28 GHz, and with two pilot tones at 28 ± 0.6 GHz
each with an amplitude of 10% of the QPSK signal. Such a signal
was repetitively generated. Four copies of the RF signal were
produced using RF splitters, individually amplied by broadband
ampliers, and fed to the photonic processor. A signal generator
was used to produce a tone with f
=26 GHz. Consequently,
the processed signal is frequency down-converted to 2 GHz. The
electrical output signal of the processor was sampled by a real-
time sampling scope (RTSS), and processed ofine. In all
experiments performed in this work, ofine processing involved
only essential functions: frequency down-conversion to baseband,
removal of the pilot tones through low-pass ltering, down-
sampling to 1 sample per symbol, normalization and de-rotation
of the constellation. Further details about ofine signal processing
can be found in the Supplementary Note 5. VOAs are responsible
for power equalizing the four signals, and also for introducing a
common extra attenuation.
The obtained results are displayed in Fig. 2c, d. As expected,
the amplitude of the output signal increases proportionally to the
number of enabled paths. However, the error vector magnitude
(EVM) is more than halved when the number of enabled paths is
doubled. Although the added signals are identical, these have
different noise sources as these are amplied by different
electrical and optical ampliers. Therefore, coherently adding
the signals increases the signal-to-noise ratio (SNR) of the output
signal, thus explaining why its EVM is more than halved. These
results validate the operation principle of the proposed photonic
Single-beam beamforming. As depicted in Fig. 3a, demonstrating
the proposed photonic processor as a photonic-aided payload
receiver involves connecting the processor to a PAA receiver, and
feeding multiple beams from different directions to the PAA.
However, the processor requires adding pilot tones to the received
signals. The simplest way of adding pilot tones to all signals is to
4NATURE COMMUNICATIONS | (2019) 10:1984 | |
Content courtesy of Springer Nature, terms of use apply. Rights reserved
use a dedicated antenna for transmitting the pilot tones to the
entire PAA receiver. Such approach does not impact the low-noise
amplication stage located right after the PAA receiver, as pilot
tones are by denition much weaker than the input signals. The
amplitude and phase of the pilot tones depend on the position of a
given antenna element of the PAA receiver. Nonetheless, given
that the position of the dedicated antenna is known beforehand,
such a dependency can be calibrated and therefore compensated.
The simplied experimental set-up used for demonstrating
single-beam beamforming is shown in Fig. 3a. For a matter of
simplicity only a single channel of the AWG is used, meaning that
beam and pilot tones are simultaneously generated and
transmitted by a single Tx antenna. Due to power limitations,
the Tx antenna is positioned 15 cm from the PAA receiver, which
comprises four antenna elements uniformly spaced by 20.1 mm.
The photonic processor was congured to coherently add the four
input signals with equalized power and time delay. As a result,
optimized beamforming is obtained without having to recon-
gure the processor when changing the beam launch point. In
order to validate such a statement, the performance of the system
is assessed for three different position of the Tx antenna. As
depicted in Fig. 3a, at position kthe Tx antenna is equidistant
from the antenna elements kand k+1.
The results displayed in Fig. 3bd show that beamforming was
achieved for all positions of the Tx antenna. When taking as
reference a given signal amplitude, the system provides a
performance similar to the observed in Fig. 2c. Such allows to
conclude that the introduction of a wireless link did not produce
any impairment other than the unavoidable free-space path loss.
Assuming that all four paths of the photonic processor provide
identical performances, the output signal with lowest EVM
should be obtained when the PAA receiver receives the maximum
amount of power. Such case corresponds to placing the Tx
antenna at position 2. A higher EVM should thus be obtained for
positions 1 and 3. Both statements are conrmed by the
experimental results. However, the output signal with the highest
amplitude is not obtained when placing the Tx antenna at
position 2, but at position 3. Such observation does not prove any
inconsistency, as a signal with a higher amplitude does not
necessarily have a higher SNR.
Multi-beam beamforming. The demonstration of a exible
photonic-aided payload receiver must assess its capability of
separating multiple beams. In order to achieve so, the experimental
set-up depicted in Fig. 4a is considered. An additional Tx antenna
is connected to the second channel of the AWG for producing the
second beam. The AWG generates two identical RF signals with
parameters as previously dened, but carrying distinct symbol
patterns. Pilot tones are added only to one of the signals. Given
that the photonic processor automatically points the receiver
towards the direction from where the pilot tones originate, the
processor automatically beamforms the beam to which pilot tones
are added. Both Tx antennas should be as far as possible from the
PAA receiver such that the angle of incidence of each beam is
identical for all antenna elements of the PAA receiver. However,
limited transmitting power forces placing the Tx antennas near the
PAA receiver, at about 40 cm, even when using extra broadband
ampliers for boosting the transmitted power. Nonetheless, by
setting the antenna Tx
12 cm apart from the antenna Tx
estimated power of the beamformed beam is 18.8 dB higher than
the other (interfering) beam. The set-up thus poses no physical
restriction to separating one beam from the other. Four cases are
thus considered: beam 1 and pilot tones launched from Tx
(S), beam 1 and pilot tones launched from Tx
and beam 2
launched from Tx
(S) +Tx
(I), beam 2 and pilot tones
launched from Tx
(S), and beam 2 and pilot tones launched
from Tx
and beam 1 launched from Tx
(S) +Tx
The results displayed in Fig. 4b, c show once again that single-
beam beamforming is achieved for both beams, regardless of the
number of added signals. However, such is not the case when
transmitting two beams. When a single path is enabled, the
receiver is unable to distinguish one beam from the other. As both
beams produce a similar RF power at any antenna element of the
PAA, the photonic processor basically adds the two beams with
identical weights, thus corrupting the beam intended to be
beamformed. The activation of two paths enables six different
combinations of pairs of antenna elements. Each combination
results in a different nulling capability. Such explains why beam
separation is achieved in some combinations (e.g. 1 +2 and 3 +
4), and not in others (e.g. 1 +3 and 2 +4). The activation of all
four paths provides clear beam separation as expected. The results
also show that the performance of the various paths is not
identical. Such a non-uniformity in performance in unavoidable,
given that signals input to different paths are processed by
different devices, such as electrical ampliers, modulators and
optical ampliers. Nonetheless, such non-uniformity in perfor-
mance is mitigated as more paths are enabled, and performance
ends up being very similar when all four paths are enabled.
Comparison with state-of-art OBFN. Before discussing the
obtained results, it is important to compare the proposed
photonic-aided payload receiver with state-of-art OBFNs.
Wireless link
15 cm
Extra attenuation [dB]
Extra attenuation [dB]
1 path
2 paths
4 paths
Pos. 1
Pos. 2
Pos. 3
EVM [dB]
1 path 2 paths 4 paths
Position 1Position 3 Position 2
Fig. 3 Demonstration of single-beam beamforming. aSet-up: the Tx antenna is placed at three different positions. bAmplitude and cerror vector
magnitude (EVM) of the output signal when adding one, two and all four signals, for the different positions of the Tx antenna. dRepresentative
constellation diagrams of the output signal obtained without extra attenuation
NATURE COMMUNICATIONS | (2019) 10:1984 | | 5
Content courtesy of Springer Nature, terms of use apply. Rights reserved
The main motivation for photonic implementations of BFNs
was, and still is, the fact that RF phase shifters are bulky, lossy
and increasingly challenging to produce at high RF frequencies.
Therefore, almost all of the research done in OBFNs has focused
on developing photonic phase shifters, able to surpass such
limitations. As explained in the following paragraph, the various
photonic phase shifters can be divided in three generations.
The rst-generation photonic phase shifters consist of mere
optical bres with a length tailored to provide the target RF phase
shift29. Discretely adjustable phase shifting was demonstrated by
switching among optical bres with different lengths30.Thesecond-
generation photonic phase shifters exploit linear and nonlinear
properties of photonic devices. Chromatic dispersion was the most
exploited linear property3133. Dispersive devices such as optical
bres and bre Bragg gratings have a propagation delay that
depends on the wavelength of the input signal. Such a property was
exploited to produce continuously tunable photonic phase shifters
in a single device. In terms of nonlinear properties, slow light was
profoundly exploited as it enables modulating the refractive index
of the medium with the input power3436. Such an effect was used
to induce a tunable phase shift between RF sidebands and optical
carrier of an RF signal modulated onto an optical carrier. These
phase shifters were the rst not to resort to delay lines. The third-
generation photonic phase shifters are based on adjustable optical
ltering. TODLs based on all-pass lters implemented in resonant
and non-resonant interferometers were, respectively, proposed in
refs. 9,10,19,37,38.Bothlters have a periodic frequency response,
which is tuned to provide the correct delay to at least one of the
RF sidebands of the RF signal modulated onto an optical carrier.
The TODLs used in the present work are based on non-resonant
MZDIs. Its operation principle as well as its application to an
OBFN are detailed in ref. 20. Programmable lters based on liquid
crystal on silicon (LCoS) matrices were also proposed both for
providing a TODL and a phase shift between the optical carrier and
While a plethora of phase shifters and TODLs were proposed,
few of these were experimentally demonstrated within an
OBFN10,21,30,32,4143, especially within a receiving stage10,41.
A receiving stage combines signals from the PAA to form one
or multiple beams. Signal combination can be performed
coherently or incoherently. The latter option typically multiplexes
signals into a wavelength-division multiplexing (WDM) signal,
which is then directly detected40,43,44. Despite being a straightfor-
ward approach, it supports a limited number of channels, it
does not resort to coherent detection, and consequently does
not support heterodyne reception. Conversely, coherent signal
combination does not have such drawbacks. In fact, coherent
signal combination enables building an OBFN identical to an RF
BFN, that is, signals are coherently combined without bandwidth
limitations, RF frequency down-conversion is achieved by means
of heterodyne reception, and optical phase shifting is equivalent
to RF phase shifting25,28. The latter advantage enables the use
of the simplest and smallest photonic phase shifterthe optical
phase shifter. However, to the best of the authorsknowledge, an
OBFN resorting to coherent signal combination was demon-
strated only once10, without any active stabilization loop.
A comparison between the proposed system and state-of-art
OBFNs would be desirable concerning key metrics such as size,
weight, power consumption and performance degradation.
However, such a comparison cannot be made as other proposed
systems either lack dimensioning or were not dimensioned for
communications satellites. Nonetheless, the proposed system is
evaluated in detail according to such key metrics in ref. 5.
Future HTCSs will have to do better than just increasing capacity:
they must become capable of providing exible coverage for best
serving fast-evolving user needs. Both requirements can only be
achieved by adopting a recongurable antennaa PAAwhich
in turn requires a massive signal processor, the BFN. While dif-
ferent implementations of the BFN have been discussedanalo-
gue, digital or photonic-aidedthe unique miniaturization and
low-loss capabilities enabled by the latter puts photonics in a
leading position. Nonetheless, a viable photonic-aided payload
must comply with key features such as being scalable, modular,
miniaturizable and having an architecture resembling an RF
payload. Devising a simple, scalable and low-speed stabilization
loop has proved to be the last challenge to overcome before such a
photonic-aided payload could be demonstrated. The work
Wireless link
40 cm
12 cm
1 path 2 paths 4 paths
4 paths
EVM [dB]
Tx1(S) + Tx2(I)Tx2(S)
Tx2(S) + Tx1(I)
2 paths 1 path
4 paths 2 paths 1 path
Fig. 4 Demonstration of two-beam beamforming. aSet-up. bRepresentative constellation diagrams of the output signal obtained when one Tx antenna
transmits the signal beam (S) and the other the interfering beam (I). cError vector magnitude (EVM) of the output signal for different combinations of
enabled paths and beams. All tests were made without introducing extra attenuation
6NATURE COMMUNICATIONS | (2019) 10:1984 | |
Content courtesy of Springer Nature, terms of use apply. Rights reserved
presented in this paper shows that such a challenge was indeed
overcome, paving the way for a photonic revolution in HTCS
Scaling the proposed system to more antenna elements and
more beams depends on the scalability of individual modules.
The photonic BFN is both the most important and most chal-
lenging module to scale, as its complexity is proportional to both
the number of antenna elements and number of beams. On the
one hand, self-heterodyne detection makes its basic elements
phase shifters and couplersequivalent to their RF counterparts,
thus enabling the miniaturization of the BFN by a factor of
50005. On the other hand, as the number of required
phase shifters is proportional to the number of antenna elements
of the PAA and to the number of beams, about 25,000 phase
shifters are required5. Therefore, the development roadmap of a
high-capacity and exible photonic-aided payload should focus
on developing dense arrays of low-speed phase shifters with low
power consumption and low insertion losses.
Even though the focus of the present work is the real-time
demonstration of a modular and coherent photonic-aided pay-
load receiver, such a demonstration included several novelties
worth highlighting: rst-ever use of a EDMCFA in an OBFN,
rst-ever integrated OBFN including a photodiode, rst-ever
demonstration of an OBFN performing RF frequency conversion
by means of heterodyne reception and rst-ever separation of two
beams by an OBFN.
A discussion of future directions should be split in two
parts: system and devices. From a system perspective, although
challenging, scaling the proposed receiver to more antenna ele-
ments and more beams does not require deep modications to
the basic architecture and stages presented in Fig. 2a. None-
theless, although the proposed system is based on a single laser
source, multiple laser sources with different wavelengths can be
used where suitable, as the proposed system is compatible with
WDM. Adapting the proposed architecture to the transmitting
stage would be straightforward, also requiring a simpler MCL,
as N-to-1 signal combination becomes signal splitting.
From a device perspective, we believe that a signicant progress
in PICs is mandatory such that beamforming one beam in a
single PIC can be envisaged. Focus should be given to developing
low-loss interfacing as well as low-loss, compact, low-power
phase shifters.
The method used for interfacing depends on whether WDM is
used. If not, as considered in the present work, it requires N+1
input bres, provided that polarization multiplexing is also not
used. Such a large number of bres should be bundled into a two-
dimensional bre array, which should be precisely aligned with a
set of grating couplers. While such interfaces with these many
inputs/outputs have already been investigated45, developing a
space-qualied package requires a signicant effort. Using WDM
means that only a few interfacing bres are required, which does
not pose a problem in interfacing. However, an OLO must be
allocated for each channel, and all resulting signals have to
be demultiplexed on chip, which requires large and lossy
demultiplexers43. It is therefore likely that a trade-off between
both solutions should prove to be the best solution.
As discussed in ref. 28, silicon PICs offer three kinds of phase
shifters based on different effects: thermo-optic, carrier injection
and carrier depletion. These phase shifters have different trade-
offs among insertion loss, voltage-dependent loss, footprint and
power consumption, all being unacceptably underperforming in
at least one of such parameters. For instance, thermo-optic and
carrier-injection phase shifters consume more than 10 mW28,
which according to the model presented in5puts the power
consumption of phase shifters at the same level of the low-noise
ampliers. Conversely, carrier-depletion phase shifters have
negligible power consumption, but at the cost of being almost
1 cm long and lossy, with over 5 dB of insertion loss. Conse-
quently, a new kind of phase shifter should be developed. Liquid
crystals are known to have a very strong electro-optic effect,
orders of magnitude higher than materials widely used in pho-
tonics such as lithium niobate, enabling very small phase
shifters46,47. They are also very transparent to light, and thus low-
loss, and are routinely integrated in LCoS matrices nowadays
packing over 107phase shifters in <1 cm248. Co-integrating a
reective LCoS matrix on top of silicon PIC using grating cou-
plers to provide vertical interfacing appears to be a promising
approach, allowing to envisage a phase shifter as large as a grating
coupler, typically with 10 × 10 μm246.
Liquid crystals have a slow response on the order of 1 ms.
Although such response is fast enough for compensating phase
wandering, it might be insufcient for the modulation of
dithering tones at frequencies higher than 1 kHz. Nonetheless,
such is not a problem. Even though the number of required phase
shifters βis of N×N
, the number of required dithering tones is
of only N. Consequently, only Nmodulators are required to
produce the dithering tones. In order to avoid parasitic phase
dithering of the RF signal associated with the modulation of the
dithering tones, a suitable modulation scheme would be to resort
to a ring modulator only modulating one of the pilot tones49.
While such a modulation scheme provides intensity modulation
instead of phase modulation, the operation principle behind the
MCL would be the same.
The proposed coherent photonic processor can be generalized
from an N-to-1 weighted adder to a M×Nmatrix multiplier
without changing the MCL. Optical matrix multipliers resorting
to coherent optical signal processing have been proposed as a
promising alternative approach to microelectronic and opto-
electronic articial neural networks, as once the matrix coef-
cients are set the multiplication takes place at the speed of light50.
As a single PIC can pack a very limited number of neurons,
implementing larger and more powerful networks requires mul-
tiple PICs, and therefore modularity. Similarly to the proposed
processor, scaling up to a modular implementation must conserve
coherency between all signals in all modules, also requiring an
MCL. Therefore, the presented work may serve as a starting point
towards a modular coherent photonic neural network.
Monitoring and control loop. As explained in the main text, the MCL sets and
stabilizes the amplitude, phase and delay of each input signal. Here, we detail the
operation of the MCL.
The MCL rst sets the TODLs as described in ref. 28 to minimize the relative
delay among signals. Given that the congured delays remain stable over time, for a
matter of simplicity the MCL congures one TODL at a time, with dithering tones
deactivated. In order to avoid interference from other paths, all paths except for the
one including the TODL being congured are fully attenuated.
Once the TODLs are congured, the power of all signals is equalized as follows.
The power of each signal is measured one at a time, with all other signals fully
attenuated to avoid any measurement error. The power of the weakest signal is
then subtracted a margin of 1 dB, resulting in a reference power P
. Each signal
is attenuated until its power falls within P
, where P
=0.1 dB. At
this point the extra attenuation mentioned in the gures is of 0 dB.
After all signals are again attenuated until the target extra attenuation is
reached, the system is ready to coherently combine all signals. The MCL is then
congured to proceed similarly to a phase-locked loop, with each iteration
described as follows. Dithering tones are simultaneously generated with a default
amplitude of 50 mV, and with a frequency of f
=500(k+1). The DSC then
congures the ADC to capture 128 samples at 64 ksa s1. The dithering tones are
deactivated once all samples have been captured. The delay between the generation
of the dithering tones and the sampling by the ADC was measured to be fairly
constant, with a jitter of only a few. The samples produced by the ADC are
processed by the DSC to extract the dithering tones with the highest frequency, that
is, f
, from which the phase of each optical signal is estimated. Even though
the estimated phases have an undetermined phase reference, such a phase reference
is the same for all estimated phases. As a result, all paths have the same phase if all
estimated phases are identical, regardless of the common value. The objective of the
NATURE COMMUNICATIONS | (2019) 10:1984 | | 7
Content courtesy of Springer Nature, terms of use apply. Rights reserved
MCL thus is how to achieve so by producing the smallest possible ajustment to the
phase shifters β. Given that voltage adjustments are proportional to estimated
phases, the solution is to minimize the variance and average value of the estimated
phases. Such is performed as follows. A vector containing the estimated phases is
rst sent to the computer. Variance is minimized by adding or subtracting 2πto
each estimated phase; if the resulting variance is reduced, the estimated phase is
updated. Once the variance has been minimized, the average of the vector
containing the updated phases is set to zero. The voltage offset to be applied to a
given phase shifter is varied by the estimated phase multiplied by a gain factor,
which by default is 0.2V/π. If the new voltage offset is higher than V
=0.7 V
or lower than V
=1.5 V, it is reset to (V
)/2 =1.1 V. This is
necessary to guarantee that the phase shifters βoperate within voltage limits. The
new voltage offsets are transmitted to the DSC, which updates the DACs associated
with the corresponding phase shifters β. The measured duration of each iteration
and thus the period of the MCL was of 100 ms, dominated by the communication
time between the DSC and the computer, performed via USB.
Data availability
The data that support the ndings of this study are available from the corresponding
author upon reasonable request.
Received: 27 December 2018 Accepted: 29 March 2019
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This work was supported in part by the Fundação para a Ciência e Tecnologia/Ministério
da Educação e Ciência under the Ph.D. Grant SFRH/BD/117444/2016, by Fundo
Europeu de Desenvolvimento RegionalPortugal 2020 partnership agreement under the
project UID/EEA/50008/2013 and by the European Commission through the project
BEACON (FP7-SPACE-2013-1-607401).
Author contributions
M.V.D., R.N.N. and V.C.D. jointly developed the concept. V.C.D., G.W. and L.Z.
designed and fabricated the chip. J.G.P. and C.F.R. implemented the MCL and designed
the layout of the PCB. R.W. and S.C. conceived the array of MZMs. M.F., M.N. and T.N.
conceived the radiation-hard EDMCFA bre and fan-in/fan-out. J.C., L.S. and M.K.
assembled the EDMCFA. J.A. coordinated and gave the inputs for space applications.
V.C.D. and M.V.D. conceived the experiments, performed the measurements and ana-
lysed the data. V.C.D, M.V.D. and R.N.N. wrote the paper. M.V.D., S.C. and R.N.N.
managed and coordinated the project.
Additional information
Supplementary Information accompanies this paper at
Competing interests: The authors led a patent application on a photonic system to
perform beamforming of a radio signal: Miguel V. Drummond, Rogério N. Nogueira and
Vanessa C. Duarte, Photonic beamforming system for a phased array antenna receiver,
PCT/IB2016/052206, led on 18 April 2016. The remaining authors declare no competing
Reprints and permission information is available online at
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... To solve this issue, optical true time delay line (OTTDL) has been in-troduced as an effective solution by steering the beam at a same angle for a wide bandwidth to salve the bandwidth limitation of traditional electronic phased array antennas. By controlling the phases and the amplitudes of OTTDL transmissions, it is possible to achieve beam steering and beam forming [14][15][16]31,42,44,45 , and satellite navigation communications and automotive ranging radars for Ku and K band have attracted people's attention. The present chip-scale programmable silicon photonic processor can be configured as 4-channel OTTDL for 18 GHz microwave signal and demonstrate the beamformer covering the Ku and K band. ...
... It can be seen that the beam steering angle is frequency insensitive and the extinction ratio is below 0.2. Table 1 gives a summary of the representative delay lines for beam steering in terms of the scale, the performance of waveguides and MZSs, the overall power consumption and the footprints 3, [14][15][16]31,44,45 . As shown in Table 1, most of these delay lines were demonstrated with a single channel 14−16 . ...
... As shown in Table 1, most of these delay lines were demonstrated with a single channel 14−16 . There are a few chips with multiple channels, as demonstrated in ref. 3,31,44,45 , where the delay line is routed in either snake-shape 3,31 or S-shape 45 , resulting in large footprints. Furthermore, the waveguides have some high propagation loss, which prevents further improvement of the chip scale. ...
Full-text available
Chip-scale programmable optical signal processors are often used to flexibly manipulate the optical signals for satisfying the demands in various applications, such as lidar, radar, and artificial intelligence. Silicon photonics has unique advantages of ultra-high integration density as well as CMOS compatibility, and thus makes it possible to develop large-scale programmable optical signal processors. The challenge is the high silicon waveguides propagation losses and the high calibration complexity for all tuning elements due to the random phase errors. In this paper, we propose and demonstrate a programmable silicon photonic processor for the first time by introducing low-loss multimode photonic waveguide spirals and low-random-phase-error Mach-Zehnder switches. The present chip-scale programmable silicon photonic processor comprises a 1×4 variable power splitter based on cascaded Mach-Zehnder couplers (MZCs), four Ge/Si photodetectors, four channels of thermally-tunable optical delaylines. Each channel consists of a continuously-tuning phase shifter based on a waveguide spiral with a micro-heater and a digitally-tuning delayline realized with cascaded waveguide-spiral delaylines and MZSs for 5.68 ps time-delay step. Particularly, these waveguide spirals used here are designed to be as wide as 2 µm, enabling an ultralow propagation loss of 0.28 dB/cm. Meanwhile, these MZCs and MZSs are designed with 2-µm-wide arm waveguides, and thus the random phase errors in the MZC/MZS arms are negligible, in which case the calibration for these MZSs/MZCs becomes easy and furthermore the power consumption for compensating the phase errors can be reduced greatly. Finally, this programmable silicon photonic processor is demonstrated successfully to verify a number of distinctively different functionalities, including tunable time-delay, microwave photonic beamforming, arbitrary optical signal filtering, and arbitrary waveform generation.
... In refs. 23,24 , cascaded optical ring resonators and in ref. 25 Mach-Zehnder interferometers with tunable coupling ratios serve as broadband tunable delay lines whereas in ref. 26 switchable waveguide delay lines are used to form a photonic beamformer. Such integrated architectures have the potential to enable scalable and high-yield integrated photonic-assisted TTD phased arrays (or timed arrays), which can address some of the challenges of conventional electronic implementations, but often at a cost of increased complexity and chip area due to the use of TTD elements. ...
... Moreover, in refs. [23][24][25][26] , active mm-wave devices are used to generate the mm-wave carrier in the electrical domain limiting their operational bandwidth and further increasing the complexity and power consumption as the frequency increases. ...
Full-text available
Millimeter-wave (mm-wave) phased arrays can realize multi-Gb/s communication links but face challenges such as signal distribution and higher power consumption hindering their widespread deployment. Hybrid photonic mm-wave solutions combined with fiber-optics can address some of these bottlenecks. Here, we report an integrated photonic-assisted phased array transmitter applicable for low-power, compact radio heads in fiber to mm-wave fronthaul links. The transmitter utilizes optical heterodyning within an electronically controlled photonic network for mm-wave generation, beamforming, and steering. A photonic matrix phase adjustment architecture reduces the number of phase-shift elements from M × N to M + N lowering area and power requirements. A proof-of-concept 2 × 8 phased array transmitter is implemented that can operate from 24–29 GHz, has a steering range of 40°, and achieves 5 dBm EIRP at an optical power of 55 mW without using active mm-wave electronics. Data streams at 2.5 Gb/s are transmitted over 3.6 km of optical fiber and wirelessly transmitted attaining bit-error rates better than 10⁻¹¹.
... Microwave photonics technology brings together the worlds of radio frequency (RF) engineering and optoelectronics [1,2], exhibiting a promising potential in applications such as broadband wireless access networks [3], satellite communications [4], radar [5] and electronic warfare systems [6]. Photonic integration enables a dramatic reduction in the footprint of microwave photonics systems with a high complexity and enhances functionalities and performance, propelling it into real applications [7,8]. ...
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Photonic integration brings the promise of significant cost, power and space savings and propels the real applications of microwave photonic technology. In this paper, a multiband radio frequency (RF) signal simultaneous receiver using an optical bandpass filter (OBPF) integrated with a photodetector (PD) on a chip is proposed, which was experimentally demonstrated. The OBPF was composed of ring-assisted Mach–Zehnder interferometer with a periodical bandpass response featuring a box-like spectral shape. The OBPF was connected to a PD and then integrated onto a single silicon photonic chip. Phase-modulated multiband RF signals transmitted from different locations were inputted into the OBPF, by which one RF sideband was filtered out and the phase modulation to intensity modulation conversion was realized. The single sideband with carrier signals were then simultaneously detected by the PD. A proof-of-concept experiment with the silicon photonic integrated chip was implemented to simultaneously receive four channels of 8 GHz, 12 GHz, 14 GHz and 18 GHz in the X- and Ku-bands. The performance of the integrated microwave photonic multiband receiver—including the receiving sensitivity, the spurious free dynamic range, the gain and the noise figure across the whole operation frequency band—was characterized in detail.
Microwave photonics offers a promising solution for frequency converting microwave signals, however, demonstrations so far have either been bulky fibre implementations or lacked rejection of interfering image signals. Here, we demonstrate the first microwave photonic mixer with image rejection of broadband signals utilising chip-based stimulated Brillouin scattering and interferometry. We demonstrate frequency down-conversion of carrier frequencies ranging from 10 GHz-16 GHz, ultra-high image rejection for a single tone of up to 70 dB, and 100 MHz and 400 MHz wide analogue signals with 28.5 dB and 16 dB image rejection, respectively. Furthermore, we down-convert 200 Mb/s quadrature-phase-shift keying signals with an error vector magnitude as low as -9.6 dB when simultaneously present interfering image signals are suppressed by the mixer.
In this work, we first demonstrate a complete 2-D photonics-aided phased-array antenna (PAA) receiver with so far the largest scale of 8 $\times$ 8 by utilizing highly integrated silicon true-time delay line chips. The prototype utilizes the wavelength-division-multiplexing technology to build the crucial optical beamforming network (OBFN) module so as to dramatically reduce the type and amount of silicon binary integrated optical time delay lines. The measured beam scanning range in the frequency range from 2 to 6 GHz is 90 $^{\circ}$ $\times$ $90^{\circ}$ with a beam-pointing step of 1 $^{\circ}$ . Furthermore, due to the online channelized signal processing module, we systematically characterize sensitivity, RF gain, and dynamic range of the photonics-aided PAA. Considering that we first utilize the photonic integration technology to realize a 2-D PAA demonstrator with the largest scale and the highest technological maturity, we believe that it presents a milestone in the development of future microwave photonics-based radar systems.
Free-space optical (FSO) communication technology is a promising approach to establish a secure wireless link, which has the advantages of excellent directionality, large bandwidth, multiple services, low mass and less power requirements, and easy and fast deployments. Increasing the communication capacity is the perennial goal in both scientific and engineer communities. In this paper, we experimentally demonstrate a Tbit/s parallel FSO communication system using a soliton microcomb as a multiple wavelength laser source. Two communication terminals are installed in two buildings with a straight-line distance of ∼ 1 km . 102 comb lines are modulated by 10 Gbit/s differential phase-shift keying signals and demodulated using a delay-line interferometer. When the transmitted optical power is amplified to 19.8 dBm, 42 optical channels have optical signal-to-noise ratios higher than 27 dB and bit error rates less than 1 × 10 − 9 . Our experiment shows the feasibility of a wavelength-division multiplexing FSO communication system which suits the ultra-high-speed wireless transmission application scenarios in future satellite-based communications, disaster recovery, defense, last mile problems in networks and remote sensing, and so on.
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A lean coherent receiver based on an externally modulated laser is presented. Reception is conducted in virtue of the optical detection capability of a monolithically integrated high-bandwidth electro-absorption modulator and its distributed feedback laser section that serves as local oscillator. The in-line configuration of the receiver with its partial feed-through of the incident data signal to the local oscillator is further exploited for the purpose of optical injection locking. A locking range of 200 MHz is obtained at an input power level of -27.8 dBm. The proposed low-complexity receiver is being evaluated in an optical access scenario through use of a commercial off-the-shelf externally modulated laser device rated for 10G applications. Coherent reception of 1 Gb/s on-off keyed data through exclusive use of analogue signal processing at the receiver is demonstrated at low bit error ratios of 10<sup>-10</sup> and is further shown to be compatible with a loss budget of 40.1 dB at the hard-decision forward error correction threshold. Higher data rates of up to 6.6 Gb/s at a loss budget of 31 dB are facilitated through broadband orthogonal frequency division multiplexing, for which wavelength locking obviates optical frequency offset correction between local oscillator and incident data signal.
In this Letter, we propose a monitoring and control system (MCS) for operating tunable optical delay lines (TODLs), regardless of their operation principle and implementation technology. The monitoring system resorts to two out-of-band pilot tones added to the input optical signal. The amplitude and phase difference between tones are retrieved to the control system, which calculates and applies the TODL control signals. The MCS was validated using a Mach-Zehnder delay interferometer-based TODL, implemented in three different silicon photonic integrated circuits (PICs). The three PICs resort to different kinds of phase shifters based on thermo-optic, carrier-injection, and carrier-depletion effects. The proposed MCS enabled tuning the delay within the entire range of the TODL in all tested PICs. The scalability of the MCS for large-scale photonic beamformers is discussed.
Conference Paper
Beam forming using a 4x1 phased array with an optical feed network for K-band smart antenna systems is presented. Up to 45 degree steering angles are achieved. Wireless transmission throughputs of up to 3 Gbit/s using 32-OFDM subcarriers are achieved for the attained steering angles.
The design and dimensioning of a photonic-aided payload for a multi-beam high-throughput communications satellite is a complex problem in which the antenna, RF and photonic subsystems must be considered as a whole for achieving best performance with lowest mass and power consumption. In this paper, we propose and dimension the receiving stage of a communications satellite comprising a phased array antenna (PAA) feeding a multibeam photonic beamforming system (PBS). The PBS uses a single wavelength and resorts to heterodyne detection such that the retrieved beams are frequency downconverted. End-to-end system modeling shows that the complexity of the PAA and PBS can be traded-o for signal-to-noise ratio (SNR) or power consumption without compromising the beam width. The dimensioning of a realistic scenario is presented, showing that an SNR and beam crosstalk on the order of 20 dB are achievable with a total power consumption below 1 kW for a typical number of 100 antenna elements (AEs).
Microwave phased array antennas (PAAs) are very attractive to defense applications and high-speed wireless communications for their abilities of fast beam scanning and complex beam pattern control. However, traditional PAAs based on phase shifters suffer from the beam-squint problem and have limited bandwidths. True-time-delay (TTD) beamforming based on low-loss photonic delay lines can solve this problem. But it is still quite challenging to build large-scale photonic TTD beamformers due to their high hardware complexity. In this paper, we demonstrate a photonic TTD beamforming network based on a miniature microresonator frequency comb (microcomb) source and dispersive time delay. A method incorporating optical phase modulation and programmable spectral shaping is proposed for positive and negative apodization weighting to achieve arbitrary microwave beam pattern control. The experimentally demonstrated TTD beamforming network can support a PAA with 21 elements. The microwave frequency range is 8~20 GHz; and the beam scanning range is '60.2'. Detailed measurements of the microwave amplitudes and phases are performed. The beamforming performances of Gaussian, rectangular beams and beam notch steering are evaluated through simulations by assuming a uniform radiating antenna array. The scheme can potentially support larger PAAs with hundreds of elements by increasing the number of comb lines with broadband microcomb generation.
Plasmonics provides a possible route to overcome both the speed limitations of electronics and the critical dimensions of photonics. We present an all-plasmonic 116–gigabits per second electro-optical modulator in which all the elements—the vertical grating couplers, splitters, polarization rotators, and active section with phase shifters—are included in a single metal layer. The device can be realized on any smooth substrate surface and operates with low energy consumption. Our results show that plasmonics is indeed a viable path to an ultracompact, highest-speed, and low-cost technology that might find many applications in a wide range of fields of sensing and communications because it is compatible with and can be placed on a wide variety of materials.
Conference Paper
We report on high density optical packaging of high radix (64x64) silicon photonic MEMS switches using pitch-reducing optical fiber array. The footprint of 61-channel optical I/O is as small as 330μm x 280μm.