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Possibilities of further improvement of 1-second fluxgate variometers

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  • Lviv Center of the Institute of Space Research of NAS of Ukraine and SSA of Ukraine

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The paper discusses the possibility to improve temperature and noise characteristics of fluxgate variometers. The new fluxgate sensor with a Co-based amorphous ring core is described. This sensor is capable to improve the signal-noise ratio at the recording short-period geomagnetic variations. Besides the sensor performance, it is very important to create the high stability compensation field, which is canceling the main Earth magnetic field inside the magnetic cores. For this purpose the new digitally controlled current source with low noise level and high temperature stability is developed.
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Possibilities of further improvement of 1-second fluxgate
variometers
Andriy Marusenkov1
1Lviv Centre of Institute for Space Research, Lviv, 79060, Ukraine
Correspondence to: Andriy Marusenkov (marand@isr.lviv.ua)
Abstract. The paper discusses the possibility to improve temperature and noise characteristics of fluxgate variometers. The
new fluxgate sensor with a Co-based amorphous ring core is described. This sensor is capable to improve the signal-to-noise
ratio at the recording short-period geomagnetic variations. Besides the sensor performance, it is very important to create the
high stability compensation field, which is cancelling the main Earth magnetic field inside the magnetic cores. For this purpose
the new digitally controlled current source with low noise level and high temperature stability is developed.5
1 Introduction
Flux-gate magnetometers (FGM) are widely used for measuring weak magnetic fields in geophysical researches. For this,
reducing the FGM own noise is very important, particularly for observatory variometers compatible with 1-second INTER-
MAGNET standard (Turbitt et al., 2013). Next very important task is improving the temperature stability of both zero offset
and transformation coefficient, especially for space magnetometers and geophysical equipment operating in field conditions.10
To reach this, the most important is to improve the FGM sensor. New approaches such as the use of ferrodielectric materials
(Vetoshko et al., 2003), special excitation modes (Vetoshko et al., 2003; Koch and Rozen, 2001; Ioan et al., 2004; Sasada
and Kashima, 2009; Paperno, 2004) can significantly reduce the level of own noise of flux-gate sensor, in particular, down to
0.1 pT Hz0.5at a frequency of 1 Hz (Koch and Rozen, 2001) with further decrease to several tens of fT Hz0.5(Koch and
Rozen, 2001; Koch et al., 1999). Also in the best examples of flux-gate sensors for space research, along with moderate noise15
level, zero offset is practically independent on temperature: their zero drift is within 1 nT in the temperature range -40C to
+65C (Merayo et al., 2005; Nielsen et al., 1995). In this study we try to find possibilities for simultaneous improvements of
noise level and temperature stability of flux-gate variometers for Earth magnetic field measurements. In Sect. 2 we describe
the new flux-gate sensor with significantly reduced noise level. Then, in Sect. 3 the peculiarities of the design of a low noise
and highly stable digitally controlled current source are given.20
2 Development of low-noise fluxgate sensor with amorphous magnetic core
During last fifteen - ten years our main hopes and expectations for decreasing noise level of the fluxgate sensors were linked
with using amorphous magnetic materials instead of crystal permalloy for the sensor core. The very powerful technique to
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Geosci. Instrum. Method. Data Syst. Discuss., doi:10.5194/gi-2017-12, 2017
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suppress magnetic noise of amorphous alloy is a proper annealing procedure. The ten-fold noise level improving is possible by
selection an optimal annealing temperature. However, we found also that the sensor zero offset and its temperature dependence
often considerably degrade after annealing. It was experimentally found that an amorphous alloy with the lower optimal an-
nealing temperature usually could provide lower and more stable zero offset. As a result of this approach a new amorphous ring
core fluxgate sensor (conventionally named FGS32/11) had been recently developed. The sensor consists of a 32 mm diameter5
fiberglass bobbin with 11 turns of Co-based 0.03 mm thick and 3 mm wide tape annealed at 710 K. Besides the excitation
winding it contains the two sectoral measuring winding for sensing two orthogonal components of the magnetic field. The
sensor was tested at the following excitation parameters: driving frequency fex = 7.5kHz, the amplitude of the driving pulses
Hm= 6.8kA m1, and the relative width of these pulses αex = 0.4. The zero offset stability and the noise level measurements
were conducted in a 4-layer magnetic shield. The zero offset short-term drift lies within 40 pT during 7 hours. The sensor noise10
level estimations are given in Fig. 1, where for comparison purposes noise spectral density of the 1-second INTERMAGNET
magnetometer LEMI-025 as well as typical geomagnetic spectrum are presented. The achieved noise level (1 pT Hz1at
1 Hz) is three times less than that of LEMI-025, what could provide better signal-to-noise ratio especially at measurements of
short-period geomagnetic variations. The zero offset variations, also measured in the magnetic chamber, do not exceed ±1 nT
during sensor temperature excursions in the range +5C to +35C, what is comparable with the best permalloy sensors. Due to15
the excellent noise level, low zero offset short-term drift and good temperature stability, this sensor is very promising for using
in 1-second INTERMAGNET variometers.
Besides the sensor performance, it is very important to create the high stability compensation field, which is canceling the main
Earth magnetic field inside the magnetic cores. The possibility of constructing a digitally controlled current source (DCCS)
with temperature and noise characteristics consistent with the parameters of the best modern fluxgate sensors are considered in20
the next section.
3 Development of digitally controlled current source
For the postulated goal achievement, following noise characteristics of the compensation field were posed: noise level no more
than 0.5; 1.5 and 5 pT Hz0.5at frequencies of 1; 0.1 and 0.01 Hz, respectively, what constitutes 7.1; 22 and 71 109Hz0.5in
relative units taking account the compensation range of ±70000 nT. Instability of compensation field should not exceed 1 nT25
or 14 ppm in relative units in the temperature range -40C. . . +60C. At the condition of a linear dependence of compensation
field on the temperature, the thermal drift should be as small as 0.14 ppm C1. The digitally controlled current source,
which consists of a voltage reference (VR), a digital-to-analog converter (DAC) and current-to-voltage converter, is analyzed.
Analysis of the literature reveals that the problem of creating a stable electric current is associated with relatively high noise
level of semiconductor voltage sources. Ciofi et al. (1997) showed that radical noise reduction can be achieved using chemical30
current source; in the paper (Scandurra et al., 2014) reference voltage noise reduction was achieved using a low-pass filter that
is based on supercapacitors. The current sources’ noise characteristics achieved in (Ciofi et al., 1997; Scandurra et al., 2014)
meet the requirements to the DCCS (Table 1). Table 1 also shows that the noise of the current source (Costa et al., 2012),
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which is built using semiconductor voltage reference, several times exceed specified limits. Devices that developed by Ciofi
et al. (1997); Scandurra et al. (2014), are designed to work in the laboratory in a relatively narrow temperature range. The
stability of these current sources under the temperature, time and other factors of influence is not given, but we can assume
that, for example, voltage of galvanic elements and supercapacitor parameters could significantly depend on the temperature
and mechanical stress.5
3.1 Voltage reference selection
Basing on the detailed review of the characteristics of semiconductor integrated voltage references (Harrison, 2009) it was
found that only very few models have their own noise level, temperature and time drift acceptable to be used in high-class
FGM. The low-frequency (1 Hz) noise level characteristics of different voltage reference models are given in Table 2, which
is mainly filled with data taken from (Fleddermann et al., 2009; Halloin et al., 2014). An indisputable leader within all specified10
parameters is the buried-zener voltage reference LTZ1000 (Harrison, 2009, p. 494) based on the subsurface Zener diode, which
positive temperature coefficient is compensated by the negative coefficient of the forward-biased base-emitter voltage of the
transistor located at the same substrate. This voltage reference has also fairly weak dependence of the output voltage on the
dose of radiation (Rax et al., 1997), which may be important for space application. Achieving a record low temperature drift
(0.05 ppm C1) is due to crystal controlled heating and maintaining its operation temperature in a very narrow range. Taking15
into account the significant power consumption, this way is not always acceptable in FGM and may be unreasonable due to
thermal instability of other units of the VR.
Experimental research of three samples of VR showed significant nonlinearity of the temperature dependence of the output
voltage UREF in the temperature compensation mode without temperature stabilization (Fig. 2), especially at the edges of
the temperature range. According to Tsividis (1980) and computer modeling of the VR circuit (Fig. 3, rb= 0 Ohm, R2 =20
18.6Ohm, R3 = 120 Ohm, R4 = 68.1kOhm, R5is absent), conducted in LTSpiceIV simulation package, expected non-
linearity of the temperature dependence of Ube(VT1) and, accordingly, UREF looks like a dotted curve in Fig. 2, what does not
match with the measurement results of output voltage of LTZ1000 samples. Much better matching of the model curve (solid
line in Fig. 2 ) with experiment results was obtained by adding the resistor rb= 15 kOhm (Fig. 3) to the LTZ1000 circuit given
in the technical documentation (Linear Technology Corp., 2015).25
The UREF voltage increase at low temperatures is perhaps due to a decrease of current transfer coefficient of the transistor
VT1 and a corresponding increase in base current and voltage drop at rb,R2and dynamic resistance of the Zener diode
VZ1, since a collector current depends slightly on the temperature (0.03 %/C). To compensate the effect of the temperature
dependence of the base current, and slightly linearize temperature dependence of Ube(VT1) is possible if to generate in the VT1
collector a current ICproportional to absolute temperature (PTAT) of the crystal. This method is widely used for temperature30
compensation in bandgap voltage references (Harrison, 2009, sect. 14.1). In contrast to bandgap circuits we propose to use a
simpler schematics at the expense of less accurate proportionality of the current to absolute temperature. Approximately PTAT
current IC(VT1) could be generated in the diagram in Fig. 3 at a proper value of the resistor R5. The result of the simulation
of the modified circuit (Fig. 3 and rb= 15 kOhm, R2 = 14.5Ohm, R3 = 120 Ohm, R4 = 30.1kOhm, R5=6.2kOhm) is
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represented by dashed line in Fig. 2. Experimental study of the LTZ1000 sample #3, operating according to the modified circuit,
showed that the output voltage UREF temperature dependence becomes more linear in the temperature range from 25C to
65C (Fig. 4). Further it is planned to get temperature characteristics of the modified circuit in a wider temperature range, and
specify parameters of the LTZ1000 to obtain more reliable results of computer simulation.
3.2 DAC configuration - bipolar vs. unipolar reference input5
As a digital-to-analog converter, one of the best models – 20 bit DAC AD5791 with temperature drift 0.05 ppm C1– was
selected. As the data sheet of this component (Analog Devices, Inc., 2013) contains incomplete information regarding the
noise level at low frequencies, these characteristics were examined in two circuit configurations: with bipolar (Fig. 5, Circuit 1)
and unipolar (Fig. 5, Circuit 2) voltage reference input. The noise tests were carried out using the two samples of the DCCS:
the sample #1 with AD5791 configured for the bipolar reference input; the sample #2 – for unipolar reference input. The10
voltage references were built using LTZ1000. In case of the bipolar input DAC configuration (Fig. 5, Circuit 1) the negative
reference output UREF N was formed from the LTZ1000 positive output voltage UREF P by a voltage inverter based on a
low-noise operational amplifier AD8675 (Analog Devices, Inc., 2012) and a matched pair of the metal foil resistors DSMZ
(Vishay Precision Group, Inc., 2015). The noise levels of both unipolar and bipolar voltage references were estimated before
the DAC tests. It was found that the voltage inverter contributed practically no additional noise, so for both voltage references15
the noise level mainly depends on the LTZ1000 characteristics and are equal to 9·109Hz0.5at 1 Hz; 22 ·109Hz0.5at
0.1 Hz and 70 ·109Hz0.5at 0.01 Hz. In each case, the DAC output was connected to the voltage-to-current converter with
ungrounded load consisting of a low-noise zero-drift operational amplifier OPA2188 (Texas Instruments Inc., 2016) and high
stability metal foil resistors VSMP0805 (Vishay Precision Group, Inc., 2016). The noise level of these identical voltage-to-
current converters was checked separately and it did not exceed 4·109Hz0.5in the frequency band 0.01. . . 1 Hz. For each20
sample, Table 3 shows the noise level of output currents at the various points in the range: at zero ("Iout = 0" column), at the
minimum ("Iout =Imin" column) and at the maximum values ("Iout =Imax " column). The values of the noise level, which
exceed given limits, are marked in bold. Overpassing the requirements at a frequency of 1 Hz for both cases is caused by the
voltage reference noise. The noise of the AD5791 configured with an unipolar reference input (Fig. 5, Circuit 2) is considerably
bigger than that of the version with a bipolar reference input and exceeds the requirements at zero and minimum values of the25
output current. Probably, this is due to the additional 1/fnoise generated by the resistors R1,RFB (see Fig. 5, Circuit 2) when
a larger current is flowing through them. At the bipolar input configuration (Fig. 5, Circuit 1) these resistors are excluded from
the signal pass and can not contribute an excessive noise. So, for better noise characteristics achievement, the AD5791 has to
be connected in the version of the bipolar voltage reference input.
3.3 The temperature stability of the DCCS30
The two prototypes of the DCCS with AD5791 configured for a bipolar reference input were intensively tested in order to
estimate the temperature dependences of both the separate units and the device in whole. The measurements were carried
out at the five values of the DAC input code: 0x100000 (UDAC =URE F N ), 0x13FFFF (UDAC = 0.5UREF N ), 0x17FFFF
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(UDAC = 0 V), 0x1BFFFF (UDAC = 0.5URE F P ) and 0x1FFFFF (UD AC =UREF P ).
Let the DAC output voltage be given by expression
UDAC =UREF P SDAC +UREF N (1 SD AC ),(1)
where SDAC = 0 ...1– transformation factor of the DAC.
It could be shown, that the relative deviation δSinv of the voltage inverter scale factor Sinv leads to inconsistency of the positive5
UREF P and negative UREF N reference voltages and causes the relative change of the DAC output voltage as follows:
δUDAC =UD AC
UREF P
=(1 SDAC )δSinv (2)
From other side the DAC output voltage could change due to the drift of the reference voltage UREF P and imperfection
of the internal components of the AD5791. In order to estimate full-scale and zero-scale error temperature coefficients of
the AD5791 we measured simultaneously UREF P ,(UREF P +UREF N )and UD AC during the temperature tests and applied10
necessary corrections during post-processing. For instance, we estimated AD5791 zero-scale error temperature coefficient
measuring (UREF P +URE F N )and UDAC (0) at at the nominal zero DAC output (input code 0x17FFFF, SDAC = 0.5). Then
we calculated
δUDAC =UD AC
UREF P
,(3)
15
δSinv =(URE F P +UR EF N )
UREF P
(4)
and analyzing sum (δUDAC (0) + 0.5δSinv)found zero-scale error temperature coefficient of the AD5791. The results of such
measurements for the DCCS #1 are given in Fig. 6. The total temperature drift of the DAC output voltage δUDAC (0) (markers
"" in Fig. 6) is mainly caused by instability of the voltage reference inverter (markers "" in Fig. 6) and contribution of the
AD5791 zero-scale error temperature drift (markers "" in Fig. 6) is negligible. For both samples of the DCCS the AD579120
zero-scale error temperature coefficient did not exceed 0.03 ppmC1at the worst case, what is approximately compatible
with data sheet specifications (Analog Devices, Inc., 2013). In both prototypes the drift of the voltage inverters’ scale factors
Sinv linearly depends on temperature with coefficients 0.27 ppmC1in the prototype #1 and 0.45 ppmC1in the prototype
#2. These values coincide well with 0.5 ppmC1maximum temperature coefficient of the resistors ratio (Vishay Precision
Group, Inc., 2015). The full-scale temperature coefficients of the both AD5791 did not exceed 15 ppm in the temperature range25
from -40C to +60C.
The voltage-to-current converter transforms the DAC output voltage UDAC =7.2···+ 7.2V into the output current Iout =
3.6...+ 3.6mA. As expected, the main contribution to the temperature drift of the transformation factor Sui comes from
the resistor Rui, which was combined from the three surface mount resistors VSMP0805 680 Ohm connected in series. The
temperature tests of the voltage-to-current converter in the DCCS #1 are given in Fig. 7. The temperature drift of the converter30
zero-offset (at UDAC = 0 V, markers "" in Fig. 7 ) does not exceed 5 ppm in the temperature range from -60C to +80C
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and is negligible in comparison with the temperature drift of the transformation factor. The similar results were obtained
for the converter in the DCCS #2, but the transformation factor temperature dependence was much more non-linear. The
temperature tests of the resistors, conducted without removing them from the boards, had confirmed that Rui1and Rui2depend
on temperature in different ways (Fig. 8) and temperature characteristics of Rui2is not compatible with data sheet specifications
(Vishay Precision Group, Inc., 2016) . Probably, the reason of the Rui2unexpected temperature dependence was a mechanical5
strain occurred, when the resistors were soldered at the board. The repeat temperature tests of the same resistors re-soldered in
order to minimize possible package deformation reveal considerable change of the resistance temperature characteristic (see
Fig. 8), which become more linear and similar to the data sheet curve.
4 Conclusions
The ways to improve noise level and temperature stability of 1-second fluxgate variometers are considered. The noise param-10
eters of the new fluxgate sensor with Co-based amorphous magnetic alloy are discussed. The achieved sensor noise level is
equal to 1 pT Hz0.5at 1 Hz, what is considerably better than that of the modern observatory fluxgate magnetometers. The
short-term zero offset stability of the sensor is also quite good and lies within 40 pT during 7 hours. The zero offset changes
do not exceed ±1 nT during temperature excursions in the range +5C to +35C. Besides the sensor performance, it is very
important to create the high stability compensation field, which is canceling the main Earth magnetic field inside the magnetic15
cores. The voltage reference in the electronic unit and the compensation windings in the sensor are the most critical elements
in terms of generating the stable compensation signals. The noise level, the temperature drift, the long-term stability of the
best semiconductor voltage references are compared and it is stated that only few of them are suitable for applying in high
class fluxgate variometers. Using the best available electronic components the prototypes of the digitally controlled highly
stable current source is designed and tested. As a part of this work it is experimentally showed, that one of the best voltage20
references LTZ1000 exhibits considerable non-linearity of the temperature dependence in the uncontrolled temperature mode.
The curvature-compensating circuitry is proposed. The source of the extra noise in the digital-to-analog converter AD5791 is
revealed and the appropriate configuration of its inner structure is selected. It is supposed that the application of the discussed in
the paper results and recommendations will allow creating an FGM with outstanding level of noise and temperature parameters.
Acknowledgements. This work was supported by the grant of the National Academy of Science of Ukraine. The author would like to thank25
the Local Organizing Committee of XVIIth IAGA Workshop on Geomagnetic Observatory Instruments, Data Acquisition and Processing
for financial support.
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10-4 10-3 10-2 10-1 100
f [Hz]
10-12
10-11
10-10
10-9
10-8
10-7
10-6
S [T Hz-0.5 ]
geomagnetic variations
FGS32/11 X component
FGS32/11 Y component
FGS32/11 noise fit
LEMI-025
Figure 1. Noise spectra of flux-gate variometers.
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−60 −40 −20 0 20 40 60
T[°C]
−200
−150
−100
−50
0
50
100
150
±UREF [ppm]
IC~const
IC~PTAT
Ube(VT1)
LTZ1000 #1
LTZ1000 #2
LTZ1000 #3
Figure 2. Temperature dependence of LTZ1000 reference voltage – simulated results (lines) and experimental data (marks).
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VT2 VT1
6 8
VZ1
5
7
3
4
+12 V
UREF
R4R2R1
R3
rb
R5
С1
LTZ1000
VD1
Figure 3. LTZ1000 configuration.
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10 20 30 40 50 60 70
T[°C]
−20
−15
−10
−5
0
5
10
±UREF [ppm]
IC(VT1)~const
IC(VT1)~PTAT
Figure 4. LTZ1000 voltage curvature correction.
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Geosci. Instrum. Method. Data Syst. Discuss., doi:10.5194/gi-2017-12, 2017
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VRef
RFB
R1
RFB
INVVout
AD5791
R
R
DAC
Circuit 1
VRef
RFB
R1
RFB
INVVout
AD5791
DAC
Circuit 2
Figure 5. Digital to analog converter configurations.
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Geosci. Instrum. Method. Data Syst. Discuss., doi:10.5194/gi-2017-12, 2017
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−60 −40 −20 0 20 40 60 80
T [°C]
−15
−10
−5
0
5
10
15
20
±UDAC(0);¡0:5±Sinv [ppm]
±UDAC(0)
¡0:5±Sinv
±UDAC(0)+ 0:5±Sinv
Figure 6. Temperature drift of the voltage reference inverter scale factor and the DAC zero offset.
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Geosci. Instrum. Method. Data Syst. Discuss., doi:10.5194/gi-2017-12, 2017
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−60 −40 −20 0 20 40 60 80
T [°C]
−40
−20
0
20
40
60
¢Sui=Sui [ppm]
UDAC =7:2 V
UDAC =3:6 V
UDAC =0 V
UDAC =¡3:6 V
UDAC =¡7:2 V
R¡1
ui1 fit
Figure 7. Temperature drift of the voltage-to-current converter transformation factor.
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−60 −40 −20 0 20 40 60 80
T [°C]
−100
−50
0
50
100
¢R=R [ppm]
Rui1
Rui1 fit
Rui2
Rui2 fit
Rui2 resoldered fit
Figure 8. VSMP0805 resistors temperature dependences.
16
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Table 1. Noise level of some digitally controlled current sources.
Noise level (109Hz0.5)
Current source 0.01 Hz 0.1 Hz 1.0 Hz
Current source A (Ciofi et al., 1997) 7.5 1.4 0.27
Current source B (Scandurra et al., 2014) no data 12 3
Current source C (Costa et al., 2012) no data no data 30
Requirements to the DCCS 71 22 7.1
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Table 2. Noise level of some precision voltage references.
Noise level (109Hz0.5)
Voltage reference 0.01 Hz 0.1 Hz 1.0 Hz
AD587LN (subsurface Zener) (Fleddermann et al., 2009) 250 70 19
LT1021BCN8-5 (subsurface Zener) (Fleddermann et al., 2009) 700 230 70
LT1236ACN8-5 (subsurface Zener) (Fleddermann et al., 2009) 600 130 35
LTZ1000 (subsurface Zener) (Linear Technology Corp., 2015) no data 33 9
LTC6655BHMs8-5 (bandgap) (Halloin et al., 2014) 2000 1000 700
MAX6126 (proprietary) (Fleddermann et al., 2009) 230 100 50
MAX6250ACSA (subsurface Zener) (Fleddermann et al., 2009) 400 150 40
MAX6350 (subsurface Zener) (Halloin et al., 2014) 1000 600 100
VRE305 (subsurface Zener) (Halloin et al., 2014) 4000 2000 400
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Table 3. Noise level of the digitally controlled current source
Frequency (Hz) Noise level (109Hz0.5) Requirements
Iout = 0 Iout =Imin Iout =Imax
circuit 1 circuit 2 circuit 1 circuit 2 circuit 1 circuit 2
1.0 4 8 9 16 9 9 7.1
0.1 7 14 22 39 22 22 22
0.01 15 42 70 150 70 70 71
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ResearchGate has not been able to resolve any citations for this publication.
Article
Full-text available
The degradation of precision reference devices is investigated to determine the relative importance of ionization and displacement damage. The results are compared with theoretical calculations of a basic bandgap reference circuit. Several of the device types were degraded severely at 20 krad(Si), with about the same degradation as that predicted for the basic bandgap reference circuit. One very high precision device with an internal heater performed far better than any of the other devices in the study
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We propose a new approach for the realization of very low noise programmable current sources mainly intended for application in the field of low frequency noise measurements. The design is based on a low noise Junction Field Effect Transistor (JFET) acting as a high impedance current source and programmability is obtained by resorting to a low noise, programmable floating voltage source that allows to set the sourced current at the desired value. The floating voltage source is obtained by exploiting the properties of a standard photovoltaic MOSFET driver. Proper filtering and a control network employing super-capacitors allow to reduce the low frequency output noise to that due to the low noise JFET down to frequencies as low as 100 mHz while allowing, at the same time, to set the desired current by means of a standard DA converter with an accuracy better than 1%. A prototype of the system capable of supplying currents from a few hundreds of μA up to a few mA demonstrates the effectiveness of the approach we propose. When delivering a DC current of about 2 mA, the power spectral density of the current fluctuations at the output is found to be less than 25 pA/√Hz at 100 mHz and less than 6 pA/√Hz for f > 1 Hz, resulting in an RMS noise in the bandwidth from 0.1 to 10 Hz of less than 14 pA.
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This paper presents a Vacquier type fluxgate sensor noise measurement and the constructive principle of our sensor. The sensor contains two cores made by amorphous wires and permits two kinds of excitation modes: excitation by coils and "direct" excitation (excitation current flowing through the cores). In both excitation modes we used the same frequency. We obtained a sensor's noise reduction from 500 pT (pp) to 30 pT (pp) in 0 – 10 Hz frequency range.
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We have fabricated ring-core and single-domain rod-core flux-gate magnetic field sensors with 1/f noise levels at 1 Hz of 1.4 pT/ and 3.5 pT/, respectively. These noise sensitivities were achieved by applying an electrical current through the core of the flux gate to magnetically bias the magnetic rotation of the core perpendicular to the easy-axis direction. We also found that in the rod-core sensor, the spatial correlation lengths of the magnetic fluctuations were 25 and 40 mm with and without the biasing current. The cross-power spectrum magnitude at 1 Hz was less than 200 fT/. © 2001 American Institute of Physics.
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In this letter we determine the theoretical limit of the magnetic-field sensitivity of the flux-gate magnetometer. In order to do so, we have developed a model for the white noise of a flux gate based on the fundamental dynamics of the magnetic material forming the flux-gate core. Solving this model, we predict that the white noise of a physically realizable flux gate with a volume of 2×10−8 m3 is less than 100 fT/. The white noise varies with the lossy susceptibility of the core and inversely with the volume. We also compare the measured white noise of a thin-film flux gate with the predictions of our model and find that the measured and predicted noise agree reasonably well. © 1999 American Institute of Physics.
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We require voltage references with a relative output voltage noise of at most 10<sup>-6</sup>/radicHz in the frequency range of 0.1 mHz-1 Hz. We have investigated the output voltage noise and the external noise sources of a selection of references. Two external noise sources, i.e., supply voltage and temperature, and their coupling to the output voltage of the references were studied. The effect of supply voltage noise was negligible, whereas temperature noise had to be suppressed by stabilization. We found that samples of AD587LN- and MAX6126AASA50-type references fulfilled the requirements if the ambient temperature was stabilized to a level of approximately 50 mK/radicHz.
Current sources & voltage references, Embedded technology series, Newnes, Amsterdam, digitaler nachdr
  • L T Harrison
Harrison, L. T.: Current sources & voltage references, Embedded technology series, Newnes, Amsterdam, digitaler nachdr. edn., 2009.
DSMZ (Z foil) Ultra High Precision Bulk Metal® Z-Foil Surface Mount Voltage Divider, TCR Tracking of <0.1 ppm/ @BULLET C, PCR of ±5 ppm at Rated Power and Stability of ±0.005 % (50 ppm)
  • Vishay Precision Group
  • Inc
Vishay Precision Group, Inc.: DSMZ (Z foil). Ultra High Precision Bulk Metal® Z-Foil Surface Mount Voltage Divider, TCR Tracking of <0.1 ppm/ @BULLET C, PCR of ±5 ppm at Rated Power and Stability of ±0.005 % (50 ppm), http://www.vishaypg.com/docs/63121/dsmz.pdf, 15 2015.
8 nV/ √ Hz Rail-to-Rail Output Op Amp
References Analog Devices, Inc.: 36 V Precision, 2.8 nV/ √ Hz Rail-to-Rail Output Op Amp. Rev. E., http://www.analog.com/media/en/ technical-documentation/data-sheets/AD8675.pdf, 2012.