Recent Advances on Radio-Frequency Design in Cognitive Radio

Abstract

With the growth of mobile data applications, the spectrum allocation is becoming very scarce. To ease congestion and boost speeds, cognitive radio (CR) is currently seen as a major solution and expected to be the key player in the new wireless technologies. In this paper, we will start by introducing the cognitive radio systems, followed by exploring the challenges in designing RF engine, along with an investigation of its antennas, amplifiers, oscillators, and the components that are expected to operate over a wide range of frequencies.

Full text

Review Article
Recent Advances on Radio-Frequency Design in Cognitive Radio
H. M. El Misilmani, M. Y. Abou-Shahine, Y. Nasser, and K. Y. Kabalan
ECE Department, American University of Beirut, P.O. Box 11-0236, Beirut 1107 2020, Lebanon
Correspondence should be addressed to Y. Nasser; yn@aub.edu.lb
Received  October ; Accepted  January 
Academic Editor: Symeon Nikolaou
Copyright ©  H. M. El Misilmani et al. is is an open access article distributed under the Creative Commons Attribution
License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly
cited.
With the growth of mobile data applications, the spectrum allocation is becoming very scarce. To ease congestion and boost speeds,
cognitive radio (CR) is currently seen as a major solution and expected to be the key player in the new wireless technologies. In this
paper, we will start by introducing the cognitive radio systems, followed by exploring the challenges in designing RF engine, along
with an investigation of its antennas, ampliers, oscillators, and the components that are expected to operate over a wide range of
frequencies.
1. Introduction
With the increasing demand for high data rates requiring
high resources, such as energy and frequency bandwidth,
cognitive radio (CR) is thought of as a very promising solu-
tion. e basic operating principle of CR device relies on a
cycle of observation, analysis, and decision and an oppor-
tunistic access to the available bandwidth. Hence, a CR
device, usually referred to as secondary terminal, has to rstly
sense the existence of a primary transmission and oppor-
tunistically transmits whenever a frequency/time slot is
vacant. If the authorized (primary) terminal restarted trans-
mission, the secondary terminal jumps o into a dierent
band or alters its transmission parameters so that it does not
aect the primary transmission [].
In practice, the cognitive radio devices are expected to
sense the occupancy of any channel at any band in the
entire spectrum and autonomously adapt to the primary
transmission []. is continuous (or discontinuous) sensing
process on a large bandwidth imposes dierent constraints
on the radio-frequency front-ends of the secondary terminal.
More precisely, these requirements constrain strict issues
on antenna design, low noise amplication, frequency syn-
thesizers providing a carrier frequency from tens of mega-
hertz to about  GHz, mixing spurs, and spectrum sensing.
Broadband and tunable antennas, multiband ampliers, RF
lters, broadband direct-conversion mixers, baseband lters,
and ADCs/DACs are needed to realize soware-dened cog-
nitive radio equipment. ese RF components are expected
to operate over a wide range of frequencies [].
In cognitive radio, a recongurable radio front-end can be
programmed to transmit, steer to any band, tune to a channel
of any bandwidth, and receive any acceptable modulation
scheme. e ability to design linear and spectrally agile
components and architectures in the radio-frequency front-
end of the transceiver is considered a primary technological
concernincognitiveradioarchitectures.
eobjectiveofthispaperistorevisethemajorcon-
cerns of the front-end design of a CR system including the
antennas, ampliers, and oscillators. In the literature, very
few works have investigated the design issues of the CR front-
end from end-to-end. Moreover, a complied review of the
dierent constraints has been very little tackled. We cite, for
instance, the work in [] as a rst attempt to harmonize all
these constraints. Hence, this paper belongs to the research
workswhichcouldbeusedasareferencefortheRFengineers
working on CR. To complete this work, we detail and
investigate the challenges to overcome in the next few years
in order to design a complete and smoothly tunable RF front-
end for CR applications.
Hindawi Publishing Corporation
International Journal of Antennas and Propagation
Volume 2016, Article ID 9878475, 16 pages
http://dx.doi.org/10.1155/2016/9878475

 International Journal of Antennas and Propagation
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F : A proposed sensing and communicating antenna cong-
uration in [].
2. Antennas for Cognitive Radio
Cognitive radio communication is envisaged to be a new
paradigm of methodologies for enhancing the performance
of radio communication systems through the ecient uti-
lization of radio spectrum. A key enabler for realization
of a cognitive communication system and one of its main
challenges is the capability of recongurability in the under-
lying hardware and the associated protocol suite. From the
antenna design perspective, the demand for multiwideband
antennas which can be easily integrated with the communi-
cation system is continuously increasing. Recongurable and
frequency agile architectures are mostly designed nowadays
in order to solve the broad frequency allocation and to reduce
the number of functional blocks. Intensive work has been
done in designing antennas for cognitive radio applications.
euseofwidebandantennasforspectrumsensingandnar-
rowband antennas for transmission has been proposed by the
research community [, ]. e sensing and transmitting
antennascouldbealsofoundinthestructure,asshownin
Figure .
In general, there are three dierent categories of recong-
urable antenna:
() Frequency recongurable antennas: the frequency of
theantennaistunedtohavesinglemultifunctional
antennaasasmallterminalformanyservices,with
radiation pattern remaining unchanged while the
frequency is changing [–].
() Radiation patterns recongurable antennas: the
antenna can steer its radiation patterns beams to dif-
ferent direction. e frequency remains unchanged
while the radiation pattern changes upon the system
requirements [–].
() Polarization recongurable antennas: providing an
additional degree of freedom to improve link quality
as a form of switched antenna diversity with improved
signal reception performance in a multipath fading
environment [].
So far, recongurable antennas for cognitive radio com-
munications can be classied into three types: electronically
recongurable antennas, mechanically recongurable anten-
nas, and optically recongurable antennas.
T : Operating bands achieved when switches are ON/OFF [].
Switch  Switch  Switch  Switch  Frequency band (GHz)
ON ON ON ON .–. and .–.
ON ON ON OFF .–. and .–.
ON ON OFF ON .–.
ON ON OFF OFF .–.
ON OFF ON ON .–.
OFF ON ON ON .–.
OFF OFF ON ON .–.
71
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Switch 4
Switch 1
Switch 2
Switch 3
F : A proposed sensing and communicating antenna cong-
uration in [] (dimension is mm).
2.1. Electronically Recongurable Antennas. Electronic recon-
gurability is usually achieved by incorporating switches,
variable capacitors, lumped components such as PIN diodes,
varactor diodes, MEMS switches, or phase shiers in the
topology of the antenna [, , –]. An example of a
frequency recongurable antenna for cognitive radio is pro-
posed in []. e sensing and communicating antenna are
positioned at the same one volume, with a printed hour glass
shaped coplanar waveguide (CPW) fed monopole sensing
antenna operating at . GHz to  GHz, accomplishing the
UWB characteristic. e reverse side of the substrate con-
tains the communicating antenna designed to operate from
. GHz to . GHz. Another broadband antenna designed
for a frequency and radiation patterns recongurability for
cognitive radio is investigated in [] and shown in Figure .
e antenna consists of two branches. e le branch involves
a short transmission line and a short dipole, and the right
branch involves a long transmission line and a long dipole.
By setting switching pairs of the four controlling switches,
mutual coupling between a dipole and a nearby transmission
is disturbed. e currents on the transmission line will be
unbalanced, and hence radiation will occur. An example of a
summarizing table of such recongurability of the proposed
antennain[]isshowninTable.
In [] PIN diodes are used in the design of a recong-
urable C-slot microstrip patch antenna that can operate in
dual-bandorinverywidebandmode.Adual-bandrecon-
gurable double C-slot microstrip patch antenna is also pro-
posed in [] and shown in Figure . Two PIN diode switches
are used here to generate a dual-band and wideband by
changing the states of the PIN diode. e antenna can operate

International Journal of Antennas and Propagation 
PIN diodes
Capacitors
F : A proposed sensing and communicating antenna cong-
uration in [].
indual-bandorinverywidebandmodein,,andGHz
bands. e wideband mode can be obtained when both
switchesareintheONstate.
In [] a quad-band antenna for cognitive radio is pre-
sented. It has a direction radiation pattern in four frequency
bands, covering most of the spectrum used for existing
wireless applications. MEMS switch is used to adjust the
operating frequency of the quad-band antenna. A two-
element antenna array was further developed to increase the
antenna gain for base station applications.
A dual port sensing and communicating antenna design
forcognitiveradiosystemisalsopresentedin[].Atunable
narrowband frequency operation is also proposed in [].
AnSMVvaractorisusedtoadjustthelengthofan
open loop resonator (OLR) based band-stop lter. GaAs eld-
eect-transistor(FET)isusedin[]todesignanUWB
microstrip monopole antenna with recongurable multiband
function. In [], a novel design of a recongurable minia-
turized planar spiral monopole antenna suitable for TVWS
applications is reported. By inserting a tunable inductor on
the spiral monopole and modifying its inductance, frequency
recongurability is attained.
2.2. Mechanically Recongurable Antennas. Mechanic recon-
gurability is usually achieved by incorporating some phys-
ical alteration of the antenna structure using a rotational
movement. e advantage of this method is that no biasing
circuits for switch activation are needed, which might aect
the antenna performance. An example of this implementa-
tionisshownin[].eantennapresentedconsistsoftwo
structures incorporated together into the same substrate: the
rst structure is an ultrawideband (UWB) antenna covering
the spectrum from . GHz for channel sensing, and the
F : Implemented front-end using rotatable controlled recon-
gurable antennas [].
second structure is a frequency recongurable triangular-
shaped patch for establishing communication with another
RF device. e frequency recongurability is achieved via a
rotational motion, for which the rotating part of the antenna
is responsible to produce the required frequency tuning. By
rotation of the antenna with dierent rotation angles dierent
resonances are produced, making the antenna suitable to
communicate at the frequency specied by the “sensing”
antenna.
Another example of this implementation in the design
of a cognitive radio front-end is presented in [] using also
rotatable controlled recongurable antennas. e frequency
agility is also achieved via a rotational motion of the antenna
patch controlled by a stepper motor mounted on the back
oftheantennastructure.isisshowninFigure.In[],
an UWB sensing antenna is presented with slotted polygon
shapedpatchwithpartialgroundonthereversesideofthe
patch. e frequency conguration is achieved by rotational
movement of the triangular-shaped patch communicating
antenna.
Moreover, in [], a frequency recongurability of an
antenna for cognitive radio is also achieved via rotation
motion of a part of the antenna patch. e rotating part has
theformofacircleandcontainsfourdierentshapesfor
which each shape corresponds to a dierent antenna struc-
ture. With every rotation, a dierent antenna structure is fed
in order to produce a dierent set of resonant frequencies.
Using the same rotation mechanism, a multiband recong-
urable antenna is presented in [] using dierent rotating
slot congurations of the antenna.
2.3. Optically Recongurable Antennas. Additional work has
been done on the design of optically recongurable antennas
[, –]. An example of a photoconductive switch that
uses an 𝜂-type silicon switch doped with phosphorus to
increase its conductivity is proposed in []. A frequency
andradiationpatternrecongurabilityhasbeenachieved
by eectively changing the dipole arm length. Optical-ber
cables are used to feed the printed dipole antenna. Another
example of optically controlled frequency recongurable

 International Journal of Antennas and Propagation
Antenna
substrate
Copper piece for
thermal radiation and
laser diode xture
Laser diode
F : Integration of the laser diode into the antenna structure
[].
microstrip antenna is also presented in []. Moreover, eld-
eect-transistor- (FET-) based electronic switches used with
optical control are proposed in [] to control planar arrays
of electrically small metallic patches.
Furthermore, dynamically changing the material proper-
ties of parts of an antenna can signicantly alter the antenna
performance.isispossiblethroughchangingtheconduc-
tive properties of some materials through an electric current
or optical signal. In [] a frequency recongurable rectangu-
lar patch antenna is achieved using LED that is used to alter
the conductivity of the semiconductors used. e variable
conducting material changes the resonant frequency of the
patchbyincreasingordecreasingthesizeofthepatch[].
Another work that reduces the complexity of the systems
negating the need for optical-ber cables [] is proposed in
[]. e design is based on integrating laser diodes within the
antenna structure, as shown in Figure . is technique does
not require any biasing lines for switch activation purposes
in the antenna radiating plane, as is the case with RF MEMS
[] or PIN diodes [].
3. Amplifier Design for Cognitive Radio
3.1. Power Ampliers. Ampliers are basic building blocks
in electronics communications systems. For the transmit-
ter design in cognitive radio applications, in order not to
interfere with the primary user and operate at multistandard
frequency range, a high power gain is required, and thus a
broadband linear power amplier (PA) is desired. It must
provide a high output power, simultaneously assuring a wide
bandwidth, high eciency, and linearity behavior.
e power amplier, which is a crucial element of wireless
transmitters, consumes a large portion of energy in RF
circuits during transmission. Consequently, an ecient PA
design with high eciency capabilities is required. As the
system requirements vary, the specic constraints on the
amplier design also vary considerably. ere are, however,
common requirements for nearly all ampliers, including
frequency range, gain atness, output power, linearity, match-
ing, and stability. Oen there are design trade-os required
to optimize any parameter over the other, and performance
compromises are usually necessary. Dierent classes and
modes of operation were dened, each achieving certain
criteria in such performance metrics. Popular examples are
the basic classes such as Classes A, B, C, D, E, and F PAs.
Because of their highly versatile circuit function, PAs have
always been the rst to benet from developments in the
device and semiconductor technologies, which helped in
dening even new techniques for operation like the Doherty
ampliers and Class J PAs to meet the requirements imposed
on PAs due to the evolution of new communication systems
and standards as CR systems.
When the PA is driven towards saturation, the nonlinear
distortion will increase signicantly. On the other hand, the
highest PA power eciency is obtained at the saturation
point. In fact, there is a trade-o between the power eciency
of the PA and its linearity. e nonlinear behavior of the PA
leads to spectral regrowth of its out-of-band output signal
and, as a result, to adjacent channel interference (ACI). e
ACI power is a nonlinear increasing function of the PA
input power. erefore, it is important to consider the PA
nonlinear behavior of the CR transmitter and the resulting
ACI power for allocating power in CR networks by noting the
interference temperature limits. e CR system design should
be aware of the nonlinear behavioral model of the PA and of
the other users constraints in the environment.
ere are several methods to develop wideband PAs:
(i) e traveling-wave and distributed amplier topolo-
gies have excellent characteristics in terms of band-
width, gain atness, and input voltage standing-wave
ratio, so that they are widely used for broadband PA
development [, ].
(ii) e multistage LC combination technique is pro-
posed [] to improve the GaAs HBT PA in order to
be used in broadband wireless applications from .
to . GHz.
(iii) e push-pull PAs can reach wide bandwidth using
broadband transformers [–].
(iv) e shunt-feedback technique and multisection dis-
tributed matching networks are demonstrated to be
useful in [].
Several works have been proposed in designing power
ampliers for cognitive radio applications. In [], a broad-
band PA is demonstrated in InGaAs HEMT technology using
load impedance tracking. e power-added eciency and
the  dB compression point of the PA are better than %
and . dBm, respectively. A wideband power amplier for
the application of intelligent cognitive radios is proposed in
[]. In this PA design, the broadband frequency response is
enhanced using transformer matching network and resistive
feedback. As shown in Figure , series stack topology is used
to achieve the broadband load impedance match by dis-
cussing the constraints of stack PA in both GaAs and CMOS
methods. e main dierence between these two PAs is the
feed point. In the rst one, transformers are employed using
RFinputatthebottomandgroundatthetop.InFigure(b),
the RF feed point is upside-down compared to Figure (a).
DuetothefactthatthesetwoPAsarevoltagecombined,
the total biasing current is controlled by the transistor. To
verify the design models, a high eciency broadband PA in
commercial . m CMOS process with the best PAE of %
and the  dB compression point of  dBm is established. is

International Journal of Antennas and Propagation 
RFout
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F : Two kinds of transformer orientations of stack PA. In (a), the transformers are opposite coupled. (b) e upside-down input fed
conguration [].
PA also demonstrates the widest bandwidth performance
among CMOS PAs below GHz. In [], a high eciency
Doherty power amplier suitable for TV band applications
isproposed.ItisdesignedfollowingtheClassABscheme
for the main amplier and a Class C scheme for the peak
one. It attained a high power-added-eciency of .%,
a . dBm output power, an associated gain of .dB,
and an operating frequency bandwidth between  and
 MHz (.% fractional bandwidth) which make it
suitable for cognitive radio applications in the TV band.
In [], the interference control in time-windowed OFDM
Systems with realistic power ampliers for cognitive radio
applications is investigated. e load-tracking technique
basedonthefrequency-variedloadpullisproposedforthe
PA design. e interference in holes generated inside an
orthogonal frequency division multiplexing (OFDM) spec-
trum, for applications related to cognitive radio, is discussed.
Due to its simplicity, the time-windowing technique is
selected in order to obtain interference reduction in the out-
band region and inside the spectral hole. In this design,
it is shown how to choose system parameters (number of
guard subcarriers, window type, and extended guard interval
duration) in the presence of a nonlinear power amplier,
whose generated interference is of dominant importance in
assessing the real gain provided by the OFDM technique.
3.1.1. Tunability for Power Ampliers. Broadband PA modules
with frequency tuning capabilities have been also investi-
gated. ese modules having multiple narrowband power
amplier chips and corresponding matching circuits are
joined into a platform and controlled by switches in order to
obtain frequency band tuning [, ].
In order to lessen the number of components in PAs and
corresponding matching circuits, alternative single chip solu-
tions have been recommended. For example, while a balanced
amplier is a good applicant, it is usually not practical to
use quarter wavelength size of the couplers. A distributed
amplier is another consideration, where it suers from low
eciency and low gain and requires a relatively large chip
size. On the contrary, the feedback ampliers have a relatively
small chip size, but with low gain at microwave frequencies
and compromised eciency when resistive feedback is used.
Asshownin[,],lossymatchingnetworksareused
to attain better gain atness, making a trade-o with the
power gain. In [], a novel recongurable output matching is
applied in the dual mode broadband InGaP HBT PA through
employing PIN diodes to adjust the LC networks as shown
in Figure . Using the compensating matching technique,
the -stage broadband power amplier is achieved and the
distribution of power gain among stages is optimized. e
output matching circuit is realized with parallel LC tank
circuits using PIN diodes in order to control the inductor
value. is broadband amplier module oers the advantage
of using less components, less power insertion loss, high
linearity, and small size.
3.2. Low Noise Amplier. For the receiver side in cognitive
radio,alownoiseamplierisrequired.IntheCRapplica-
tions, it is very essential to detect the presence of primary
users, creating spectrum opportunities for secondary users
(CR users) in order to increase the capacity of wireless
communications. For realizing this function, the sensitivity
of the spectral sensing in CR must be much higher than con-
ventional radio receivers. High sensitivity integrated receivers

 International Journal of Antennas and Propagation
Broadband
power amplier
Output
matching circuit
CT1 CT2
Input port Output port
(a)
+
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C4
C3
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(b)
F : (a) Proposed broadband power amplier module; (b) schematic of the novel recongurable output matching circuit [].
Spectrum access engine (management, access, allocation, etc.)
Sensor and control plane
Reconguration
and radio resource manager
RF front-
end, PA,
lters,
mixers
Multiple
receivers
Rx
Multiple
transmitters
Tx
ADC/DAC Digital base band
processing
Cross-
layer design
and
optimization
Upper
layers
mechanisms
and
applications
Data
Control
(soware, core +memory)
F : e spectrum access scheme including RF front-end.
require low noise amplier (LNA) that can deliver suciently
low noise gure (NF), acceptable linearity, quite high gain,
andlowpowerdissipation.eserequirementsmustbe
achieved over a wide frequency range in CR. Commonly
wideband LNAs consist of MOSFETs and resistors which are
far from the above-mentioned requirements. Some research
intends to enhance the noise performance in LNA design
using noise cancellation methods. However, the gain of all
those LNAs is less than  dB when the NF is about  dB, and
all of them have relative large power dissipation [–].
Several works have been proposed in designing low noise
ampliers for cognitive radio applications. A new wideband
low noise amplier, operating in UHF band for spectral
sensing in the receiver of cognitive radios, is reported in [].
e noise canceling technique is employed in this circuit,
and it yields higher gain and lower power dissipation. e
method of shunt-resistive feedback is adopted for achieving
a broad bandwidth. e noise gure is less than  dB, the
gain is – dB, and the LNA consumes only .mW at
. V power supply. In [], a new full on-chip CMOS LNA
topology,workingintherangeofMHztoGHzwith
very low power consumption, is introduced. In this LNA,
the common-gate (CG) stage for wideband input matching is
combined with the common-source (CS) stage for canceling
the noise and distortion of CG stage. Moreover, the CS stage
used both nMOS and pMOS transistors to improve the IIP.
is LNA achieves an input return loss (S) less than − dB
over the whole bandwidth and a noise gure of . dB to
. dB while consuming only  mW from a  V power supply.
e average power gain (S) is  dB. e achieved IIP and
IIP are about − dBm and  dBm, respectively.
4. Oscillator Design for Cognitive Radio
4.1. Introduction. Figures  and  present the main blocks
of a suitable architecture for CR systems. In practice, the
spectrum sensing architecture is composed of two stages.
In the rst stage, the target spectrum undergoes a coarse
detection and analysis on relatively large subbands of interest.
is allows recognizing those channels occupied by primary
(strong power) signals and marking them as busy. en, a
sweep over the band of interest is applied to have a ne
detection of the presence or absence of the primary signal.
e second stage consists of the RF transmission block able
to generate the necessary RF signals whenever a spectrum
opportunity occurs.
e signal band in CRs is shared between primary
and secondary transmission. In practice, there are various
architectures of receivers, such as direct-conversion (homo-
dyne), super-heterodyne, low-IF, and bandpass sampling
radio architecture. However, contrarily to the conventional
tuners, the spectrum sensing receivers should have similar

International Journal of Antennas and Propagation 
Main LO
Tx front-end
Rx front-end
Rx front-end
Wideband
LO coarse
tuning
ADC
ADC
DSP
DSP
FFT
ADC
ADC
DAC
DAC
Spectrum mapping
Main radio
Spectrum monitor
F : Spectrum monitoring and mapping scheme [].
architectures to spectrum analyzer. e frequency of the
local oscillator (LO) and its intermediate frequency should
be selected outside the signal band boundary []. On the
other hand, as the CR system should call for exible RF
architectures, with a wide frequency range, a wideband
high performance controlled local oscillator with a relatively
relaxed phase noise performance should be designed for
the use of both spectrum monitoring and transmission
functionalities.
In this section, we investigate techniques related to the
design and implementation of VCOs and to the increase of
their tuning range while maintaining acceptable design speci-
cations such as phase noise, power dissipation, accuracy, and
stability. In practice, two popular oscillator architectures have
been proposed for spectrum sensing applications: LC tank
based and ring based oscillators. e LC tank oscillators are
well-known for their phase noise and low power consump-
tion at radio frequencies and they are usually recommended
in many applications. However, they suer from low tuning
range as compared to ring oscillators, a necessary step in
spectrum sensing techniques. In practice, the tuning range of
the LC tank based oscillator is restricted to only % to %
if implemented without compensation techniques yielding
then to vulnerability to process variations. e utilization
of an inductor with a high quality factor however increases
the chip area, the cost, and complexity of the oscillator [].
Moreover, in a system requiring multiple output phases,
additional circuits, such as I/Q generators, make the system
more complex.
On the other hand, ring oscillators are considered to
be suitable candidates as the core block in the required
frequency synthesizer naturally with wide tuning range.
Moreover, the relatively small area makes it a good choice
for low cost portable devices. ey are able to generate
wide range frequency signals with multiple output phases
without any supplementary circuits which is very practical to
cognitive radio applications. However, they suer from high
level of phase noise.
4.2. e LC Tank Oscillator. In the literature, dierent LC
resonant tanks with multiple frequencies and based on
transformers or multitapped inductors have been proposed
to design tunable multibands and/or wideband oscillators
[–]. In [–] transformers with small turn ratio and
adequate coil coupling are utilized to design dierential
dual-band oscillators. As the target frequency ranges from
. GHz to  GHz, two congurations are proposed. A one-
port oscillator is used for the low-band mode, while a two-
port oscillator design is used for the high band.
Multitapped inductors are proposed in [] to design
an LC tank of order equal to , covering the two bands
. GHz and . GHz with target application GSM/DCS/PCS
standards. In [], transformers with high coupling ratio and
largeturnratioareusedtodesignadual-bandoscillator
( GHz/ GHz). To stabilize their one-port oscillator, a
notch-peak cancellation technique is proposed. Similarly, a
single multitapped inductor and two-port oscillations are
used in [] to design a . GHz/ GHz dual-band dif-
ferential VCO for area-ecient wideband applications. In
[],atriple-modewidebandVCOtunablefrom.GHz
to . GHz is proposed by utilizing loosely coupled -coil
transformer with one-port oscillations.
All this work shows that VCOs with high order LC tanks
present large prospective to design wideband or multiband
applications. Nevertheless, the comparison of these VCOs
with the conventional LC-VCO schemes reveals that the
stability of the VCO becomes a real issue when the order of
the resonator increases. is leads to an increased complexity

 International Journal of Antennas and Propagation
which yields to the inclusion of many design parameters
in the design of the VCO end-product. For instance, the
inductor, capacitor, and the coupling between the inductors
ratios should be carefully (and maybe jointly) considered in
order to optimize the VCO outputs.
Fortunately, some attempts in the literature have been
established [] to derive analytical results and close-form
expressions of the oscillation frequencies and of the condi-
tions of the transformer-based one-port and two-port oscil-
lators.ishasled,fromoneside,tosuccessfulrealizations
of one-port and two-port oscillators with high order resonant
tanks and to a comprehensive evaluation and comparison of
dierent tanks topologies and designs at dierent bands for
particular applications on the other side. Nevertheless, it is
considered in [] that the capacitors of the oscillators are
lossless, a basic assumption yet invalid in real networks. To
solve this problem, a complete understanding of the two-
oscillator congurations (one-port and two-port) including
capacitance loss, phase noise, and tank Qs is provided in [].
Itisshownthatone-portoscillatorsconsumelesspowerbut
need to be stabilized if the designer is seeking oscillation at
thehigherpeakfrequency.etwo-portoscillatorshaveno
stability issue and have superior phase noise performance
for a given output swing but are less ecient in convert-
ing the bias current to the tank swing. Based on these
observations, both congurations are combined to exploit
their corresponding advantages at dierent frequency for
spectrum sensing (and soware-dened radio-SDR) appli-
cations. Accordingly, a transformer-based dual-band Q-VCO
frequency synthesizer is proposed and designed [], which
supports existing wireless technologies.
In [], a VCO with tuning range extension circuit is
designed using an injection-locked frequency divider (ILFD)
and ip op dividers. e -stage dierential ILFD can
generate quadrature outputs, with a tunable divide ratio equal
to , , , and  and very wide output frequency range.
e proposed oscillator achieves a large frequency from
. MHz to . GHz with  dBc/Hz of gure of merit. It is
implemented using a  nm CMOS process. Other wideband
oscillatorshavebeenalsoproposedintheliteratureusing
the same tuning range extension technique and QVCO. We
cite, for instance, the works of [, ] which design VCOs
for multistandard transceivers using a tuning range extension
technique and QVCO. ese VCOs achieve quite wide tuning
range and high spurious rejection using single-sideband
mixer (SSBM) with I/Q signals. However this is obtained at
the detriment of large phase noise and power consumption.
e work of [] is also interesting in this regard. A tuning
range extension technique is utilized using dierential VCO,
a mixer, and dividers. However, the VCO does not generate
I/Q signals, while spurious frequencies appear at the output
of the mixer.
4.3. e Ring Oscillator. Generally, a ring oscillator is com-
posedofacascadeofdierentstages.Itsoscillatingfrequency
is proportional to the number of stages (𝑁)andthetrans-
mission delay of each delay cell stage (TD). In practice, the
number of stages is usually xed for a given design while the
TD reects the charging and discharging time for the load
capacitance. For instance, with a CMOS inverter based delay
cell used in an oscillator, the transmission delay could be
tuned by changing the bias current of the design []. For a
dierential amplier base stage, either the tail current source
orPMOSloadcouldbevariedfortuningpurposes[].ese
design parameters allow to oer large VCO gain variation
over the overall tuning band. In particular, very high gain is
oen observed in the middle of the tuning range, where the
oscillator would suer from high sensitivity to control voltage
imperfections.
ere is no doubt that a linear relationship between
the voltage control and the oscillating frequency is required
for wideband systems to ensure the locking range of the
frequency synthesizer. Hence, most of the oscillators designs
were based on this basic yet important approach. For CR
applications using ring oscillator approach, we distinguish
the work of [] on standard  nm CMOS technology
targeting a wide range frequency tuning range. e three-
stage dierential ring oscillator proposed therein is tuned by
an array of MOS varactors, controlled by a staggered voltage
osetsystemforimprovedtuninglinearity.eproposed
ring oscillator oers a measured tuning range from . GHz
to . GHz, consuming . mA current from a . V supply
voltagewhilethephasenoisemeasurementsareappropriate
for such designs (−. Bc/Hz at  MHz oset from an operat-
ing frequency of . GHz). In [], a novel voltage-controlled
ring oscillator fabricated in TSMC .-m CMOS technology
is designed. To maintain the linearity between frequency and
voltage characteristics and phase noise specications over
a wide tuning range, a transmission gate is implemented.
e proposed ring oscillator has achieved a wide operating
frequency range from MHz to  MHz covering particu-
larlyTVchannelbands(orTVWhiteSpacesTVWS).e
interesting point of the proposed ring design resides in its
phase noise which measures  dBc/Hz at  MHz oset from
 MHz and its power consumption of  mW.
Alinearcurrent-controlledoscillator(CCO)isproposed
in[].eoutputfrequencyoftheproposedoscillator
is implemented using a linear CMOS switched-capacitor
frequency detector (SCFD) that operates appropriately over
the frequency range of the designed oscillator. en, a neg-
ative impedance converter (NIC) is utilized to control the
supply current and complete the ring oscillator design. e
interesting part of the proposed oscillator resides in the
fact that it does not use operational ampliers or resistors
which makes it very easy and widely applicable in all types
of oscillators. is oscillator has power consumption less
than . mW in the frequency range of  MHz to . GHz
fromasupplyvoltageof.Vwithveryhighphasenoise
performance (in the order of . dBc/Hz at  MHz oset
from the carrier frequency of . MHz). In [], a compact
and low power quadrature local oscillator is designed and
implemented for the .–GHz band using a dierentially
tuned LC-VCO. e same design is then converted to cover
the band – GHz with continuous frequency coverage. A
-stage dierential injection-locked ring oscillator (ILRO) is
used subsequently to the latch-based divider to generate
quadrature output phases without restricting % duty cycle
from input signals as those of conventional divide-by-

International Journal of Antennas and Propagation 
approaches. When implemented in a  nm general purpose
CMOS IC technology, the integrated quadrature-phased LO
consumes  mA of current at a  V supply and oers very
good phase noise performance across the entire – GHz
band targeting the CR applications.
In [, ], an all-digital phase-locked loop using a com-
bination of a digitally controlled ring oscillator with an LC
tank is proposed to extend the tuning range of the LC tank
and reduce its power dissipation. In the suggested design,
an adaptive frequency calibration, based on binary search, is
introduced to accelerate the frequency settling. e proposed
architecture is fabricated in a -nm CMOS while the fre-
quency synthesizer has an active area of . mm2and achieves
a frequency tuning range of . to . GHz, with power
consumption of less than  mW from a .-V supply.
To improve the poor phase noise performance common
in ring oscillators, a series of works have been proposed to
dealwiththisissue[,–].In[],aringbasedoscillator
using a saturated-type dierential delay cell with a positive
feedback path is proposed. It is shown that the phase noise of
the proposed oscillator has comparable performance as the
conventional LC tank VCO. However, the frequency tuning
mechanism implemented is not practical as it changes the
strength of the latch in the positive feedback. Indeed, the
latter could not be proportional to the control voltage which
means that the oscillator could not have a linear frequency
versus the input voltage. In addition, the oscillator could not
operate in low frequency range such as VHF bands. To over-
come this problem, a transmission gate is proposed in [, ]
to linearize the voltage-frequency tuning relation. In [], a
monolithic ring VCO is proposed. It is based on delay cell and
a -stage scheme taking into account the trade-o between
the frequency tuning range, phase noise performance, and
power consumption of the oscillator. e delay cell consists
of one NMOS input pair, one PMOS positive feedback pair,
and one transmission gate which connects the output of one
delay cell to the input of the next one.
4.4. Other Oscillators. In [], a Distributed Voltage-Con-
trolled Oscillator (DVCO) suitable for CR applications is
proposed and optimized. To do so, Harmonic Balance (HB)
based optimization techniques have been used. e proposed
DVCO aims at minimizing the output power variation along
thefrequencybandanditconsistsofadistributedamplier
with a feedback loop. e oscillator tuning is achieved
through biasing all the drain terminals with a xed voltage
𝑉DD and varying the gate bias voltages of the active devices
in pairs. e measured DVCO frequency band, shown in
Figure , covers from .GHz to .GHz, dissipates a
power equal to  mW, and generates average output power of
.dBmwhiletheaveragemeasuredphasenoiseatMHzo-
set from the carrier across the tuning range is −. dBc/Hz.
Another interesting work in this eld concerns the con-
tributionof[],basedonACMOSspectrumsensoraim-
ing at detecting spectral usage and spectrum holes in the
. GHz industrial scientic-medical (ISM) band. e pro-
posed design consists of a swept oscillator and frequency
discriminator, both of which use the injection locking of
the VCO to process the sensed signal without requiring
0
100
200
300
400
500
600
700
800
900
0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
Frequency (MHz)
Simulated
Measured
V
ctrl (V)
F : Frequency-voltage control characteristics [].
a frequency synthesizer. e proposed sensor can detect
thefrequencyandpoweroftheprimarysignalwithhigh
accuracy at a spectrum scanning speed of MHz/ms. It is
worth mentioning that the sensitivity of the proposed design
can be below − dBm when an external low noise amplier
(LNA)isusedinfrontofthesensorIntegratedCircuit.
5. Challenges
Although a large number of research works on CR exist,
the challenges of this technology remain numerous. Besides
the challenges related to intelligence distribution and imple-
mentation, decision making, sensing algorithms and learn-
ing process, delay/protocol overhead, geolocation, and exi-
ble hardware design, major diculties lie in the implementa-
tion and the design of antennas, ampliers, and oscillators.
e complexity found in the implementation of these RF
parts is lowering the pace of its development []. Several
workshavebeendoneinanalyzingsuchchallenges[,,,
, , –].
Referring to Figure , for this system to operate on
multiband or a broadband simultaneously, the employment
of parallel processing from antennas to analog to digi-
tal interfaces is materialized. Multiantennas are necessary
for MIMO operation and/or multibands operation. Passive
module used for RF ltering, switching, or duplexing and
impedance matching between power ampliers and antennas
are located aer the antennas. Aer that, multireceiver
(Rx) and multitransmitter (Tx) are followed before a multi-
ADC/DACs module. e high performance of this system
requires coding and encoding, in addition to conventional
processing for modulation and demodulation, and conse-
quently digital ltering, digital automatic gain control, dc
oset cancellation, nonlinearities, and correction and calibra-
tion of analog errors. In order to improve the performance of
the analog part, a feedback from baseband to RF front-end
and transceiver combined with control plane and sensor is an

 International Journal of Antennas and Propagation
essential step. e challenges of RF front-end and transceiver
in the short-midterm are
(i) reducing the chip and passive components,
(ii) increasing their frequency-tunability,
(iii) minimizing the dissipation of power,
(iv) reducing their area.
5.1. RF Front-End: Tx and Rx. Investigating the RF front-end,
the RF section needs to be particularly exible. Flexibility
requires the cognitive radio transceiver being able to adapt
to multiple access methods and adaptive modulation scheme
senseanduseanyavailablefrequencyband,establishcom-
munication with several points, switch quickly between links,
and most importantly handle very large peak-to-average
power ratios (PAPRs) []. e eciency of the PA suers
dramatically in case of very large PAPR. Also, the large PAPR
highly aects the linear upconverters leading to high power
consumption.
Investigating the RF part in the transmitter, important
design challenges are faced when the linearity requirement
has to be encountered for high lower levels. Although the
power levels in the RF part of the receiver are much lower,
the presence of adjacent channel interference from dierent
radio systems also adds to the design challenges. For linearity
requirements, the expected level of blocker power above the
desired channel should be also included in the design of
received components. Moreover, the receiver must operate in
a wide dynamic range in order to handle a large interferer
and at the same time receive a much smaller wanted signal.
Two challenges arise from this diculty related to achieving
acceptable noise gure performance for the overall receiver
and sucient dynamic range for the ADCs. e receiver
should then have a good sensitivity required to achieve low
noise factor (< dB) and low insertion losses (<dB)andhigh
LNA and mixer linearity (IP, IP). It also needs to be a good
blocker immunity and low local oscillator (LO) phase noise
[]. ese requirements are dicult to be handled by the
band limited traditional front-end technology and even by
multimode/multiband and wideband transceivers. Switching
thefront-endsasrequiredisusuallyadoptedinthiscase[].
Typically, these challenges and requirements are attained
through the use of additional components such as surface
acoustic wave (SAW) lters and crystal oscillators. However
the added components are mostly costly and increase the
power consumption, in addition to added lack of exibility.
On-chip lters are suggested to limit these disadvantages, but
at the cost of added signal corruptions.
5.2. Low Noise Ampliers. A CR receiver (Rx) must provide
a relatively at gain and a reasonable input return loss across
BWCR . Consequently, it is challenging to employ traditional
RF circuit techniques. As an example, the switched-band
circuitsorstaggeredtuningformedbyacascadeofstageswith
staggered resonance frequencies proves to be impractical
for such a large bandwidth. is problem is illustrated
in several papers [, ]; however the solutions are still
inconvenient for CR systems. e design of broadband LNA
LD
RD2
Y
M3
M2V
b2
Ls
Rs
M1
X
V
in
V
out
V
b1
V
n1
LD
RD1
V
DD
F : CG/CS stage [].
is an interesting challenge due to the trade-os it governs
between input matching, gain, noise gure, bandwidth, and
voltage headroom. e choice of the topology begins with the
input matching requirement []. e input matching of the
LNA can take on one of several forms:
() a common-gate (CG) stage,
() a common-source (CS) stage with inductive degener-
ation,
() a gain stage with resistive feedback,
() a combination of CS and CG stages.
However, each one of these approaches has its drawbacks.
For example, the rst one suers from a relatively high noise
gure in addition to severe gain-noise trade-os. e second
one does not lend itself to broadband operation.
e fourth approach could be considered as a develop-
ment of the rst two methods by combining the CG and
GS approaches, shown in Figure . is provides additional
voltage gain and forms a dierential output along with the CG
stage, with reduced noise (canceled 𝑀1,𝑉𝑛1)[].However,
this topology still suers from the drawbacks of the CG
LNA, facing serious headroom issues and degraded reection
coecient. Noise cancellation can be realized when the input
signal appears with opposite polarities and the noise of
a device []. e cancellation technique also suppresses
nonlinear components produced by the input device.
5.3. Nonlinearity and LO Harmonics. In addition to third-
order intermodulation, nonlinearity in cognitive radios cor-
rupts the signal path in the presence of large interferers. As
illustrated in Figure  two interferers at 𝑓1and 𝑓2generate a
beat at 𝑓2−𝑓
1as they face even-order distortion in the LNA
and the input stage of the mixer. Due to random asymmetries

International Journal of Antennas and Propagation 
Desired band
(primary CR
signal)
Image
band
RFin IFout
LO oscillator
−Δf
−Δf
Δf
Δf
f
0
f
0
−f
0
−f
0
@f
0
F : Example of even-order distortion in spectrum sensing receivers due to image bands.
Multimodal
quadrature LO
Inputs for other divisions
f
0
N
f
0
M
f
0
P
f
0
N2
f
0
M2
f
0
P2
f
0
N3
f
0
M3
f
0
P3
%N%N%N
%P%P%P
%M%M%M
I/Q
F : Multidecade carrier generation using a single multimodal LO.
within the mixer, a fraction of this random beat leaks to
thebasebandwithoutfrequencytranslation,thusdestroying
thedownconvertedsignal.eLNAitselfalsoproduces
components at 𝑓2+𝑓1and 𝑓2−𝑓1,bothofwhichmayliewithin
BWCR .atis,theLNAbecomesthebottleneck.Another
eect arising from even-order distortion is the demodulation
of AM interferers.
5.4. LO Path Design. Another challenge in CR arises from the
design of Local Oscillators. e carrier synthesis for cognitive
radios must follow three principles []:
() Each frequency component must be produced in
quadrature form.
() Single-sideband (SSB) mixing must be avoided
becauseofitslargespuriouscontent.
() For a frequency divided by an odd number, it must be
divided by  so as to generate quadrature phases.
Using one decade of carrier frequencies, some of the
issues related to the LO path design shoot from the supply
coupling within divider chains. An alternative approach to
multidecade carrier synthesis is illustrated in Figure . e
circuit consists of a quadrature LC oscillator operating at
one of two frequencies (e.g., . GHz and  GHz) and three
dividerchainsprovidingdivideratiosof,,,,,,and
. e worst-case oscillator tuning range in this case is %
compared to .% in the rst case.
e circuit produces quadrature phases at all outputs. An
exception to the third principle prescribed above is given
through the use of quadrature Miller dividers [], in which
an SSB mixer is used to form a Miller loop, and all of
the unwanted frequencies generated by the SSB mixer are
translated to zero, or to its harmonics as they travel to the
output.However,theprincipaldisadvantageofquadrature
Miller dividers is the need for quadrature LO inputs, which
suer from higher phase noise [] and exhibit two possible
but poorly controlled oscillation frequencies [].
5.5. Antenna Cancellation for Simultaneous Cognitive Radio
Communication and Sensing. For real time and accurate
monitoring of the radio spectrum, concurrent communica-
tion and sensing is a vital CR functionality. is depends
on the ability of isolating the communication path from

 International Journal of Antennas and Propagation
12
3
Sensing antenna
Transmit antennas
(a)
Transmit far-eld
Null
(b)
F : (a) e numerical model of the array prototype comprised of three short monopoles and (b) the far-eld beam pattern when
out-phasing the transmit antennas [].
the sensing path to a level which ensures that the local
transmissions do not overload the sensing unit when the
transmitting and sensing antennas are collocated in the same
CR device. is can be achieved by equipping the CR with
redundanttransmitantennastoformaspatiallterthat
selectively cancels the transmit signal in the sensing direction
[]. Such antenna cancellation could result in an isolation
level of  dB as depicted in [] that can be further increased
when combined with active power cancellation in the RF or
baseband stage.
e concept of using multiple antennas has also been
used to cancel the interference via zero forcing or far-eld
null steering [, ]. e transimpedance between the
transmit and sensing ports is hence diminished and the
transmit power is spatially canceled in the sensing direction
before arriving at the sensing antenna. is is dierent from
active ltering where the leaked power is canceled aer the
sensing port in the RF front-end and also dierent from
the passive antenna cancellation [] in the sense that the
destructive interference is designed by properly weighting
the transmit antennas rather than by physically locating
the transmit antennas at proper distances from the sensing
antenna. Consequently, as long as the frequency response of
the phase-shiing components and the antennas is wideband,
this results in a wideband performance. Part of the antenna
isolation is owed to the mutual coupling loss between the
antennas; in addition, an adaptive spatial lter that selectively
nulls the transmit signal in the sensing direction can be
formed, as shown in the example in Figure (a) for which
two antennas are dedicated for data transmission and the
third antenna is dedicated for sensing. e null steering
shouldbeperformedinanadaptivemanneraccordingto
the scattering channel environment; thus adaptive antenna
cancellation should be implemented in a real-life scenario.
e two transmit signals should be equal and out-phased
by 𝜋radians. Inspecting Figure (b) showing the power
beampattern of the transmit antennas under such excitation,
it is clearly shown that an articial null is formed toward
the sensing antenna via the spatial lter thus enhancing the
antenna isolation.
6. Conclusions
In this paper, a literature review of the RF front-end of CR
systems has been investigated. Starting with a brief descrip-
tion of CR systems, an extensive review of the antennas,
ampliers, and mixers design schemes used in CR systems
has been presented. Moreover, an investigation of some of the
challenges facing the design of RF front-end for CR systems
in the near future has been illustrated. It is very clear that the
design issues include but are not limited to power eciency,
chip dimension, phase noise, and the recongurability, yet the
power amplier remains one of the main bottlenecks.
Conflict of Interests
e authors declare that there is no conict of interests
regarding the publication of this paper.
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