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ARL-TR-7513 ● OCT 2015
US Army Research Laboratory
Harmonic Phase Response of Nonlinear Radar
Targets
by Sean F McGowan, Dr Gregory J Mazzaro,
Kelly D Sherbondy, and Ram M Narayanan
Approved for public release; distribution is unlimited
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Destroy this report when it is no longer needed. Do not return it to the originator.
ARL-TR-7513 ● OCT 2015
US Army Research Laboratory
Harmonic Phase Response of Nonlinear Radar
Targets
by Sean F McGowan and Kelly D Sherbondy
Sensors and Electron Devices Directorate, ARL
Dr Gregory J Mazzaro
Department of Electrical & Computer, Engineering, The Citadel, The
Military College of South Carolina, 171 Moultrie St, Charleston, SC
29409
Ram M Narayanan
Professor of Electrical Engineering, 202 Electrical Engineering East
Building, The Pennsylvania State University, University Park, PA
16802, USA
Approved for public release; distribution is unlimited.
ii
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4. TITLE AND SUBTITLE
Harmonic Phase Response of Nonlinear Radar Targets
5a. CONTRACT NUMBER
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5c. PROGRAM ELEMENT NUMBER
6. AUTHOR(S)
Sean F McGowan, Dr Gregory J Mazzaro, Kelly D Sherbondy, and
Ram M Narayanan
5d. PROJECT NUMBER
5e. TASK NUMBER
5f. WORK UNIT NUMBER
7. PERFORMING ORGANIZATION NAME(S) AND ADDRESS(ES)
US Army Research Laboratory
ATTN: RDRL-SER-U
2800 Powder Mill Road
Adelphi, MD 20783-1138
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ARL-TR-7513
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Approved for public release; distribution is unlimited.
13. SUPPLEMENTARY NOTES
14. ABSTRACT
One of the latest challenges being investigated by the US Army Research Laboratory’s (ARL) Electronics and Radio
Frequency (E&RF) Division is the development of a radar system that can accurately detect and range an electronically
nonlinear target, such as a detonator of an improvised explosive device (IED). Previous nonlinear radar systems detect targets
via transmission of a single frequency ω, stepping (incrementally increasing) this frequency through a wide bandwidth, and
then listening for a response of the 2nd harmonic 2ω; however, the phase information that this harmonic contains and its
relationship to target distance has been largely assumed and unconfirmed. Our most recent experimental tests, both wired and
wireless, have confirmed that this harmonic phase response is constant versus frequency at the target. Using inverse Fourier
transforms, the range of an electronic nonlinear target can be determined from that phase.
15. SUBJECT TERMS
nonlinear, phase, ranging
16. SECURITY CLASSIFICATION OF:
17. LIMITATION
OF
ABSTRACT
UU
18. NUMBER
OF
PAGES
36
19a. NAME OF RESPONSIBLE PERSON
Kelly D Sherbondy
a. REPORT
Unclassified
b. ABSTRACT
Unclassified
c. THIS PAGE
Unclassified
19b. TELEPHONE NUMBER (Include area code)
301-394-2533
Standard Form 298 (Rev. 8/98)
Prescribed by ANSI Std. Z39.18
iv
List of Figures
Fig. 1 General model of harmonic radar,3 where transmitter = Tx and
receiver = Rx ..........................................................................................4
Fig. 2 Wireline experimental setup3 .................................................................6
Fig. 3 Power and phase of reflection from the 24-ft cable with open-circuit
termination at a) fundamental and b) 2nd harmonic3 ..............................8
Fig. 4 Power and phase of 2nd harmonic reflection from ZX60-3011 amp
input: a) amp connected to Port 1 and b) amp connected through the
24-ft cable to Port 1................................................................................9
Fig. 5 Power and phase of 3rd harmonic reflection from the ZX60-3011 amp
input: a) amp connected to Port 1 and b) amp connected through the
24-ft cable to Port 1..............................................................................10
Fig. 6 Power and phase of 2nd harmonic reflection from the ZX60-V63+ amp
input: a) amp connected to Port 1 and b) amp connected through the
24-ft cable to Port 1..............................................................................11
Fig. 7 Power and phase of 3rd harmonic reflection from the ZX60-V63+ amp
input: a) amp connected to Port 1 and b) amp connected through the
24-ft cable to Port 1..............................................................................12
Fig. 8 Power and phase of 2nd harmonic reflection from the ZLW-186MH
mixer RF port: a) mixer connected to Port 1 and b) mixer connected
through the 24-ft cable to Port 1 ..........................................................13
Fig. 9 Power and phase of 3rd harmonic reflection from the ZLW-186MH
mixer RF port: a) mixer connected to Port 1 and b) mixer connected
through the 24-ft cable to Port 1 ..........................................................14
Fig. 10 Power and phase of 2nd harmonic reflection from the ZFM-2000+
mixer RF port: a) mixer connected to Port 1 and b) mixer connected
through the 24-ft cable to Port 1 ..........................................................15
Fig. 11 Power and phase of 3rd harmonic reflection from the ZFM-2000+
mixer RF port: a) mixer connected to Port 1 and b) mixer connected
through the 24-ft cable to Port 1 ..........................................................16
Fig. 12 Power and phase of 2nd harmonic reflection from the Motorola FV300
radio: a) target connected to Port 1 and b) target connected through the
24-ft cable to Port 1..............................................................................17
Fig. 13 Power and phase of 3rd harmonic reflection from the Motorola FV300
radio: a) target connected to Port 1 and b) target connected through the
24-ft cable to Port 1..............................................................................18
Fig. 14 Power and phase of 2rd harmonic reflection from the Motorola T4500
radio: a) target connected to Port 1 and b) target connected through the
24-ft cable to Port 1..............................................................................19
v
Fig. 15 Power and phase of 3rd harmonic reflection from the Motorola T4500
radio: a) target connected to Port 1 and b) target connected through the
24-ft cable to Port 1..............................................................................20
Fig. 16 Unwrapped harmonic phase response for the Motorola FV300 ..........21
Fig. 17 Unwrapped harmonic phase response for the Motorola T4500 ...........22
Fig. 18 Wireless experiment setup ...................................................................23
Fig. 19 Power and phase of 2nd harmonic from the a) Motorola T4500 radio at
2 m away and b) Motorola FV300 radio at 3 m away ........................25
List of Tables
Table 1 Range-to-target for devices tested in wireline experiment ..................23
vi
Acknowledgments
I appreciate the mentorship of Dr Anthony Martone and Mr Kelly Sherbondy. I
would also like to thank Dr Andy Sullivan, Dr Matthew Higgins, Dr Gregory
Mazzaro, Mr Marc Ressler, Mr Edward Viveiros, Dr Ronald Polcawich, Mr Brian
Phelan, Mr Kyle Gallagher, and Mr Roger Cutitta for their assistance and support.
I also wish to acknowledge my fellow interns Richard Pooler, Kristopher Young,
Jason Cornelius, and Philip Saponaro.
1
Student Bio
Sean F McGowan is a senior undergraduate at Pennsylvania State University
pursuing a BS degree in electrical engineering through Penn State’s Schreyer
Honors College. It is his first summer interning at the US Army Research
Laboratory, though he has worked in Penn State’s Optics & Laser Lab earlier in his
undergraduate career. In the future he hopes to further his education by attending
graduate school and obtaining his Masters in electrical engineering, with a focus on
signal processing.
2
INTENTIONALLY LEFT BLANK.
3
1. Introduction/Background
Unlike its traditional, linear counterpart, nonlinear radar offers the ability to detect
and range targets that contain electronically nonlinear components, such as
transistors, diodes, and semiconductors. While many circuit devices, such as
amplifiers, mixers, and converters, exhibit nonlinear behavior, the true potential of
nonlinear radar is shown in its real-world application of detecting a range of targets
from handheld radios to electronic detonators of improvised explosive devices
(IEDs). Targets such as these are difficult to detect with linear radar, due to their
typical size and shape – often small and thin, with a slim geometric profile. This
profile lends itself to a small radar cross section, which can easily be obscured and
interfered with by nearby objects or clutter. Nonlinear radar offers the feature of
high clutter rejection, but this must be weighed against 2 disadvantages: 1) the
harmonic response from the target is extremely weak, and thus difficult to capture,
and 2) the incident power required to generate a detectable response is significantly
higher than that of linear radar.1,2
The nonlinear radar model used in my experiments is akin to a harmonic, stepped
frequency radar. The radar transmits a single frequency, ω; if the radar receives a
harmonic of this fundamental (i.e., 2ω and 3ω), a nonlinear target is present. The
original fundamental frequency ω is then stepped, or increased by an increment Δω,
across a bandwidth until enough harmonic data are collected to verify detection and
calculate distance to target.3
A general diagram of harmonic radar is shown in Fig. 1. In the transmission chain,
the fundamental ω is amplified to increase its incident power on the target, and then
fed through a low pass filter to attenuate any artifacts created from amplification.
This signal is then transmitted to a target, and then reflected back to the receive
chain, the effective reverse of the transmission chain. This time, the signal is fed
through a high pass filter to attenuate the fundamental, while allowing its harmonics
to pass through. The weak harmonic responses are then amplified to allow for easier
detection and measurement.
4
Fig. 1 General model of harmonic radar,3 where transmitter = Tx and receiver = Rx
In earlier studies of nonlinear radar, the phase of these reflected harmonics has
largely been assumed to be constant versus the bandwidth of frequencies
transmitted to the target.3 This previously unconfirmed relationship is necessary to
calculate the distance to target using an inverse Fourier transform. My experimental
designs reexamine this assumption to affirm it and uphold that the harmonic
stepped-frequency methods of radar are widely applicable for the detection and
ranging nonlinear targets.
2. Harmonic Phase Response Theory
Let the transmitted electric field be a single-frequency sinusoid, represented as a
complex exponential:
(1)
Equation 1 shows Et and
φ
t as the initial amplitude and phase of the transmitted
electric field, and ω is its fundamental frequency in radians.
As the transmitted wave travels a distance d to the target in time τ, the amplitude of
its electric field will attenuate as it propagates through the lossy medium, such as a
transmission line or air. Additional, the wave will undergo a phase shift that
proportional to its operating frequency. Arriving at the target, the incident electric
field may be written as
(2)
( )
t
trans t jjt
E t Ee e
φω
=
( )
inc i j jt
E t Ee e
ωτ ω
−
=
5
where Ei < Et. The incident electric field incident Einc can be related to the electric
field that reflects off the target by the standard power series model for nonlinear,
memoryless targets:12–15
(3)
where ap are complex power-series coefficients. The value of a1 is the linear
response of the target, whereas the following coefficients {a2, a3, …} depend on a
variety of properties of the device, such as the specific clutter around the device,
the orientation of the target, and other radio frequency (RF) interference in the
environment.
When the incident electric field described in Eq. 2 is substituted into Eq. 3, and the
expanded, the result is
( )
3
12
22 2 33 3
refl 1 i 2 i 3 i
22 2 33 3
1i 2 i 3 i
...
...
j jt j j t j j t
j
jj
j jt j j t j j t
E t aEe e aE e e aEe e
a e Ee e a e Ee e a e Ee e
ωτ ω ωτ ω ωτ ω
φ
φφ
ωτ ω ωτ ω ωτ ω
−− −
−− −
=+ ++
=+ ++
(4)
This is the reflected electric field, the wave after the transmitted signal has reached
the target and begins to travel back to the radar’s receiver. Assuming the radar is
monostatic (the transmitter and receiver are combined and thus the same distance
to the target), the reflected electric field will experience a second, identical time
delay of τ on its return, so that the electric field received can be described as
(5)
In Eq. 5, EM denotes the electric-field amplitude received at each harmonic M of
fundamental frequency ω. Each harmonic of the stepped fundamental frequency
experiences a phase delay of 2Mωτ –
φ
M. This delay is the key phase information
needed to obtain range-to-target data.
If we describe Eq. 5, the electric field received by the radar, as a phasor, the result
is the complex equation
. (6)
To obtain distance from the equation, a simple substitution of τ = d/up is required.
This relationship simply states the time it take for wave to travel to the target is the
distance to the target divided by the wave’s speed.
( ) ( )
refl in
1
p
p
p
E t aE t
∞
=
=
∑
( )
( )
( )
( )
( )
( )
( )
( )
12
3
12
22 2
rec 1 2
2 24 36
12 3
2
1
...
...
M
jj
jjtj jjtj
j
jj
jtj jtj jtj
jM
jM t
M
M
E t Ee e e e Ee e e e
Eeee Ee ee Ee ee
Ee e
φφ
ωτ ω ωτ ωτ ω ωτ
φ
φφ
ω ωτ ω ωτ ω ωτ
ωτ φ
ω
−− − −
−−−
∞−−
=
=++
=+ ++
= ⋅
∑
( ) { }
rec 2
MM
EM E M
ω φ ωτ
=∠−
6
. (7)
Finally, range-to-target may be calculated by solving for the derivative of the phase
of Erec with respect to radian frequency of the fundamental. However, this is only
true of the harmonic phase response from the target is constant versus frequency.
. (8)
The experimental data collected and delineated in this study confirm that target
distance can be calculated from Erec using Eq. 8.
3. Wireline Experiment
To begin verification of the assumed phase-frequency relationship, a basic wireline
experimental design is used, as depicted in Fig. 2. Both the radar’s transmitter and
receiver is simulated by using the Keysight N5242A PNA-X nonlinear vector
network analyzer (NVNA). The NVNA transmits a series of stepped signals from
Port 1 along 2 cascaded 12-ft MegaPhase F130 transmission lines, whose loss and
distortion is known. These cables simulate the distance between the radar and a
nonlinear target. The incident signal, notated as A1, reaches the target, and is
reflected back into Port 1 of the NVNA, as B1.
Fig. 2 Wireline experimental setup3
The targets used in this wireline experiment are as follows:
• MiniCircuits ZX60-3011+ amplifier (input port)
• MiniCircuits ZX60-V63+ amplifier (input port)
• MiniCircuits ZLW-186MH mixer (RF port)
• MiniCircuits ZFM-2000+ mixer (RF port)
• Motorola FV300 radio (antenna port)
( )
( )
{ }
rec 2
MM p
E M E M du
ω φω
=∠−
( )
( )
rec
rec
rec
2
02
2
Mp
p
p
E M M du
d
EM
u
u
dE
M
ωφ ω
ω
ω
∠=−
∂∠=−
∂
∂
=−∠
∂
7
• Motorola T4500 radio (antenna port)
Each target had SMA input and output ports. For the amplifiers and mixers, unused
ports were terminated with a 50-Ω load. For the Motorola handheld radios, their
antennas were replaced with an SMA end launch soldered to their printed circuit
board (PCB). Every target exhibits nonlinear behavior, with the amps and mixers
typically being components to detect in a larger electronic target, and the radios
more accurately representing a real-world target for a nonlinear radar system to
detect.
For this wireline experiment, the previously derived distance, Eq. 8, can be
simplified to account for the wave’s propagation through a lossy cable like the
MegaPhase F130s.
( ) ( )
21
21
B1
22
pMM
r
uc
dMM
φω φω
ε
ω ωω
−
∂
=− ∠=−
∂−
(9)
In Eq. 9, the propagation speed has been broken down to the speed of light and the
dielectric constant of the transmission line. Additionally, the derivative of the phase
of the return wave has been simplified to a discrete slope equation between 2
collected data points.
Finally, the sweep the NVNA transmits to the target is from 800 to 900 MHz with
a 1.25-MHz step. Each transmitted sinusoid has the same amplitude, but is applied
to the target at 2 different power levels: –10 and 0 dBm. The results are displayed
Figs. 3 through 15, with the frequency reported in Hz.
8
(a)
(b)
Fig. 3 Power and phase of reflection from the 24-ft cable with open-circuit termination at
a) fundamental and b) 2nd harmonic3
9
(a)
(b)
Fig. 4 Power and phase of 2nd harmonic reflection from ZX60-3011 amp input: a) amp
connected to Port 1 and b) amp connected through the 24-ft cable to Port 1
10
(a)
(b)
Fig. 5 Power and phase of 3rd harmonic reflection from the ZX60-3011 amp input: a) amp
connected to Port 1 and b) amp connected through the 24-ft cable to Port 1
11
(a)
(b)
Fig. 6 Power and phase of 2nd harmonic reflection from the ZX60-V63+ amp input: a) amp
connected to Port 1 and b) amp connected through the 24-ft cable to Port 1
12
(a)
(b)
Fig. 7 Power and phase of 3rd harmonic reflection from the ZX60-V63+ amp input: a) amp
connected to Port 1 and b) amp connected through the 24-ft cable to Port 1
13
(a)
(b)
Fig. 8 Power and phase of 2nd harmonic reflection from the ZLW-186MH mixer RF port:
a) mixer connected to Port 1 and b) mixer connected through the 24-ft cable to Port 1
14
(a)
(b)
Fig. 9 Power and phase of 3rd harmonic reflection from the ZLW-186MH mixer RF port:
a) mixer connected to Port 1 and b) mixer connected through the 24-ft cable to Port 1
15
(a)
(b)
Fig. 10 Power and phase of 2nd harmonic reflection from the ZFM-2000+ mixer RF port:
a) mixer connected to Port 1 and b) mixer connected through the 24-ft cable to Port 1
16
(a)
(b)
Fig. 11 Power and phase of 3rd harmonic reflection from the ZFM-2000+ mixer RF port:
a) mixer connected to Port 1 and b) mixer connected through the 24-ft cable to Port 1
17
(a)
(b)
Fig. 12 Power and phase of 2nd harmonic reflection from the Motorola FV300 radio:
a) target connected to Port 1 and b) target connected through the 24-ft cable to Port 1
18
(a)
(b)
Fig. 13 Power and phase of 3rd harmonic reflection from the Motorola FV300 radio:
a) target connected to Port 1 and b) target connected through the 24-ft cable to Port 1
19
(a)
(b)
Fig. 14 Power and phase of 2rd harmonic reflection from the Motorola T4500 radio: a) target
connected to Port 1 and b) target connected through the 24-ft cable to Port 1
20
(a)
(b)
Fig. 15 Power and phase of 3rd harmonic reflection from the Motorola T4500 radio: a) target
connected to Port 1 and b) target connected through the 24-ft cable to Port 1
21
The assumption that harmonic phase response is constant with frequency is
empirically confirmed by the data shown in Figs. 4a through 15a, which delineate
a mostly flat phase response when the target is directly connected to Port 1 of the
NVNA. Distance can be calculated using this phase response data at either the 2nd
or 3rd harmonic of the fundamental. This is shown more clearly in Figs. 16 and 17,
which contain the unwrapped phase plots for the FV300 radio (Figs. 12b and 13b)
and T4500 radio (Figs. 14b and 15b), respectively, at –10 dBm.
Fig. 16 Unwrapped harmonic phase response for the Motorola FV300
22
Fig. 17 Unwrapped harmonic phase response for the Motorola T4500
For the FV300 radio, at f = 800 MHz,
φ
2 = –75°, and at f = 900 MHz,
φ
2 = –4375°.
Using a delay of 1/up = 1.27 ns/ft, as listed in the Megaphase F130 manufacturing
specifications,12 range-to-target may be calculated from the phase response at 2f
using Eq. 9:
(10)
With MATLAB, this process can be repeated to generate the distance to each tested
target from the phase response of the 2nd or 3rd harmonic (Table 1).
23
Table 1 Range-to-target for devices tested in wireline experiment
Target
M
d
Target
M
d
MiniCircuits
2
23.5 ft
MiniCircuits
2
23.6 ft
ZX60-3011+
3
23.4 ft
ZFM-2000+
3
23.7 ft
MiniCircuits
2
23.3 ft
Motorola
2
23.5 ft
ZX60-V63+
3
23.4 ft
FV300
3
23.8 ft
MiniCircuits
2
23.6 ft
Motorola
2
23.6 ft
ZLW-186MH
3
23.6 ft
T4500
3
23.6 ft
open circuit
1
23.2 ft
Each distance calculated from the phase data is well within 5% of the actual
distance (24 ft), confirming that using harmonic phase information to determine
range is a valid technique for use in nonlinear radar.
4. Wireless Experiment
Figure 18 shows the next experimental design used to validate the harmonic phase
response, a wireless setup slightly more complex than the wired system previously
discussed.
Fig. 18 Wireless experiment setup
Like the wireline tests, this experimental design uses the Keysight NVNA to
generate the series of transmit signals, as well as receive the harmonic response;
though in this wireless configuration, the received signal enters Port 2, instead of
returning to the same port, Port 1, as in the previous design. Similar to the wireline
tests, the transmission chain begins with signal generation, amplification, and low
pass filtering; however, the wireless configuration employs a much more powerful
24
amp, the MiniCircuits ZHL-42W, with a gain of roughly 38 dB. Again, the low
pass filter acts to attenuate harmonics or artifacts that could be generated during
amplification. The first major difference between the wired and wireless setups is
the next component in the transmitter chain, a configuration of diplexers that serve
3 functions: 1) divert any remaining harmonics that were not sufficiently attenuated
by the first filter to a 50-Ω load, 2) send the desired, artifact-free signal through
another series of filters to the vertically aligned Schwartzbeck 9120E antenna to be
transmitted to the target, and 3) on the receive chain, attenuating the received
fundamental frequency, while passing the harmonic response to Port 2 of the
NVNA to be captured.
The targets under test for this wireless experiment were the same handheld
Motorola radios used in the wireline experiment, the FV300 and T4500. Both
targets were powered-on and tested at distances of both 2 and 3 m, directly in front
of the antenna. The bandwidth of frequencies the NVNA generates is 750 to
850 MHz with a step of 0.625 MHz, at a power of –10 dBm.
The same range equations used to determine distance from phase previously
derived and described in Eqs. 8 and 9 can be used in the wireless case, with one
small modification. Because of the more complex and thus longer transmitter and
receive chains, the time it takes to travel the additional distance to and from the
NVNA must be accounted for. This, mathematically, simply translates to a
subtraction of d0, or the distance that the signal travels from the NVNAs Port 1 to
the antenna plus the distance that the harmonic response traverses from the antenna
to the NVNAs Port 2, divided in half.
( ) ( )
( )
21
rec 0 0
21
2 22
pMM
uff
c
d Ed d
M M ff
φφ
ωπ
−
∂
=− ∠ −=− −
∂−
(11)
For the radar system used in this experiment, the distance d0 is 3.2 m. The 2nd
harmonic response is shown in Fig. 19 for both radios and 2 different distances.
25
(a)
(b)
Fig. 19 Power and phase of 2nd harmonic from the a) Motorola T4500 radio at 2 m away and
b) Motorola FV300 radio at 3 m away
26
Though these data are less strong than their wireline counterpart, range-to-target
can still be calculated using Eq. 11, when the target’s amplitude response has a
sufficient enough signal-to-noise ratio to discern it from the noise floor. Using the
2nd harmonic (M = 2) reflected from the T4500 radio and data near 760 MHz (the
most visibly straight linear portion of the phase plot):
(12)
The results of the wireless experiment, as shown in Eq. 12, corroborate the data
collected in the previous wired experiment. Again, the conjecture that the received
harmonics are linear as a function of frequency is verified, so that range can be
calculated from the harmonic’s phase.
5. Conclusions
Previous nonlinear harmonic radar systems detect targets via transmission of a
single frequency ω, stepping (incrementally increasing) this frequency through a
wide bandwidth, then listening for a response of the 2nd harmonic 2ω; however, the
phase information that this harmonic contains and its relationship to target distance
has been largely assumed and unconfirmed. This assumption was verified through
2 experiments, 1 wired and 1 wireless, where the phase of the 2nd and 3rd harmonic
of the received electromagnetic wave from nonlinear targets was measured and
plotted against the frequency. The result was a linear relationship, in which range-
to-target could be calculated from slope.
27
6. References
1. Kosinski JA, Palmer WD, Steer MB. Unified understanding of RF remote
probing. IEEE Sensors. Dec. 2011;11(12):3055–3063.
2. Steer MB, Wilkerson JR, Kriplani NM, Wetherington JM. Why it is so hard to
find small radio frequency signals in the presence of large signals. 2012
Workshop on Integrated Nonlinear Microwave and Millimetre-Wave Circuits
(INMMIC), pp. 1–3, Sept. 2012.
3. Mazzaro GJ, McGowan SF, Gallagher KA, Sherbondy KD, Martone AF,
Narayanan RM. Phase responses of harmonics reflected from radio-frequency
electronics. SPIE DSS 2016 Conf. Aug. 2015.
4. Mazzaro GJ, McGowan SF, Gallagher KA, Sherbondy KD, Martone AF,
Narayanan RM. Phase Responses of Harmonics Reflected from Radio-
Frequency Electronics,” in preparation for SPIE DSS 2016, Baltimore, MD,
Apr. 2016.
5. Mazzaro GJ, McGowan SF, Gallagher KA, Martone AF, Sherbondy KD.
Harmonic phase responses of radio-frequency electronics: wireline test.
Adelphi (MD): Army Research Laboratory (US); in preparation.
6. Mazzaro GJ, Ranney KI, Gallagher KA, McGowan SF, Martone AF.
Simultaneous-frequency nonlinear radar: hardware simulation. Adelphi (MD):
Army Research Laboratory (US); August 2015. Report No.: ARL-TN-0691.
28
1 DEFENSE TECHNICAL
(PDF) INFORMATION CTR
DTIC OCA
2 DIRECTOR
(PDF) US ARMY RESEARCH LAB
IMAL HRA
RDRL CIO LL
1 GOVT PRINTG OFC
(PDF) A MALHOTRA
11 DIRECTOR
(PDF) US ARMY RESEARCH LAB
ATTN RDRL SER U
T DOGARU
M HIGGINS
D LIAO
A MARTONE
D MCNAMARA
G MAZZARO
K RANNEY
M RESSLER
K SHERBONDY
G SMITH
A SULLIVAN
