Conference PaperPDF Available

Performance and Implementation of a Multi-Rate IR-UWB Baseband Transceiver for IEEE802.15.4a

Authors:
  • Institute for High Perfomance microelectronics and TU-Cottbus

Abstract and Figures

Design, simulation, implementation and performance of IR-UWB baseband conforming to IEEE802.15.4a are discussed. The baseband can support various data rates such as 850 Kb/s, 6.81 Mb/s and 27.24 Mb/s. The design and parameter selection were considered carefully taking into account all possible imperfections that IR-UWB high frequency signal can experience. Energy detection receiver employing a comparator clocked at 499.2 MHz was adopted for the digitisation. Using I and Q path both positive and negative pulses were detected with a high reliability leading to a very good synchronisation performance. Simulation results confirm that the synchronisation is very robust being always correct for office NLOS environment and a large clock deviation between transmitter and receiver. The algorithm presented in this paper was implemented with discrete components, FPGA and signal generators. Experimental results show a good agreement with the simulation for all the data rates and the implemented baseband offers around six meter communication range tested along with a high frequency frontend from discrete components.
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Performance and implementation of a multi-rate
IR-UWB baseband transceiver for
IEEE802.15.4a
S. Olonbayar, D. Kreiser, R. Kraemer
IHP
Im Technologiepark 25
15236 Frankfurt Oder, Germany
sonom@ihp-microelectronics.com
Abstract: Design, simulation, implementation and performance of
IR-UWB baseband according to IEEE802.15.4a is discussed. The
baseband can support the various data rates such as 850 Kb/s,
6.81 Mb/s and 27.2 Mb/s. The design and parameter selection
were considered carefully taking into account all possible
imperfections that IR-UWB high frequency signal can
experience. Energy detection receiver employing a comparator
clocked at 499.2 MHz was adopted for the digitisation. Using I
and Q path negative pulses were detected with a high reliability
leading to a very good synchronisation performance. Simulation
results confirm that the synchronisation is very robust being
always correct for office NLOS environment and large clock
deviation between transmitter and receiver. The algorithm
presented in this paper was implemented with discrete
components, FPGA and signal generators. Experimental results
show good agreement with the simulation for all the data rates
and the implemented baseband offer around six meter
communication range tested along with high frequency discrete
components.
Key words IR-UWB, Digital baseband, clock deviation,
synchronisation, comparator, pulse shaping, non-coherent energy
detection, IEEE802.15.4a
I. I
NTRODUCTION
There exist a number of ways to convey data wirelessly from
one point to another which are geographically separated. The
most common way is to use continuous wave known as carrier
with certain frequency and vary its parameters in accordance
to the data signal being sent. Such systems are widely in use
today such as mobile communications, WiFi, Bluetooth etc.
The other method is to transmit pulses whose presence signals
the receiver data. If transmit pulses possess duration of less
than 2 ns taking a Gaussian shape it falls under the category of
impulse radio ultra wideband (IR-UWB) due to its extremely
large bandwidth exceeding 500 MHz. IR-UWB was identified
as a promising technology in enabling low power consumption
and precise range determination. IEEE has recognised its
suitability and the physical layer based on IR-UWB were
specified leading to the emergence of the standard
IEEE802.15.4a. It describes four data rates namely 110 kb/s,
850 kb/s, 6.81 Mb/s and 27.2 Mb/s to be transmitted across the
frequency band 3.1-10.6 GHz occupying the bandwidth of 500
MHz. Furthermore, ranging is supported with the standard.
This paper describes the implementation and performance of
the baseband along with the interfacing to the radio frequency
frontend fully compatible to the standard mentioned.
II. IR-UWB
T
RANSMITTER
According to the standard, the transmitter performs several
tasks preparing data to be radiated through an antenna. These
involve channel doing schemes for improving link
performance such as convolutional coding and Reed-Solomon
coding. For the data coming from MAC layer with the length
of up to 128 bytes firstly Reed –Solomon coding is performed
by adding 48 redundancy bits to each 330 bits of data for error
correction.
Figure 1 Block diagram of IR-UWB Transmitter
A half rate R=1/2 convolutional coder with the constraint
length of three (K=3) and the generator polynomial g
0
=[010]
2
,
g
1
=[101]
2
is used for correcting random errors that can occur
during transmission and reception. With Reed-Solomon
encoding a block of I bits is encoded into a codeword of I+48.
The notation RS
6
(63,55) is applied to the encoding showing
that 55 symbols are encoded to 63 having the redundancy of 8
symbols. The RS coding is designed to correct burst errors by
replacing the burst covering up to 8 consecutive bytes by the
correct one. The coded data then modulated with pulse
position modulation PPM and BPSK. PPM is performed by
positioning pseudo random sequence generated from LFSR in
either first or second half of the symbol duration. Symbol
durations are variable depending on the data rate to be
supported as shown below.
850 Kb/s 1024 ns
6.81 Mb/s 128 ns
27.24 Mb/s 64 ns
R-S
Conv
PPM-BPSK
LFSR
Pream.
16:1
Ser
Pulse
gen.
Mixer
7.9872
Ampli
fier
MAC
As can be seen with increasing data rate, more and more bits
are packed in shorter period of time. For each data rate, within
a symbol period, two bits are sent one is carried in the position
and the other is carried in the phase of the burst.
After the error correction and protection is done preamble and
start of frame delimeter (SFD) are placed in front of the
modulated data for signalling and synchronisation purposes.
Creating a frame which contain preamble, SFD and data
payload concludes the function to be performed by the
baseband.
All the baseband operations stated before can be realised with
the clock rate of 31.2 MHz for power saving as well as
suitability reasons to the chip fabricating technology SGB25V
from IHP Germany. With a 16:1 serialiser running at 499.2
MHz clock, 16 bit 31.2 MHz digital signal is serialised into a
499.2 MHz stream. A pulse generator generates Gaussian
pulses having a duration of 2 ns for the data coming from the
serialiser.
A frame corresponding to the data rate 850Kb/s
A frame for 6.81 Mb/s
A frame for 27.24 Mb/s
Figure 2. IEEE802.15.4a IR-UWB frame in analogue
baseband corresponding to different data rates captured from a
scope
The frame signal is then processed through the radio
frequency frontend such as up conversion and amplification to
the level suitable for radiation through the antenna. The main
issue for an IEEE802.15.4a IR-UWB transmitter is to ensure
compatibility to the spectrum mask issued by FCC. It is
achieved by utilising suitable pulse shaping techniques. One
way is given in [7].
III. IR-UWB
R
ECEIVER
The performance of IR-UWB is highly dependent on the
receiver. As IR-UWB signal strength is very weak, the
receiver needs to have a good sensitivity and possess some
processing capabilities to recover data from such low signal
level being around background noise. Since IR-UWB is
intended for use in battery driven devices, power consumption
is expected to be low. Achieving a good performance by
keeping the power low is the challenge the designers of IR-
UWB face. Especially if certain mechanisms have to be
implemented to cope with multipath fading, the design can
become complex leading to increased consumption.
For lower data rates, receive signal can be integrated in
analogue domain which permits to use slower analogue to
digital conversion afterwards. Such receiver which does an
analogue integration over 16 ns and digitises it at 62.5 MSPS
for the data rate of 850 kb/s is presented in [1]. It achieved a
range of approximately 12 m with a reasonable packet error
rate. The negative side for such type of detection is that it fails
to receive polarity bits degrading the performance. This work
extends it by introducing another two higher data rates namely
6.81 Mb/s and 27.41 Mb/s. As these data rates transmit only
two and one narrow pulses (2 ns) respectively within a symbol
period compared to 850 Kb/s which sends 16 pulses, the
digitisation has to be carried out fast enough to be able to
resolve individual pulses. For this reason, analogue baseband
signal is digitised with a comparator which outputs digital
signal when it exceeds the threshold set.
Figure 3. Comparator operation for I path
At the receiver, down-converted and amplified baseband
signal can have the waveform shown in Figure 3. For clarity,
very clean and good shaped signal is illustrated and in reality
it might be bit distorted and subjected to background noise.
Two thresholds are used as the signal contains both positive
and negative pulses. It can be seen that the preamble can be
detected by setting an appropriate threshold. Most analogue
comparators produce an output only when the input signal
goes higher the threshold. Since the duration of pulses is 2 ns,
the comparator produces a signal lasting shortly maybe less
than 1 ns depending on the threshold and the noise floor.
Figure 4. IR-UWB receiver architecture
A level shifter could be used to bring the comparator output to
CMOS level making it digital. Therefore, it can be said that
the comparator and level shifter act like an analogue to digital
converter with the resolution of 1,5 bit and the sampling rate is


Non-
coherent
HF frontend
I path
Q path
Comp
Comp
S/P
S/P
I+
I
-
1
32
1
32
Synch
PA.
det
PHR det.
data rate
packet length
PPM BPSK
Viterbi
R.S dec
FIFO
Dec.
block
499.2 MHz. The digitised signal is then parallelised to 16 bit
31.2 MHz stream for digital baseband processing.
For the implementation of IR-UWB transceiver the following
factors have to be considered with care 1) clock accuracy and
2) multipath propagation. As IR-UWB devices are assumed to
be used for wireless sensor communication, clock accuracy is
particularly important especially for ranging and for higher
data rate. The standard IEEE802.15.4a allows clock drift up to
20 ppm. Our baseband as well as HF frontend uses the
reference clock 31.2 MHz and derives the other necessary
clocks from it. 20 ppm translates to 160 KHz deviation for the
carrier frequency of 7.9872 GHz. Since transmitter and
receiver always operate independent of each other and the
reference clock is not the same, there will be certain deviation
of clock frequencies used at transmitter and receiver. In
practice, crystal oscillators can deviate depending on factors
like temperature. For analytical purposes, transmit and receive
frames are generated where certain clock deviation is
introduced and these appear in Figure 5.
a) receive signal waveform corresponding to carrier frequency
(7.9872 GHz) deviation of 160 kHz
b) receive signal waveform corresponding to carrier frequency
(7.9872 GHz) deviation of 10 kHz for I and Q paths
Figure 5. Waveform of signals corresponding to various
deviation of carrier frequency at transmit and receive side
As can be seen from the Figure 5, depending on the degree of
deviation, the waveform is distorted in different ways. For
larger deviation the phase of the signal is altered periodically
quite often (see Figure 5a). Polarity detection for individual
pulses is not simple in this case, particularly for payload data
represented by a burst consisting of pulses. It is due to the
limited performance of pulse shaping filter which always need
some time for transitioning from 0 to 1 and vice versa lacking
a sharp change to other state. Therefore, it necessitates a very
high sampling rate for delivering required accuracy or to hit
the exactly the peak of the pulse which makes the realisation
quite hard.
At the output of comparator the constellation diagram for
digitised signal looks like as illustrated in Figure 6. The signal
at any instance of time (sampling every 2 ns) has the following
constellation: 0 is for zero, 1 for positive pulse and -1 for
negative pulses.
Q
1
I
-1
a) Constellation diagram b) Decision block
Figure 6 Constellation diagram of PPM-BPSK for IR-UWB
For detecting signal, the constellation diagram is observed
continuously every 2 ns and the decision is made whether 1, 0
or -1 is received. Using I and Q path is important to detect
polarity of pulses. These are generated at the down-conversion
by multiplying the signal with the local carrier for I path and
Q path is produced by multiplying the signal with the same
local carrier with the phase shifted by 90
0
. With these I and Q
paths it is possible to detect polarity. Due to the deviation of
carrier frequency at receiver, the signal may even disappear at
certain point as presented in Figure 5. As I and Q paths are 90
0
apart from each other in phase, when one experiences low
signal level the other shows higher signal strength. It can
effectively be exploited for reception especially in detecting
reversed frames which is the case for SFD.
The detection was done in the following way: considering
both I and Q for detection and seeing which one delivers the
expected signal. In the preamble phase it can easily be seen
which one is delivering better signal. By performing a
correlation it can be established where the reception is more
reliable. Since the preamble is made up from repetition of
ternary signal (1, 0, -1) with the length of 31 by inserting 15
zeros between its elements, we observe the correlator output
for both I and Q. The one which spots preamble is used as a
reference. If for instance, I path finds preamble, it is assumed
that the signal is not shifted in phase by 180
0
. Essentially we
perform four correlations. These are I and Q signals correlated
with the known expected sequence and with its reversed one.
After the decision is made as to whether 1, 0 or -1, the decided
bits are parallelised at 31.2 MHz creating 16 bit at this clock
rate. The digital baseband receives this parallelised signal and
does the correlation. 16 parallel operating correlators are used
to achieve the accuracy of 2 ns. Such a parallelism allows the
digital baseband to run at lower clock rate which is 31.2 MHz
and offers at the same time the above accuracy which is
necessary for polarity detection.
0 2 4 6 8 10
12
x 10
5
-1.5
-1
-0.5
0
0.5
1
1.5
0 2 4 6 8 10 12
x 10
5
-1.5
-1
-0.5
0
0.5
1
1.5
0 2 4 6 8 10 12
x 10
5
-1.5
-1
-0.5
0
0.5
1
1.5
1
-1
Dec.
Block
499.2
MHz
I
I-
Q
Q-
1,0, -1
I path
Q path
In the preamble phase, if there is no deviation in clock or in
other words if clocks at both transmit and receive side are of
the same frequency, I or should Q path should deliver signal
that can be detected. In case the start phases for transmit and
receive signal happen to be 0
0
or 90
0
, one of them shows a
good detectable signal. If I path was for instance, to deliver
signal then no phase shift is assumed and the detection is
proceeded with this path. If Q path was found to be strong,
phase shift by 180
0
degree is assumed on it and polarity for
pulses are perceived to be reversed. This way detection can be
performed effectively exploiting both I and Q path.
When clocks start to deviate, both paths would deliver signal
with a period dependent on the extent of deviation. Hence I
and Q paths are taken into account interchangeably permitting
polarity detection. Furthermore, the amount of deviation could
be estimated observing the rate of switch between the paths. It
is possible if the preamble is long enough and the deviation is
large so that the change occurs at least once within a preamble
period (see Figure 5a for the preamble length of 64). Once this
deviation is known, it can be considered for data detection by
shifting the detection window left and right in time. One other
way of dealing with such deviation is to compensate it by
sensing the degree of the drift in baseband and adjust the
clock. As clock deviation or drift is rather deterministic it can
be estimated and its influence can be effectively reduced.
Multipath propagation
Multipath propagation plays and important role in the design
of IR-UWB receiver. Especially detecting polarity becomes
challenging in multipath environment. If there is no multipath
the reception is performed as described earlier. In the presence
of multipath, the link performance has to be improved by
reducing the packet error rate. Multipath path describes copies
of signal arriving at the receiver via different ways due to
reflection, diffraction and refraction. These copies are then
superimposed making it demanding to sort out which signal
belong to which replica. As long as different paths are not
resolved correctly, the detection may not be reliable.
Since signal replicas are usually slightly displaced in time
from each other, all types of overlapping is possible. Some
signal may overlap with out of phase delivering unexpected
constellation. Therefore, a sort of signal processing is required
to resolve multipath. Well known method for that is rake
receiver which combines signal received in different paths.
Since shorter paths are more likely to occur, resolving them
may require quite intensive processing.
The delay spread for ultra wideband signal can be as much as
few hundred nanoseconds. As a consequence, bursts in data
field can still interfere with each other despite of guard
interval being 256 ns for 850 Kb/s and 32 ns for 27.24 Mb/s.
As can be seen, higher data rates are more vulnerable to
intersymbol interference.
Data detection
Once synchronisation is achieved with the accuracy of 2 ns
detecting SFD part commences. 19 bits comprising SFD
which are transmitted always at 850 Kb/s deliver information
such as data rate, packet length and etc needed for receiving
the packet. This part is partly guarded against errors with
Hamming coding. Based on the data rate to be received,
payload data detection starts by looking at the energy level at
both first half and second half of symbols the duration of
which is extracted from data rate bits.
The receiver has to detect not only energy corresponding to
position bit but also the polarity of pulses making up a burst.
For lower data rates especially at 850 Kb/s, position bits can
still be recovered regardless of a shift of burst due to the clock
drift as the burst duration is as large as 32 ns. Higher data rates
require more accurate clock as burst consist only one or two
pulses.
Parallel to detecting position bits, polarity bits can be
recovered too. In order to do that it has to be established if
whether the sign or polarity of pulses creating a burst have
been reversed. If so, polarity bit is detected to be “1”
otherwise “0”. As stated earlier due to the clock drift the burst
could be received with reverse even the burst is sent without
reversing the phase. If at the beginning of payload detection
the polarity of the first data bit is known, the rest can be
detected referencing to it. Polarity is received by doing
correlation between expected polarities and the received
polarities within a sequence (burst). A burst with varying
signs is generated for each symbol at the transmitter. The
receiver performs two correlations for each burst firstly
receive burst with the expected one and secondly with the
expected that is reversed. Each time receiver detects bit, it sees
if the constellation has been changed compared to the previous
one. If yes, it can be seen which direction is it clockwise or
counter-clockwise. By observing all four inputs of the decision
block therefore the polarity of pulses can be recovered. The
phase received at I or Q is compared with the previous phase.
If it is shifted by 180
0
it is detected to be a negative otherwise
positive pulse.
IV. S
IMUATION RESULTS
Extensive simulations were carried out before starting the
implementation. At the transmitter, the baseband signal was
generated with appropriate shaping after channel encoding and
modulation were done. The frame signal for different rates
was produced and up-converted to 7.9872 GHz. AWGN noise
was introduced and wireless IR-UWB channel was simulated
according to the popular known model of Saleh-Valenzuela.
At the receiver after down-conversion, the signal was made
digital through a comparator by comparing it with the
threshold set and digitising the comparator output. Extensive
searching of preamble as well as SFD was performed to
establish synchronisation and get prepared for detecting data.
Simulation was done for various signal to noise ratio (SNR),
clock drift and multipath propagation. For the data rate of 850
Kb/s the synchronisation was found to be always 100%
reliable for quite large carrier frequency difference at the
transmitter and receiver which is 160 kHz and in office NLOS
channel (see Figure 7). The performance was starting to
degrade only for very high deviation that can be higher than
500 MHz.
Figure 7 Synchronisation performance for various frequency
deviation at transmitter and receiver, SNR=0 dB, office NLOS
and 500 different channel realisations
Synchronisation performance stays the same for all data rates
as the preamble structure is identical. Position bits were
always received correctly indicating a perfect synchronisation
performance of the receiver for all implemented data rates. For
SNR of -5 dB data bits are received correctly in NLOS office
channel as can be seen from Figure 8 without needing a
channel decoding. Moreover, all data rates show similar
performance indicating again a very accurate synchronisation
performance and a suitable way of tackling deviation of carrier
frequency at transit and receive sides.
Figure 8. Frame reception percentage (position bits) for
different data rates, frequency deviation of 160 kHz, office
NLOS
For a quite high deviation in frequency amounting 500 MHz
the receiver still delivers acceptable performance. The
obtained results refer only to the received position bits which
contain whole data bits.
Experiment
Besides carrying out some simulations into the detection
algorithm described, experiments were done. IR-UWB signal
with different data rates compatible to IEEE802.15.4a was
generated using an arbitrary waveform generator N8242A
from Agilent and the corresponding waveforms are shown in
Figure 2. The generator was programmed with Matlab codes
and for shaping the pulses rcosine filter with F
s
=998.4MHz
and F
d
=499.2 MHz was used which produces Gaussian shaped
pulses having the duration of 2 ns. This baseband signal is
then up-converted to 7.9872 GHz with E8257D PSG generator
making the signal ready for radiation. It can also be done with
mixers available as discrete component for instance from
Mini-circuits.
Figure 9. Experimental setup
At the receiver after the down-conversion and low pass
filtering using discrete components down converted baseband
signal was digitised with AD9286 from analogue devices
operating at 499.2 MHz sampling rate. For this experiment
AD9286 evaluation board and a capture board with FPGA
Virtex IV from Xilinx on it was used for doing digital
processing on the captured signal from the ADC (see Figure
9).
The signal applied to the ADC was recorded with the capture
board and it is shown in Figure 10.
Figure 10. Captured preamble
As can be seen the preamble can easily be recognised having
three states. Therefore synchronisation performance with this
algorithm is quite acceptable. However, as the sampling rate is
not high enough (2 ns duration pulses were sampled with
499.2 MHz), usually pulses were sampled two times within 2
ns period. This is partly due to imperfect pulse shaping
however by setting the threshold higher, the peak may be hit.
If we see it from analogue domain, the baseband exhibits the
bandwidth of 250 MHz and the sampling criteria is thus
satisfied. The sampling frequency could have been made 1
GHz for better performance. However, as an ASIC solution in
250 nm CMOS technology is intended, 499.2 MHz was
chosen for suitability to implement it. 499.2 MHz sampling
permits to attain a satisfactory synchronisation performance as
16 parallel operating correlators used which are displaced in 2
ns from each other. Despite the double samples for each pulse,
the synch algorithm can still spot preamble and SFD.
0 160 320 640
94
95
96
97
98
99
100
Carrier frequency deviation at trans mitter and receiver (KHz)
Percentage of succes ful
synchronisation
-10 -5 0 10
0
20
40
60
80
100
SNR (dB)
Succe ssful detection
percentag e of bits
27 Mb/s
6 Mb/s
850 Kb/s
3.99 4 4.01 4.02 4.03 4.04
4
-1
-0.5
0
0.5
1
Eval and
capture board
AWG
N8242A
E8257
PSG
However, phase detection in data field is not simple as the
transition from positive to negative pulse always requires
certain time and the slope is not steep. This leads to degraded
performance in detecting phases as illustrated in Figure 11.
Burst representing 2 bits at
850 850 Kb/s
Transmitted burst in ternary
format
Recorded burst in ternary
format
Figure 11. Comparison of sent and recorded burst
As visualised in Figure 11, there is some discrepancy between
sent and the recorded burst even though the clocks are
synchronous. This algorithm can detect energy and therefore
position bits are usually received correctly. However, polarity
bits are not always reliable in part due to pulse shaping,
inaccurate synchronisation and improper threshold. Polarity
bits that are received incorrectly prior to channel decoding
could be corrected through Reed Solomon and half rate
convolution decoding.
V. C
ONCLUSION
In this paper the baseband architecture, design,
implementation and performance are presented. An energy
detection receiver incorporating a comparator sampled at
499.2 MHz was adopted. The baseband supports data rates
850 Kb/s, 6.81 Mb/s and 27.2 Mb/s according to the standard
IEEE802.15.4a. For the simulation all necessary imperfections
that the radio frequency signal can experience were
considered. Moreover, for the design, simulation and
implementation clock drift, frequency deviation at receiver
and transmitter and multipath propagation were taken into
account. Simulation results suggest that the synchronisation
performance is quite acceptable even for large carrier
frequency differences at transmit and receive sides being
100% correct for the deviation of 160 kHz. I and Q processing
at the digital baseband offered a good performance in
detecting negative pulses. Experiments involving transmission
and reception using the FPGA baseband and radio frequency
frontend show good agreement with the simulation results
obtained. Experimental system supported all the data rates at
the communication range of around six meter.
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x 10
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Impulse Radio Ultra-Wideband (IR-UWB) communication system according to the standard IEEE.802.15.4a supporting non-coherent reception is discussed with the focus on the automatic gain controller (AGC). The performance of the transceiver is evaluated under noisy and common multipath channel environments. Simulation results confirm that a simple non-coherent, low power IR-UWB receiver can achieve the sensitivity of -70 dBm for the input signal with 250 MHz analogue using a 6 bit analogue to digital converter with a sampling frequency of 62.4 MHz. It presents the maximum achievable performance from the system under consideration where integrating circuit and slower analogue to digital converter are incorporated. The sensitivity of the receiver can be further improved by using a higher resolution ADC where -100 dBm received signal is still detected with 12 bit ADC.
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Using UWB for wireless short range, low rate communication is attracting growing interest due to its low power consumption and very high bandwidth. Moreover, it is able to offer accurate localization in the range of few centimeters. Considering these qualities, it is desirable to design an IR-UWB transceiver which consumes as small power as possible when it is applied for battery driven wireless sensors. For this reason, an IR-UWB transceiver based on the standard IEEE.802.15.4a was investigated in this paper with particular emphasis on its implementation, power consumption and area need for both FPGA and ASIC solutions. This paper presents results obtained with real hardware.
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This paper describes a monolithic integrated transceiver chipset intended for impulse radio (IR) Ultra-wide band (UWB) applications including indoor communication and indoor localization. The chipset operates in the higher UWB band centered at 7.68 GHz and it is optimized for a pulse bandwidth of about 1.5 GHz. The average pulse repetition rate of 60 MHz and an octagonal pulse position modulation (8-PPM) allow for raw data rates up to 180 MBit/sec. The available high bandwidth is used for precise indoor localization employing a dedicated time-of-arrival (TOA) measurement extension. This unit runs with an on-chip system clock of 3.84 GHz, which allows a measurement accuracy of 260 picoseconds. As demonstrated this UWB transceiver chipset is well suited for two-way ranging (TWR) in potentially harsh RF propagation environments. Under perfect line-of-sight conditions a spatial resolution of about 3.9 centimeter could be achieved.
Conference Paper
Using UWB for wireless short range, low rate communication has been attracting growing interest due to its low power consumption and very high bandwidth. Moreover, it is able to offer accurate localization in the range of few centimeters. Recognising these qualities, it is desirable to design IR-UWB transceiver which can draw minimum possible power when it is applied for battery driven wireless sensors. For this reason, in this paper, baseband design and performance of IR-UWB transceiver based on the standard IEEE.802.15.4a was investigated with particular emphasis on reducing power consumption. Synchronization algorithm that achieves 16 ns is presented and its performance is very promising offering nearly 100% synchronization for as low SNR as 8 dB. Different resolutions of analogue to digital converter (ADC) are investigated to find out the optimum with respect to power consumption and performance. 4 bit ADC was found to be the most optimal for the sample rate of 62.4 MHz. BER performance of pulse position modulation was evaluated under realistic channel conditions with multipath components.
Conference Paper
Ultra Wideband (UWB) has been gaining growing research and industry interests in many different areas of wireless communications. Among them using UWB as a radio interface for wireless sensor network is of high interest due to its precise localisation and low power consuming capabilities. In this paper we carry out quantitative analysis into the synchronisation performance of UWB when it operates under usual wireless channel environments. Simulation results confirm that coherent receiver needs around 10 dB less signal to noise ratio to achieve synchronisation compared to the noncoherent detection. Higher spreading factors lead to noticeable improvement for both coherent and noncoherent detection.
Conference Paper
This paper presents a fully differential baseband pulse generator intended for Impulse-Radio Ultra-Wideband (IR-UWB) direct up-conversion transmitter architectures. The generator provides impulse related binary phase shift keying (BPSK) and on/off keying (OOK) modulation in accordance with the IEEE 802.15.4a standard. The logic part of the generator runs on a clock of 499.2 MHz allowing direct generation of single preamble impulses as well as data impulse bursts. While the pulse generator is digitally controlled at the input, the output provides an analogue signal ready to be shaped by a low-pass filter (LPF) and up-converted to the desired channel.
Evaluation and optimisation of robustness in the IEEE.802.15.4a standard
  • R Olonbayar
  • C Kraemer
  • Schwingenschloegel
Olonbayar, R. Kraemer, C. Schwingenschloegel " Evaluation and optimisation of robustness in the IEEE.802.15.4a standard ", ICUWB10, Sep. 2010, Nanjing, China, pp. 747-750
IRUWB Transceiver design and performance for wireless sensors
  • S Olonbayar
  • G Fischer
  • R Kraemer
S. Olonbayar, G. Fischer, R. Kraemer "IRUWB Transceiver design and performance for wireless sensors", ICUWB09, Sep. 2009, Vancouver, Canada, pp. 809-813
Evaluation and optimisation of robustness in the IEEE.802.15.4a standard
  • Schwingenschloegel
Schwingenschloegel "Evaluation and optimisation of robustness in the IEEE.802.15.4a standard ", ICUWB10, Sep. 2010, Nanjing, China, pp. 747-750