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Performance Analysis of a Microstrip Printed
Antenna Conformed on a Cylindrical Body
for TM
10
mode Using Two Different Substrates
Ali Elrashidi
1
, Khaled Elleithy
2
, Hassan Bajwa
3
1
Department of Computer and Electrical Engineering, University of Bridgeport, Bridgeport, CT 06604, USA
(aelrashi@bridgeport.edu)
2Department of Computer and Electrical Engineering, University of Bridgeport, Bridgeport, CT 06604, USA
(elleithy@bridgeport.edu)
3Department of Computer and Electrical Engineering, University of Bridgeport, Bridgeport, CT 06604, USA
(hbjwa@bridgeport.edu)
Abstract Curvature has a great effect on fringing field of a microstrip antenna and consequently fringing field affects
effective dielectric constant and then all antenna parameters. A new mathematical model for input impedance, return loss,
voltage standing wave ratio and electric and magnetic fields is introduced in this paper. These parameters are given for
TM
10
mode and using two different substrate materials RT/duroid-5880 PTFE and K-6098 Teflon/Glass. Experimental
results for RT/duroid-5880 PTFE substrate are also introduced to validate the new model.
Keywords Fringing field, Curvature, effective dielectric constant and Return loss (S11), Voltage Standing Wave Ratio
(VSWR), Transverse Magnetic TM
10
mode.
1. Introduction
Due to the unprinted growth in wireless applications and
increasing demand of low cost solutions for RF and
microwave communication systems, the microstrip flat
antenna, has undergone tremendous growth recently.
Though the models used in analyzing microstrip structures
have been widely accepted, the effect of curvature on
dielectric constant and antenna performance has not been
studied in detail. Low profile, low weight, low cost and its
ability of conforming to curve surfaces [1], conformal
microstrip structures have also witnessed enormous growth
in the last few years. Applications of microstrip structures
include Unmanned Aerial Vehicle (UAV), planes, rocket,
radars and communication industry [2]. Some advantages
of conformal antennas over the planer microstrip structure
include, easy installation (randome not needed), capability
of embedded structure within composite aerodynamic
surfaces, better angular coverage and controlled gain,
depending upon shape [3, 4]. While Conformal Antenna
provide potential solution for many applications, it has
some drawbacks due to bedding [5]. Such drawbacks
include phase, impedance, and resonance frequency errors
due to the stretching and compression of the dielectric
material along the inner and outer surfaces of conformal
surface. Changes in the dielectric constant and material
thickness also affect the performance of the antenna.
Analysis tools for conformal arrays are not mature and fully
developed [6]. Dielectric materials suffer from cracking due
to bending and that will affect the performance of the
conformal microstrip antenna.
2. Background
Conventional microstrip antenna has a metallic patch print-
ed on a thin, grounded dielectric substrate. Although the
patch can be of any shape, rectangular patches, as shown in
Figure 1 [7], are preferred due to easy calculation and mod-
eling.
Figure 1. Rectangular microstrip antenna
Fringing fields have a great effect on the performance of a
microstrip antenna. In microstrip antennas the electric filed
in the center of the patch is zero. The radiation is due to the
fringing field between the periphery of the patch and the
ground plane. For the rectangular patch shown in the
Figure 2, there is no field variation along the width and
thickness. The amount of the fringing field is a function of
the dimensions of the patch and the height of the substrate.
Higher the substrate, the greater is the fringing field.
Due to the effect of fringing, a microstrip patch antenna
would look electrically wider compared to its physical
dimensions. As shown in Figure 2, waves travel both in
L
W
ɛ
r
y
x
L
ε
r
R
d
d
d
s
s
d
z
substrate and in the air. Thus an effective dielectric constant
ε
reff
is to be introduced. The effective dielectric constant ε
reff
takes in account both the fringing and the wave propagation
in the line.
Figure 2. Electric field lines (Side View).
The expression for the effective dielectric constant is
introduced by A. Balanis [7], as shown in Equation 1.
(1)
The length of the patch is extended on each end by ΔL is a
function of effective dielectric constant and the
width to height ratio (W/h). ΔL can be calculated according
to a practical approximate relation for the normalized
extension of the length [8], as in Equation 2.
(2)
Figure 3. Physical and effective lengths of rectangular microstrip patch.
The effective length of the patch is L
eff
and can be
calculated as in Equation 3.
L
eff
= L+2ΔL (3)
By using the effective dielectric constant (Equation 1) and
effective length (Equation 3), we can calculate the
resonance frequency of the antenna f and all the microstrip
antenna parameters.
Cylindrical-Rectangular Patch Antenna
All the previous work for a conformal rectangular
microstrip antenna assumed that the curvature does not
affect the effective dielectric constant and the extension on
the length. The effect of curvature on the resonant
frequency has been presented previously [9]. In this paper
we present the effect of fringing field on the performance of
a conformal patch antenna. A mathematical model that
includes the effect of curvature on fringing field and on
antenna performance is presented. The cylindrical
rectangular patch is the most famous and popular conformal
antenna. The manufacturing of this antenna is easy with
respect to spherical and conical antennas.
Effect of curvature of conformal antenna on resonant
frequency been presented by Clifford M. Krowne [9, 10] as:
(4)
Where 2b is a length of the patch antenna, a is a radius of
the cylinder, 2θ is the angle bounded the width of the patch,
ε represents electric permittivity and µ is the magnetic
permeability as shown in Figure 4.
Figurer 4. Geometry of cylindrical-rectangular patch antenna[9]
Joseph A. et al, presented an approach to the analysis of
microstrip antennas on cylindrical surface. In this approach,
the field in terms of surface current is calculated, while
considering dielectric layer around the cylindrical body. The
assumption is only valid if radiation is smaller than stored
energy[11]. Kwai et al. [12]gave a brief analysis of a thin
cylindrical-rectangular microstrip patch antenna which
includes resonant frequencies, radiation patterns, input
impedances and Q factors. The effect of curvature on the
characteristics of TM
10
and TM
01
modes is also presented in
Kwai et al. paper. The authors first obtained the electric
field under the curved patch using the cavity model and
then calculated the far field by considering the equivalent
magnetic current radiating in the presence of cylindrical
surface. The cavity model, used for the analysis is only
valid for a very thin dielectric. Also, for much small
thickness than a wavelength and the radius of curvature,
only TM modes are assumed to exist. In order to calculate
the radiation patterns of cylindrical-rectangular patch
antenna. The authors introduced the exact Green’s function
approach. Using Equation (4), they obtained expressions for
the far zone electric field components E
θ
and E
φ
as a
functions of Hankel function of the second kind H
p
(2)
. The
input impedance and Q factors are also calculated under the
same conditions.
Based on cavity model, microstrip conformal antenna on a
projectile for GPS (Global Positioning System) device is
designed and implemented by using perturbation theory is
h
W
ΔL
L
ΔL
introduced by Sun L., Zhu J., Zhang H. and Peng X [13].
The designed antenna is emulated and analyzed by IE3D
software. The emulated results showed that the antenna
could provide excellent circular hemisphere beam, better
wide-angle circular polarization and better impedance
match peculiarity.
Nickolai Zhelev introduced a design of a small conformal
microstrip GPS patch antenna [14]. A cavity model and
transmission line model are used to find the initial
dimensions of the antenna and then electromagnetic
simulation of the antenna model using software called
FEKO is applied. The antenna is experimentally tested and
the author compared the result with the software results.
It was founded that the resonance frequency of the
conformal antenna is shifted toward higher frequencies
compared to the flat one.
The effect of curvature on a fringing field and on the
resonance frequency of the microstrip printed antenna is
studied in [15]. Also, the effect of curvature on the
performance of a microstrip antenna as a function of
temperature for TM
01
and TM
10
is introduced in [16], [17].
3. General Expressions for Electric and
Magnetic Fields Intensities
In this section, we will introduce the general expressions of
electric and magnetic field intensities for a microstrip
antenna printed on a cylindrical body represented in
cylindrical coordinates.
Starting from Maxwell’s Equations, we can get the relation
between electric field intensity E and magnetic flux density
B as known by Faraday’s law [18], as shown in
Equation (2):
(2)
Magnetic field intensity H and electric flux density D are
related by Ampérés law as in Equation (3):
(3)
where J is the electric current density.
The magnetic flux density B and electric flux density D as a
function of time t can be written as in Equation (4):
and (4)
where μ is the magnetic permeability and ɛ is the electric
permittivity.
By substituting Equation (4) in Equations (2) and (3), we
can get:
and (5)
where ω is the angular frequency and has the form of:
. In homogeneous medium, the divergence of
Equation (2) is:
and (6)
From Equation (5), we can get Equation (7):
or
(7)
Using the fact that, any curl free vector is the gradient of the
same scalar, hence:
(8)
where φ is the electric scalar potential.
By letting:
where A is the magnetic vector potential.
So, the Helmholtz Equation takes the form of (9):
A+ -J (9)
k is the wave number and has the form of: , and
is Laplacian operator. The solutions of Helmholtz
Equation are called wave potentials:
(10)
3.1 Near Field Equations
By using the Equations number (10) and magnetic vector
potential in [19], we can get the near electric and magnetic
fields as shown below:
(12)
E
φ
and E
ρ
are also getting using Equation (7);
(13)
(14)
To get the magnetic field in all directions, we can use the
second part of Equation (10) as shown below, where H
z
= 0
for TM mode:
(15)
(16)
3.2 Far Field Equations
In case of far field, we need to represent the electric and
magnetic field in terms of r, where r is the distance from the
center to the point that we need to calculate the field on it.
By using the cylindrical coordinate Equations, one can
notice that a far field ρ tends to infinity when r, in Cartesian
coordinate, tends to infinity. Also, using simple vector
analysis, one can note that, the value of k
z
will equal to
[19], and from the characteristics of Hankel
function, we can rewrite the magnetic vector potential
illustrated in Equation (12) to take the form of far field as
illustrated in Equation (17).
(17)
Hence, the electric and magnetic field can easily be
calculated as shown below:
(18)
(19)
(20)
The magnetic field intensity also obtained as shown below,
where H
z
= 0:
(21)
(22)
4. Input Impedance
The input impedance is defined as “the impedance presented
by an antenna at its terminals” or “the ratio of the voltage
current at a pair of terminals” or “the ratio of the appropriate
components of the electric to magnetic fields at a point”. The
input impedance is a function of the feeding position as we
will see in the next few lines.
To get an expression of input impedance Z
in
for the
cylindrical microstrip antenna, we need to get the electric
field at the surface of the patch. In this case, we can get the
wave equation as a function of excitation current density J as
follow:
(23)
By solving this Equation, the electric field at the surface can
be expressed in terms of various modes of the cavity as [15]:
(24)
where A
nm
is the amplitude coefficients corresponding to the
field modes. By applying boundary conditions,
homogeneous wave Equation and normalized conditions
for , we can get an expression for as shown below:
1. vanishes at the both edges for the length L:
(25)
2. vanishes at the both edges for the width W:
(26)
3. should satisfy the homogeneous wave
Equation :
(27)
4. should satisfy the normalized condition:
(28)
Hence, the solution of will take the form shown below:
(29)
with
The coefficient A
mn
is determined by the excitation current.
For this, substitute Equation (29) into Equation (23) and
multiply both sides of (23) by , and integrate over area
of the patch. Making use of orthonormal properties of ,
one obtains:
(30)
Now, let the coaxial feed as a rectangular current source
with equivalent cross-sectional area centered
at , so, the current density will satisfy the Equation
below:
(31)
Use of Equation (31) in (30) gives:
(32)
So, to get the input impedance, one can substitute in the
following Equation:
(33)
where is the RF voltage at the feed point and defined as:
(34)
By using Equations (24), (29), (32), (34) and substitute in
(33), we can obtain the input impedance for a rectangular
microstrip antenna conformal in a cylindrical body as in the
following Equation:
(35)
5. Voltage Standing Wave Ratio and
Return Loss
Voltage Standing Wave Ration VSWR is defined as the
ration of the maximum to minimum voltage of the antenna.
The reflection coefficient ρ define as a ration between
incident wave amplitude V
i
and reflected voltage wave
amplitude V
r
, and by using the definition of a voltage
reflection coefficient at the input terminals of the antenna Γ,
as shown below:
(36)
where, Z
0
is the characteristic impedance of the antenna. If
the Equation is solved for the reflection coefficient, it is
found that, where the reflection coefficient ρ is the absolute
vale of the magnitude of Γ,
(37)
Consequently,
(38)
The characteristic can be calculated as in [14],
(39)
where : L is the inductance of the antenna, and C is the
capacitance and can be calculated as follow:
(40)
(41)
Hence, we can get the characteristic impedance as shown
below:
(42)
The return loss s
11
is related through the following
Equation:
(43)
6. Results
For the range of GHz, the dominant mode is TM
10
for
h<<W which is the case. Also, for the antenna operates at
the ranges 2.49 and 2.23 GHz for two different substrates
we can use the following dimensions; the original length is
41.5 cm, the width is 50 cm and for different lossy substrate
we can get the effect of curvature on the effective dielectric
constant and the resonance frequency.
Two different substrate materials RT/duroid-5880 PTFE and
K-6098 Teflon/Glass are used for verifying the new model.
The dielectric constants for the used materials are 2.2 and
2.5 respectively with a tangent loss 0.0015 and 0.002
respectively.
6.1 RT/duroid-5880 PTFE Substrate
The mathematical and experimental results for input
impedance, real and imaginary parts for a different radius of
curvatures are shown in Figures 5 and 6. The peak value of
the real part of input impedance is almost 600 Ω at
frequency 2.49 GHz which gives a zero value for the
imaginary part of input impedance as shown in Figure 6 at
20 mm radius of curvature. The value 2.49 GHz represents
a resonance frequency for the antenna at 20 mm radius of
curvature.
VSWR is given in Figure 7. It is noted that, the value of
VSWR is almost 1.4 at frequency 2.49 GHz which is very
efficient in manufacturing process. It should be between 1
and 2 for radius of curvature 20 mm. The minimum VSWR
we can get, the better performance we can obtain as shown
clearly from the definition of VSWR.
Return loss (S11) is illustrated in Figure 8. We obtain a very
low return loss, -12 dB, at frequency 2.49 GHz for radius of
curvature 20 mm.
Figure 5. Mathimatical and experimental real part of the input impedance
as a function of frequency for different radius of curvatures.
Normalized electric field for different radius of curvatures
is illustrated in Figure 9. Normalized electric field is plotted
for θ from zero to 2π and φ equal to zero. As the radius of
curvature is decreasing, the radiated electric field is getting
wider, so electric field at 20 mm radius of curvature is wider
than 65 mm and 65 mm is wider than flat antenna. Electric
field strength is increasing with decreasing the radius of
curvature, because a magnitude value of the electric field is
depending on the effective dielectric constant and the
effective dielectric constant depending on the radius of
curvature which decreases with increasing the radius of
curvature.
Normalized magnetic field is wider than normalized electric
field, and also, it is increasing with deceasing radius of
curvature. Obtained results are at for θ from zero to 2π and
φ equal to zero and for radius of curvature 20, 65 mm and
for a flat microstrip printed antenna are shown in Figure 10.
For different radius of curvature, the resonance frequency
changes according to the change in curvature, so the given
normalized electric and magnetic fields are calculated for
different resonance frequency according to radius of
curvatures.
Figure 6. Mathimatical and experimental imaginary part of the input
impedance as a function of frequency for different radius of curvatures.
Figure 7. Mathimatical and experimental VSWR versus frequency for
different radius of curvatures.
Figure 8. Mathimatical and experimental return loss (S11) as a function of
frequency for different radius of curvatures.
Figure 9. Normalized electric field for radius of curvatures 20, 65 mm and
a flat antenna at θ=0:2π and φ=0
0
.
Figure 10. Normalized magnetic field for radius of curvatures 20, 65 mm
and a flat antenna at θ=0:2π and φ=0
0
.
6.2 K-6098 Teflon/Glass Substrate
The real part of input impedance is given in Figure 11 as a
function of curvature for 20 and 65 mm radius of curvature
compared to a flat microstrip printed antenna. The peak
value of a real part of input impedance at 20 mm radius of
curvature occurs at frequency 2.235 GHz at 410 Ω
maximum value of resistance. The imaginary part of input
impedance, Figure 12, is matching with the previous result
which gives a zero value at this frequency. The resonance
frequency at 20 mm radius of curvature is 2.235 GHz,
which gives the lowest value of a VSWR, Figure 13, and
lowest value of return loss as in Figure 14. Return loss at
this frequency is -21 dB which is a very low value that leads
a good performance for a microstrip printed antenna
regardless of input impedance at this frequency.
The normalized electric field for K-6098 Teflon/Glass
substrate is given in Figure 15 at different radius of
curvatures 20, 65 mm and for a flat microstrip printed
antenna.
Normalized electric field is calculated at θ equal to values
from zero to 2π and φ equal to zero. At radius of curvature
20 mm, the radiation pattern of normalized electric field is
wider than 65 mm and flat antenna, radiation pattern angle
is almost 120
0
, and gives a high value of electric field
strength due to effective dielectric constant.
The normalized magnetic field is given in Figure 16, for the
same conditions of normalized electric field. Normalized
magnetic field is wider than normalized electric field for
20 mm radius of curvature; it is almost 170
0
for 20 mm
radius of curvature. So, for normalized electric and
magnetic fields, the angle of transmission is increased as a
radius of curvature decreased.
Figure 11. Real part of the input impedance as a function of frequency for
different radius of curvatures.
Figure 12. Imaginary part of the input impedance as a function of
frequency for different radius of curvatures.
Figure 13. VSWR versus frequency for different radius of curvatures.
Figure 14. Return loss (S11) as a function of frequency for different radius
of curvatures.
Figure 15. Normalized electric field for radius of curvatures 20, 65 mm
and a flat antenna at θ=0:2π and φ=0
0
.
Figure 16. Normalized magnetic field for radius of curvatures 20, 65 mm
and a flat antenna at θ=0:2π and φ=0
0
.
7. Conclusion
The effect of curvature on the performance of conformal
microstrip antenna on cylindrical bodies for TM
10
mode is
studied in this paper. Curvature affects the fringing field and
fringing field affects the antenna parameters. The Equations
for real and imaginary parts of input impedance, return loss,
VSWR and electric and magnetic fields as a functions of
curvature and effective dielectric constant are derived. By
using these derived equations, we introduced the results for
different dielectric conformal substrates. For the two
dielectric substrates, the decreasing in frequency due to
increasing in the curvature is the trend for all materials and
increasing the radiation pattern for electric and magnetic
fields due to increasing in curvature is easily noticed.
We conclude that, increasing the curvature leads to
increasing the effective dielectric constant, hence,
resonance frequency is increased. So, all parameters are
shifted toward increasing the frequency with increasing
curvature.
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