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60 GHz radio: Prospects and future directions


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This paper addresses the basic issues regarding the design and develop-ment of wireless systems that will operate in the 60 GHz band. The 60 GHz band is of much interest since this is the band in which a massive amount of spectral space (5 to 7 GHz) has been allocated for dense wireless local communications.
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Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
60 GHz radio: prospects and future directions
P.F.M. Smulders
Eindhoven University of Technology, Department of Electrical Engineering,
P.O. Box 513, 5600 MB Eindhoven, The Netherlands, e-mail:
AbstractThis paper addresses the basic
issues regarding the design and develop-
ment of wireless systems that will operate
in the 60 GHz band. The 60 GHz band is
of much interest since this is the band in
which a massive amount of spectral space
(5 to 7 GHz) has been allocated for dense
wireless local communications.
Keywords 60 GHz, millimetre wave band
There exists an ever increasing supply of, and demand
for, broadband multimedia applications calling for an
ever increasing capacity of wireless networks.
Finally, this will cause a demand for wireless transfer
capacity far in excess of what can be accommodated
in the currently used bands at 2.4-2.5 and 5.2-5.8
GHz [1]. An obvious solution to this problem is to
resort to the 60 GHz band, where bandwidth is
abundantly available. In particular, for dense local
communica-tions, the 60 GHz band is of special
interest because of the specific attenuation
characteristic due to atmospheric oxygen of 10 to 15
dB/km. The 10-15 dB/km regime makes the 60 GHz
band unsuitable for long-range (> 2 km)
communications so that it can be dedicated entirely to
short-range (< 1 km) communications. For the small
distances to be bridged in an indoor environment (<50
m) the 10 to 15 dB/km attenuation has no significant
impact. The specific attenuation in excess of 10
dB/km occurs in a band-width of about 8 GHz
centred around 60 GHz. Thus, from physical point of
view, there is about 8 GHz bandwidth available for
dense wireless local com-munications. This makes the
60 GHz band of utmost interest for all kinds of short-
range wireless communications.
In the United States, the Federal Communications
Commission (FCC) set aside the 59-64 GHz
frequency band for general unlicensed applications
[2]. This was the largest contiguous block of radio
spectrum ever allocated. FCC rules allow 10 Watts of
equivalent isotropic radiated power in this band,
which complies with a maximum power density of 9
µW/cm2 at 3 meters distance. This means that 20 dBm
transmit power would be the legal power limit with an
antenna having 20 dBi gain. Commercial power
amplifier GaAs MMICs are now available that can
produce 16 dBm of transmit power with good
linearity. In Japan, there was a new regulation in
August 2000 for high speed data communication. The
frequency range is 54.25-59 GHz for licensed use
with a maximum output power of 100 mW and a
minimum antenna gain of 20 dBi and 59 – 66 GHz for
unlicensed use with a maximum output power of 10
mW and a maximum antenna gain of 47 dBi. In
Europe, frequency is allocated for mobile in general
in the 59-66 GHz band. No specific recommendation
or decision has been issued yet in this mobile
frequency band. However, the 54 – 66 GHz band is
considered by CEPT as a main priority issue; in a
recommend-dation document [3] CEPT considers:
“the high-frequency re-use achievable in the oxygen-
absorption band reduces the requirement for
sophisticated frequency planning techniques and
offers the possibility of a pan-European deregulated
telecom-munications environment for various low-
power, low cost, short-range applications” and “there
is an urgent need to identify and harmonize civil
requirements in the frequency range 54 – 66 GHz”.
An important statement that has to be made here is
that Europe should follow the US and Japan by
opening a significant part of this band for unlicensed
use, because license free is an important condition for
promoting 60 GHz systems towards the market!
Currently, there is only one standard addressing the
60 GHz band and that is the IEEE 802.16 standard for
Wireless MAN which covers 10 to 66 GHz [4]. It
concerns a last-mile broadband wireless connectivity
alternative to fiber-based DSL. In the design of the
physical layer, line-of-sight (LOS) propagation was
deemed a practical necessity. Therefore, the standard
specifies a single carrier interface which is designated
“WirelessMan-SC”. This opens the door for the
creation of fixed Broadband Wireless Access, which
could provide license-free network access support to
buildings with speeds that approach those offered by
high-speed fiber optic networks, which saves
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
tremendous initial investments in the deployment of
last-mile networking technology.
Recently a paradigm shift from “60 GHz is strictly for
LOS operation” to “60 GHz may also be suitable for
non-LOS situations under certain circumstances” has
gained considerable support. As a first step in the
creation of a standard, the IEEE has formed an
interest group to explore the use of the 60 GHz band
for wireless personal area networks (WPANs), which
typically have to support non-LOS operation with a
range of 10 meters. The IEEE 802.15.3(TM)
Millimetre Wave Interest Group (mmWIG) was
formed in July 2003 as part of an effort to develop a
millimetre-wave-based alternative physical layer for
the IEEE high-rate WPAN standard, IEEE
802.15.3TM 2003.
Cost efficient RF solutions for high data rate
transmission at 60 GHz still have to be determined. In
this respect, some important choices have to be made
which might by crucial for commercial success:
choice of the 60 GH radio front-end architecture,
choice of technology in which the radio front-end
should be implemented: GaAs, InP, Si, or SiGe.
A. Front-end architecture
With respect to the choice of the architecture of the
60 GHz front-end radio there are, in principle, three
employing subsampling,
employing direct conversion (i.e., “zero RF”),
employing superheterodyning.
Employing analog-to-digital conversion (ADC) and
digital-to-analog conversion (DAC) directly at the
antennas would make the complete RF and IF part
obsolete. However, this option can be ruled out
immediately because this would require ADC and
DAC devices having 60 GHz bandwidth. Low-cost
implementation of this in the medium term will be
unfeasible. Apart from this approach, the subsampling
receiver represents the “ultimate” solution for simple
low power down conversion, which essentially
consists of a sampling switch, clocked at a much
lower frequency, and an A/D converter. The
limitations of the subsampling approach, however,
demonstrate some of the inherent problems in low-
power receiver implementations. In a subsampling
receiver, image frequencies exist at integral multiples
of the sampling rate and can alias into the band of
interest. As a result, careful filtering prior to down-
conversion is required. For example, downconversion
of an RF signal having a bandwidth of 500 MHz,
would require a sample rate of at least 1 GHz,
assuming a “brick wall filter”. In practice, the sample
rate will have to be much higher –at least 2 GHz- in
order to minimise the finite bandwidth effects of the
filter. It is questionable whether 2 GHz A/D
conversion, with let-us-say 10 bit quantization, will
become feasible in the medium term. This might be
examined, in addition to the problem that the resulting
signal-to-noise ratio of the down-sampled signal will
inevitably be poorer than that of an equivalent system
employing a mixer for downconversion, due to the
noise aliased from the bands between DC and the
passband [5].
The advantages of direct conversion are that it is
uniquely well suited to monolithic integration, due to
the lack of image filtering, and its intrinsically simple
architecture, see [6,7]. FSK modulated signals are
especially well-suited to direct conversion, due to
their low-signal energy at DC. However, the direct
conversion receiver has not gained widespread
acceptance to date, especially in high performance
wireless transceivers, due to its intrinsic sensitivity to
DC offset problems, even harmonics of the input
signal, and local oscillator leakage problems back to
the antenna. The latter problem may be considered as
the most serious problem. Offset arises from three
sources [8]:
transistor mismatch in the signal path,
LO signal leaking to the antenna because of poor
reverse isolation through the mixer and RF
amplifier, then reflecting off the antenna and self-
downconverting to DC through the mixer,
a large near-channel interferer leaking into the
LO part of the mixer, then self-down-converting
to DC.
Good circuit design may reduce these effects to a
certain extend, but cannot be eliminated completely,
particularly not if quadrature phase shift keying
(QPSK) or gaussian minimum shift keying is used
since the spectra of these schemes exhibit a peak at
DC. But when Orthogonal Frequency Division
Multiplex (OFDM) is applied there may be a solution
and that is to avoid the use of those subcarriers that,
after conversion, correspond with, or will be close to,
the DC component. This is just an example of a
possible solution but there might also be other
solutions that exploit the particularities of the 60 GHz
physical layer.
As regards the superheterodyning option, let us
consider the simple architecture as depicted in Fig. 1.
This figure shows a basic 60 GHz RF front-end
architecture for application at the portable station (PS)
end. Ideally it should be an integrated on-chip
solution consisting of a receive branch, a transmit
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
branch and a frequency generation function. The
receive branch consists of the receive antenna, a low
noise amplifier (LNA) and a mixer which
downconverts to IF. The transmit branch consists of a
mixer, a power amplifier (PA) and the transmit
antenna. The antennas are (integrated) patch antennas.
The mixers are image rejecting mixers. They need not
to be IQ mixers. The IF is taken at 5 GHz with the
idea that, with appropriate modifications, IEEE
802.11a RF can serve as IF here to allow dual mode
operation and interoperability.
Figure 1: Simple 60 GHz RF architecture
The oscillator circuit could be a voltage controlled
oscillator (VCO) controlled by an (off-chip)
frequency synthesizer. In traditional designs the VCO
is mostly implemented off-chip because it takes too
much space on the chip for providing sufficient
performance. At frequencies as high as 60 GHz it
may become, however, feasible to implement the
VCO directly on chip because the minimum
dimensions to achieve a certain performance become
much smaller. The advantage of this approach is a
reduction of components that have to be mounted on
the PCB and the avoidance of on-chip frequency
multiplication circuits, saving space on the chip and
saving VCO performance degradation due to phase
noise and frequency offset. Important: An on-chip
VCO that directly generates a reference frequency
close to 60 GHz may have a relatively relaxed
performance when compared with the requirements of
a VCO that operates on a much lower frequency in
combination with a couple of frequency multipliers.
B. Front-end technology
Clearly, the traditional discrete front-end technology
based on waveguide cannot be used for our purposes
because of weight, volume and cost. The only viable
alternative is the use of Microwave Monolitically
Integrated Circuits (MMICs). From cost point of view
it is mandatory to reduce the number of components
on the PCB board as much as possible. This reduction
of the number of RF chips to a minimum is also
important for minimising losses in chip intercom-
nection which can become easily significant at high
frequencies. So, the level of integration should be as
high as possible.
As regards the choice of semiconductor technology,
the most important performance parameters are the
transit frequency fT, i.e. the frequency at which the
current gain is one, and the maximum frequency of
oscillation fmax. It is commonly agreed that for 60
GHz RF and fT and fmax of at least 120 GHz are
required. The cheapest semiconductor technology is
based on silicon CMOS. Fig. 2 shows the
performance increase of CMOS. It shows that for
CMOS the minimum fT requirement has already been
reached but not much more than that. Silicon will
always have a natural disadvantage, when compared
to its competitors, as regards electron mobility, which
frustrates the further increase of fT and fmax.
300 GHz
100 GHz
30 GHz
10 GHz
3 GHz
1 GHz
85 87 89 91 93 95 97 99 01 03 05
Figure 2: Development of CMOS technology
A strong competitor of silicon is Gallium Arsenide
(GaAs) which is known to have the following merits:
high fT and fmax (>120 GHz), driven to still higher
values by future gigabit applications,
low noise factor (4 dB),
excellent power added efficiency (> 60%),
good linearity,
commercial 60 GHz front-end MMIC’s available
(thus proven technology for 60 GHz radio).
a good impression of the state-of-the-art performance
of 60 GHz GaAs-based MMIC chips that are
currently commercially available is given by [9] –
[11]. From these data we learn that, in principle, an
RF frond-end can be composed having considerable
performance: 16 dBm transmit power, 5 dB noise
factor and –100 dBc/Hz @ 100 kHz phase noise.
GaAs-based 60 GHz devices such as low-noise
amplifiers, high power amplifiers, multipliers and
switches can nowadays be ordered in large quantities
in die form at prices in the order of 15 € a piece [12].
According to IBM [13] silicon germanium (SiGe)
technology can compete with III-V semiconductor
performance, while simultaneously maintaining the
multitude of advantage of silicon materials:
applying wafers of defect-free silicon material
with large diameter (200 mm),
low substrate and processing costs,
IF out
(5 GHz)
IF in
(5 GHz)
60 GHz RF
60 GHz RF
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
high process uniformity,
high yield (low defect densities),
high reliability,
high design robustness,
high integration level.
To this list we can add a favourable property and that
is a better heath removal due to a higher substrate
conductivity which may become of crucial
importance since we have to cope with considerable
heath generation by the power amplifier at a very
small spot.
The current generation SiGe technology features fT
and fmax values of ~120 GHz. It can be anticipated
that fT and fmax will remain to increase up to around
200 – 250 GHz. Increase of fT and fmax is coupled with
a decrease in break-down voltage which may limit the
peak power output.
Table I lists GaAs and SiGe cost estimates according
to [14]. For heterojunction bipolar transistor
technology we can conclude that the cost of GaAs is
about 4 times the cost of SiGe on a $/mm2 basis. This
conclusion is confirmed by the cost estimation made
in [15], which reads: 0.12 $/mm2 for 6” SiGe, 0.5
$/mm2 for 4” GaAs and 1.2 $/mm2 for 3” Indium
phosfide (InP).
Table I
GaAs and SiGe cost estimates
mm200200100100Wafer Diameter
$3400300020001400Raw cost
$3200280014001200Photo cost
32281412Mask steps
$200200600200Starting Material
µm Size
In the short term GaAs as well as InP technology will
be applied since these are already mature technologies
providing excellent performance. Because of the cost
advantage, however, it is likely that SiGe technology
will become the ultimate solution for low-cost high-
volume 60 GHz front-end MMIC chips.
60 GHz antennas should feature the following
low fabrication cost, readily amenable to mass
light weight, low volume,
high efficiency which implies low-loss feed,
easily integratable with MMIC RF front-end
covering the 59 – 66 GHz frequency band,
eventually, circular polarisation.
The requirements of low cost, low weight and low
volume rule out many antenna structures: Obviously,
the classical microwave aperture antennas of the type
“heavy metal” are unsuitable. Also lens antennas
cannot be used because a lens is typically an
expensive part and is not readily amenable to mass
production. The well-known whip antenna with
coaxial feed may be considered as too lossy and too
expensive, in case it is applied at 60 GHz. A
relatively new development is the application of
micro-electromechanical systems (MEMS) in antenna
structures, i.e., the use of small (chip-level) electro-
mechanical parts that can be actuated by supplying a
certain actuator voltage. Such antenna structures have
the potential to become cheap and small. The current
state-of-the-art is, however, that MEMS antennas
cannot be used for our purposes, because of the high
(25-100 volts) actuator voltage that is required to
establish a significant movement. A related problem
is that very thin hinges are required to achieve some
flexibility of the moving part, which gives rise to
ageing. Nevertheless, for the longer term when these
problems are solved, MEMS may become of
significance. In this respect, it should be noted that
MEMS may also become significant for voltage
controlled phase shifters to steer adaptive arrays.
For the shorter term, the only viable solution that
remains is the use of microstrip antennas. Microstrip
patch antennas feature all of the properties listed in
the aforementioned list of required properties. Linear
polarisations are possible with a straight forward feed
structure. Patch antennas can also have circular
polarisation. The application of circular polarisation is
considered because there are strong indications that
channel delay spread is substantially lower in case
circular polarisation is used instead of linear
polarisation, see [16]. Feed lines and matching
networks can be fabricated simultaneously with the
antenna structure. Finally, dual-frequency and dual-
polarisation antennas can be easily made. However, in
general it can be said that microstrip antennas also
have some limitations:
a) most microstrip antennas radiate into half-space,
b) low power handling capability (~100 W),
c) narrow bandwidth and associated tolerance
d) radiation from feeds and junctions,
For our application, a) has little significance; An
access point antenna should have high gain, whereas
a portable station antenna should radiate in horizontal
direction or slightly upwards, but not downwards.
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
Limitation b) is of no significance at all because
radiated power will never exceed 100 mW or so. The
other limitations can be circumvented by taking
suitable choices for, in particular, the dielectric
constant r
and the thickness of the substrate. For 60
GHz, a substrate tickness of 100 µm is quite
acceptable which is the typical thickness of GaAs and
SiGe chip substrate. r
for GaAs and SiGe is about
12 which yield a bandwidth of 2% and a radiation
efficiency of 80% [17]. Hence, a 60 GHz patch
antenna integrated with the 60 GHz RF front-end
might be a good option.
At 60 GHz there is much more free space loss than at
2 or 5 GHz since free space loss increases
quadratically with frequency. In principle this higher
free space loss can be compensated by the use of
antennas with more pattern directivity while
maintaining small antenna dimensions. When such
antennas are used, however, antenna obstruction, e.g.
by a human body, and mispointing may easily cause a
substantial drop of received power which may nullify
the gain provided by the antennas. This effect is
typical for millimetre waves because the diffraction of
millimetre waves (i.e., the ability to bend around
edges of obstacles) is only weak. As regards blocking
effects omnidirectional antennas have an advantage in
a reflective (e.g. indoor) environment since there they
have the ability to still collect contributions of
reflected power at the event of line-of-sight (LOS)
walls may considerably attenuate millimetre waves.
The transmissivity strongly depends on material
properties and thickness. At 60 GHz transmissivity of
glass may range from 3 to 7 dB whereas transmission
through a 15 cm thick concrete wall can be as high as
36 dB [18]. We may therefore expect that concrete
floors between stocks of a building act as reliable cell
boundaries. This helps to create small indoor cells for
hot spot communications.
Measured data for wall transmission loss at 60
GHz have been reported in [19], see Table II. This
table also includes corresponding figures measured at
2.5 GHz in order to give an impression of the
difference with a well known reference.
The figures in the table confirm the
presumption that the transmission loss at 60 GHz is
relatively high, which implies a more favourable
frequency reuse distance. A typical/moderate inner
wall consisting of multiple partitions of different
materials (e.g., windows and doors), on the other
hand, may neither be considered as a reliable cell
boundary nor as a transparent medium. Owing to the
possible significant attenuation of inner walls it will
be generally necessary to have at least one access
point per indoor environment (room, hall, corridor
etc.) to create a reliable shared medium.
Table II
Transmission loss at 60 and 2.5 GHz (dB)
7.710.20.3Mesh glass glass
2.5 GHz60 GHzThickness (cm)
A consequence of the confinement to smaller cells is
that channel dispersion is smaller when compared
with values encountered at lower frequencies because
echo paths are shorter on average. Rms delay spread
may range from a few to 100 ns. It is expected to be
highest if omnidirectional antennas are used in large
reflective indoor environments [18,20]. When,
instead, high gain antennas are used, rms delay spread
may be limited to a few ns only [18,21].
Movements of the portable station as well as
movements of objects in the environment cause
Doppler effects as frequency shift and spectrum
broadening of the received signal. These Doppler
effects are relatively severe at 60 GHz because they
are proportional with frequency. If persons move at a
speed of 1.5 m/s (walking speed) then the Doppler
spread that results at 60 GHz is 1200 Hz [18].
A consequence of the low wall penetration of
millimetre waves is that, in many cases, at least one
access point per indoor environment is required. From
a coverage point of view the best place for the access
point antenna would be somewhere near the centre of
the room at a high position near the ceiling. From a
network deployment viewpoint, however, the need to
mount antennas in the ceiling in each and every room
is tiresome and cables would probably have to run
over the ceiling, which would be unaesthetical.
An attractive alternative option would be the
possibility of placing a small sized access point in
each room, with its small sized antenna(s) mounted
on a wall where it can be readily connected to the
existing LAN cabling that is already installed – just as
is the case with today’s WLAN access points.
In order to allow flexible terminal use, the low
position of the access point (antenna) necessitates
measures to cope with the drop in received power due
to line-of-sight (LOS) obstruction by a person or
object. One measure is to apply macro diversity by
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
switching to another access point as soon as the
received signal drops below a certain threshold.
However, this requires the use of more than one
access point per room which may increase the costs
significantly, particularly when many small rooms
have to be covered. A more attractive solution may be
found in other direction namely that of applying
particular antenna patterns, which may be adaptive to
some extent, e.g., by applying beam switching. Low-
or medium-gain antennas may be preferred to high-
gain antennas in order to avoid stringent antenna
pointing and tracking requirements. Experimental
work on 60 GHz antenna pattern optimisation has
been carried out by the Radiocommunication Group
at the Eindhoven University of Technology.
Measurements have been conducted in many indoor
environments. Fig. 3 shows the received power
normalised on the transmitted power (NRP) in dB
measured in the 58-59 GHz band as function of the
separation distance between transmitter and receiver.
These measurements have been performed in a room
with dimensions 7.2*6*3.1 m3. The sides of the room
consist of glass window and smoothly plastered
concrete walls whereas the floor is linoleum on
concrete. The ceiling consists of aluminium plates and
light holders. The transmitting antenna was located in
a corner of the room at a height of 2.5 m. This
antenna has an antenna gain of 16.5 dBi and produces
a fan-beam that is wide in azimuth and narrow in
elevation. Its beam was aiming towards the middle of
the room. A similar fan-beam antenna was applied at
the receiving station which was positioned at various
places in the room at 1.4 m above the ground.
Figure 3: Normalized received power under
LOS conditions
The upper solid curve in Fig. 3 shows the NRP in
case the beam of the receiving antenna is pointing
exactly towards the transmitting antenna. The dotted
curve represents the situation in which the fan beam
at the receiver has an azimuth pointing deviation of
35°. The lower solid curve represents the situation in
which the fan-beam antenna at the receiver is replaced
by an antenna that has an antenna gain of 6.5 dBi and
that radiates omnidirectional in the horizontal plane.
As a reference, the dashed curves are added which
represent the respective theoretical results according
to the free-space law of Friss, i.e., a 6 dB decrease per
doubling of distance. The curvature of both solid
NRP curves is typical for indoor situations in which
antenna patterns are not well pointed towards each
other at short distances. In that area, the NRP
increases with distance. This is because the increased
free space loss is more than compensated by antenna
gain since the antennas are better directed towards
each other. If the separation distance is increased
further these curves tend to become higher than the
free-space curves because the reflections from walls
etc. contribute effectively to the received power. The
dotted curve remains lower because of the fixed 35°
antenna mispointing at all distances.
All curves in Fig. 3 refer to the situation of a
LOS path between transmitter and receiver. Fig. 4
shows the curves for non-LOS (NLOS) conditions.
On applying the fan-beam antenna the average drop
of NRP due to LOS path obstruction is about 11 dB
for 0° as well as 35° pointing deviation. With the
omnidirectional antenna this drop is about 4 dB. The
results in Fig. 3 are representative for other indoor
environments in the sense that the free-space law can
be considered as a reliable lower bound of NRP at
relatively large distances. Hence, we can estimate the
feasible link performance on the basis of the free
space loss. Let us, for instance, consider the antenna
setup as described but in a larger room with an
antenna separation distance of 10 m.
Figure 4: Normalized received power under
NON-LOS conditions
The transmitted power is 10 dBm (10 mW), which is
well feasible with todays 60 GHz MMIC amplifiers
operating in their linear region. According to the Friss
formula, the received power is –55 dBm. If only
thermal noise is encountered, the noise power at the
receiver is 10logkTBF with k is Boltzmann's constant
(1.38·10-23 J/K), T is equivalent noise temperature of
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
the receiver (room temperature = 290 K), B is noise
bandwidth and F is receiver noise figure. With a
receiver noise figure of 10 dB and a noise bandwidth
of 500 MHz the received noise power amounts to –77
dBm. This yields a signal-to-noise ratio (SNR) of 22
dB. Within 500 MHz bandwidth, a data rate of 500
Mbit/s can be accommodated by using OFDM in
combination with QPSK and ¾ rate convolutional
coding. For sufficient performance in terms of bit
error ratio (< 10-6) a SNR of about 10 dB is required.
This implies that 12 dB margin is left to cope with
shadowing and performance degrading factors
occurring in the transceiver such as phase noise and
frequency shift. As shown by Fig. 3 and Fig. 4. this
margin can be improved 16.5-6.5-11+4=3 dB by
applying fan beam antennas instead of
omnidirectional antennas. In order to avoid
cumbersome pointing, a fan beam antenna can be
used in the form of a sector antenna as presented in
Now let us compare the SNR performance at 60 GHz
with what we may expect at a much lower frequency,
say 5 GHz. Since, according to the Friss formula, the
free-space path loss is proportional to the square of
the frequency, the link budget at 60 GHz is 21 dB less
when compared with the linkbudget at 5 GHz under
equal conditions (same antenna patterns, separation
distances etc.). So, at first sight, there seems to be a
substantial disadvantage of 60 GHz transmission. It
should be realised, however, that any successful
commercial system is essentially limited by co-
channel interference, which will also be 21 dB lower
at 60 GHz, as far as it concerns free-space loss. As a
matter of fact, the signal-to-interference ratio of an
interference-limited system is commonly modelled as
being independent of the operating frequency, see e.g.
[23]. If we also take into account the attenuation due
to oxygen absorption as well as the extra severe wall
attenuation at 60 GHz, we may even expect better
signal-to-interference figures.
A good figure of merit for network capacity is the
obtainable information transfer density per square
meter. As already stated in the previous section it
should be readily possible to obtain a spectral
efficiency of 1 bps/Hz with OFDM as transmission
scheme. In that case the availability of 5 GHz spectral
space implies an aggregate capacity of 5 Gbps per
cell. With a cell radius of 10 m the information
density is then 16 Mbps/m2. Table III provides a com-
parison with corresponding figures obtainable with
802.11b, Bluetooth, 802.11a/g, and Ultra Wide Band
(UWB) based on similar calculations. It reveals
that 60 GHz radio has the potential to provide an
information transfer density that is not only far in
excess of what can be obtained with current systems
but is also more than one order of magnitude higher
than what UWB can offer. Further capacity
enhancements can be relatively easily obtained by
application of Multiple-Input Multiple-Output
techniques since, at 60 GHz, antenna dimensions and
mutual distances can be relatively small.
Table III
Information density comparison
10612103Number of
5005054111Inform. rate
per channel
10105010100Cell radius (m)
60 GHzUWB802.11aBluetooth802.11b
The principal reason for focussing on the oxygen
absorption band around 60 GHz band is the huge
amount of allocated bandwidth, which can be used to
accommodate all kind of short-range (<1 km) wireless
communication. In addition, 60 GHz front-end
technology is emerging rapidly. In principle, VCO’s
as well as antennas can be implemented so small that
they can be integrated with the RF MMIC yielding a
considerable cost saving. When compared with the
well-known 2.4-2.5 GHz band the channel dispersion
appears to be relatively small. In addition, it is
confirmed that the frequency reuse distance is
relatively small.
In the United States as well as in Japan a large
contiguous block of radio spectrum has been allocated
for unlicensed use. Up to now Europe did not follow
although it is recognized by CEPT that “the high-
frequency re-use achievable in the oxygen-absorption
band reduces the requirement for sophisticated
frequency planning techniques and offers the
possibility of a pan-European deregulated telecom-
munications environment for various low-power, low-
cost, short-range applications”. The message herewith
posed to CEPT is that it would be wise to follow the
US and Japan by opening a large part of the 60 GHz
band for unlicensed use. The resulting world-wide
license-free band would be a clear incentive for the
industry to develop standards and 60 GHz radio
technology with the prospective of a new mass
Proceedings Symposium IEEE Benelux Chapter on Communications and Vehicular Technology, 2003,
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In this paper, radio-over-free-space optic (RoFSO) transmission is investigated for the next-generation wireless networks. Here, the 60 GHz based radio frequency (RF) system is demonstrated using 16-quadrature amplitude modulation with orthogonal frequency division multiplexing scheme. System performance for FSO link is evaluated by modeling the channel with the gamma-gamma fading. The optical spectrum and constellation diagram of the received signals are analyzed. The improvement reported for error vector magnitude is 3.8–0.6% and symbol error rate of 0.03–0.005 for free-space range of 0.2–1 km. Proposed work claims that orthogonal frequency division multiplexing can be helpful for the FSO systems by providing robustness to adverse atmospheric turbulence.
... But, bandwidth demands are still rising. In such a context, 60 GHz millimeter wave (MMW) systems constitute a very attractive solution, due to the fact that there is a several GHz unlicensed frequency range available around 60 GHz, [6] almost worldwide. In Europe the frequency ranges 62-63 GHz and 65-66 GHz are reserved for wideband mobile networks MBS, (Mobile Broadband System), whereas 59-62 GHz range is reserved for wideband wireless local area networks (WLAN). ...
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