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Hindawi Publishing Corporation
EURASIP Journal on Wireless Communications and Networking
Volume 2007, Article ID 60654, 11 pages
doi:10.1155/2007/60654
Research Article
Transmit Diversity at the Cell Border Using
Smart Base Stations
Simon Plass, Ronald Raulefs, and Armin Dammann
German Aerospace Center (DLR), Institute of Communications and Navigation, Oberpfaffenhofen, 82234 Wessling, Germany
Received 27 October 2006; Revised 1 June 2007; Accepted 22 October 2007
Recommended by A. Alexiou
We address the problems at the most critical area in a cellular multicarrier code division multiple access (MC-CDMA) network,
namely, the cell border. At a mobile terminal the diversity can be increased by using transmit diversity techniques such as cyclic
delay diversity (CDD) and space-time coding like Alamouti. We transfer these transmit diversity techniques to a cellular environ-
ment. Therefore, the performance is enhanced at the cell border, intercellular interference is avoided, and soft handover procedures
are simplified all together. By this, macrodiversity concepts are exchanged by transmit diversity concepts. These concepts also shift
parts of the complexity from the mobile terminal to smart base stations.
Copyright © 2007 Simon Plass et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
1. INTRODUCTION
The development of future mobile communications systems
follows the strategies to support a single ubiquitous radio ac-
cess system adaptable to a comprehensive range of mobile
communication scenarios. Within the framework of a global
research effort on the design of a next generation mobile sys-
tem, the European IST project WINNER—Wireless World
Initiative New Radio—[1] is also focusing on the identifica-
tion, assessment, and comparison of strategies for reducing
and handling intercellular interference at the cell border. For
achieving high spectral efficiency the goal of future wireless
communications systems is a total frequency reuse in each
cell. This leads to a very critical area around the cell borders.
Since the cell border area is influenced by at least two
neighboring base stations (BSs), the desired mobile termi-
nal (MT) in this area has to scope with several signals in
parallel. On the one hand, the MT can cancel the interfer-
ing signals with a high signal processing effort to recover the
desired signal [2]. On the other hand, the network can man-
age the neighboring BSs to avoid or reduce the negative in-
fluence of the transmitted signals at the cell border. Due to
the restricted power and processing conditions at the MT, a
network-based strategy is preferred.
In the region of overlapping cells, handover procedures
exist. Soft handover concepts [3] have shown that the usage
of two base stations at the same time increases the robust-
ness of the received data and avoids interruption and calling
resources for reinitiating a call. With additional information
about the rough position of the MT, the network can avoid
fast consecutive handovers that consume many resources, for
example, the MT moves in a zigzag manner along the cell
border.
Already in the recent third generation mobile commu-
nications system, for example, UMTS, macrodiversity tech-
niques with two or more base stations are used to provide
reliable handover procedures [4]. Future system designs will
take into account the advanced transmit diversity techniques
that have been developed in the recent years. As the cell sizes
decrease further, for example, due to higher carrier frequen-
cies, the cellular context gets more dominant as users switch
cellsmorefrequently.Theubiquitousapproachofhavinga
reliable link everywhere emphasizes the need for a reliable
connection at cell border areas.
A simple transmit diversity technique is to combat flat
fading conditions by retransmitting the same signal from
spatially separated antennas with a frequency or time off-
set. The frequency or time offset converts the spatial diver-
sity into frequency or time diversity. The effective increase
of the number of multipaths is exploited by the forward er-
ror correction (FEC) in a multicarrier system. The elemen-
tary method, namely, delay diversity (DD), transmits delayed
replicas of a signal from several transmit (TX) antennas [5].
The drawback are increased delays of the impinging signals.
By using the DD principle in a cyclic prefix-based system, in-
tersymbol interference (ISI) can occur due to too large delays.
2 EURASIP Journal on Wireless Communications and Networking
This can be circumvented by using cyclic delays which results
in the cyclic delay diversity (CDD) technique [6].
Space-time block codes (STBCs) from orthogonal de-
signs [7] improve the performance in a flat and frequency
selective fading channel by coherently adding the signals at
the receiver without the need for multiple receive anten-
nas. The number of transmit antennas increases the perfor-
mance at the expense of a rate loss. The rate loss could be
reduced by applying nearly orthogonal STBCs which on the
other hand would require a more complex space-time de-
coder. Generally, STBCs of orthogonal or nearly orthogonal
designs need additional channel estimation, which increases
the complexity.
The main approach of this paper is the use and inves-
tigation of transmit diversity techniques in a cellular envi-
ronment to achieve macrodiversity in the critical cell border
area. Therefore, we introduce cellular CDD (C-CDD) which
applies the CDD scheme to neighboring BSs. Also the Alam-
outi scheme is addressed to two BSs [8] and in the follow-
ing this scheme is called cellular Alamouti technique (CAT).
The obtained macrodiversity can be utilized for handover de-
mands, for example.
Proposals for a next generation mobile communications
system design favor a multicarrier transmission, namely,
OFDM [9]. It offers simple digital realization due to the fast
Fourier transformation (FFT) operation and low complexity
receivers. The WINNER project aims at a generalized multi-
carrier (GMC) [10] concept which is based on a high flexible
packet-oriented data transmission. The resource allocation
within a frame is given by time-frequency units, so called
chunks. The chunks are preassigned to different classes of
data flows and transmission schemes. They are then used in a
flexible way to optimize the transmission performance [11].
One proposed transmission scheme within GMC is the
multicarrier code division multiple access (MC-CDMA).
MC-CDMA combines the benefits of multicarrier transmis-
sion and spread spectrum and was simultaneously proposed
in 1993 by Fazel and Papke [12] and Yee et al. [13]. In ad-
dition to OFDM, spread spectrum, namely, code division
multiple access (CDMA), gives high flexibility due to simul-
taneous access of users, robustness, and frequency diversity
gains [14].
In this paper, the proposed techniques C-CDD and CAT
are applied to a cellular environment based on an MC-
CDMA transmission scheme. The structure of the paper is
as follows. Section 2 describes the used cellular multicarrier
system based on MC-CDMA. Section 3 introduces the cellu-
lar transmit diversity technique based on CDD and the ap-
plication of the Alamouti scheme to a cellular environment.
At the end of this section both techniques are compared and
the differences are highlighted. A more detailed analytical in-
vestigation regarding the influence of the MT position for the
C-CDD is given in Section 4. Finally, the proposed schemes
are evaluated in Section 5.
2. CELLULAR MULTICARRIER SYSTEM
In this section, we first give an outline of the used MC-
CDMA downlink system. We then describe the settings of the
cellular environment and the used channel model.
2.1. MC-CDMA system
The block diagram of a transmitter using MC-CDMA is
shown in Figure 1. The information bit streams of the Nu
active users are convolutionally encoded and interleaved by
the outer interleaver Πout. With respect to the modulation
alphabet, the bits are mapped to complex-valued data sym-
bols. In the subcarrier allocation block, Ndsymbols per user
are arranged for each OFDM symbol. The kth data symbol
is multiplied by a user-specific orthogonal Walsh-Hadamard
spreading code which provides chips. The spreading length
Lcorresponds to the maximum number of active users L=
Nu,max. The ratio of the number of active users to Nu,max rep-
resents the resource load (RL) of an MC-CDMA system.
An inner random subcarrier interleaver Πin allows a bet-
ter exploitation of diversity. The input block of the inter-
leaver is denoted as one OFDM symbol and NsOFDM sym-
bols describe one OFDM frame. By taking into account a
whole OFDM frame, a two-dimensional (2D) interleaving
in frequency and time direction is possible. Also an inter-
leaving over one dimension (1D), the frequency direction,
is practicable by using one by one OFDM symbols. These
complex valued symbols are transformed into time domain
by the OFDM entity using an inverse fast Fourier transform
(IFFT). This results in NFFT time domain OFDM symbols,
represented by the samples
x(n)
l=1
NFFT
NFFT−1
i=0
X(n)
i·ej(2π/NFFT )il,(1)
where l,idenote the discrete time and frequency and nthe
transmitting BS out of NBS BSs. A cyclic prefix as a guard
interval (GI) is inserted in order to combat intersymbol in-
terference (ISI). We assume quasistatic channel fading pro-
cesses, that is, the fading is constant for the duration of one
OFDM symbol. With this quasistatic channel assumption the
well-known description of OFDM in the frequency domain
is given by the multiplication of the transmitted data symbol
X(n)
l,iand a complex channel transfer function (CTF) value
H(n)
l,i. Therefore, on the receiver side the lth received MC-
CDMA symbol at subcarrier ibecomes
Yl,i=
NBS−1
n=0
X(n)
l,iH(n)
l,i+Nl,i(2)
with Nl,ias an additive white Gaussian noise (AWGN) pro-
cess with zero mean and variance σ2, the transmitter signal
processing is inverted at the receiver which is illustrated in
Figure 2. In MC-CDMA the distortion due to the flat fading
on each subchannel is compensated by equalization. The re-
ceived chips are equalized by using a low complex linear min-
imum mean square error (MMSE) one-tap equalizer. The re-
sulting MMSE equalizer coefficients are
Gl,i=H(n)∗
l,i
H(n)
l,i
2+L/Nuσ2,i=1, ...,Nc.(3)
Furthermore, Ncis the total number of subcarriers. The op-
erator (·)∗denotes the complex conjugate. Further, the sym-
bol demapper calculates the log-likelihood ratio for each bit
Simon Plass et al. 3
User 1
User NuCOD
.
.
.
COD Πout
.
.
.
Πout Map
Map
.
.
.
MUX
d(1)
1
.
.
.
d(Nu)
1
d(1)
Nd
.
.
.
d(Nu)
Nd
CL
.
.
.
CL
+
+s1
.
.
.
sNd
Πin
X(n)
l,1
X(n)
l,Nc
OFDM
D/A
x(n)(t)
Figure 1: MC-CDMA transmitter of the nth base station.
y(t)A/D
IOFDM
.
.
.
Yl,1
Yl,Nc
Π−1
in
s1
sNd
.
.
.
Eq.
Eq.
.
.
.
CH
L
CH
L
.
.
.
DMUX
Demap.
.
.
.
Demap.
Π−1
out
Π−1
out
.
.
.
DEC
DEC User 1
User Nu
Figure 2: MC-CDMA receiver.
Desired BS
d
d0MT
d1
δ1d0
Interfering BS
Figure 3: Cellular environment.
based on the selected alphabet. The code bits are deinter-
leaved and finally decoded using soft-decision Viterbi decod-
ing [15].
2.2. Cellular environment
We consider a synchronized cellular system in time and fre-
quency with two cells throughout the paper, see Figure 3.The
nth BS has a distance dnto the desired MT. A propagation
loss model is assumed to calculate the received signal energy.
Thesignalenergyattenuationduetopathlossisgenerally
modeled as the product of the γth power of distance and a
log-normal component representing shadowing losses. The
propagation loss normalized to the cell radius ris defined by
αdn=dn
r−γ
·10η/10 dB,(4)
where the standard deviation of the Gaussian-distributed
shadowing factor ηis set to 8 dB. The superimposed signal
at the MT is given by
Yl,i=X(0)
l,iαd0H(0)
l,i+X(1)
l,iαd1H(1)
l,i+Nl,i
=S(0)
l,i+S(1)
l,i+Nl,i.(5)
Depending on the position of the MT the carrier-to-
interference ratio (C/I) varies and is defined by
C
I=E
S(0)
l,i
2
E
S(1)
l,i
2.(6)
3. TRANSMIT DIVERSITY TECHNIQUES FOR
CELLULAR ENVIRONMENT
In a cellular network the MT switches the corresponding BS
when it is requested by the BS. The switch is defined as the
handover procedure from one BS to another. The handover
is seamless and soft when the MT is connected to both BSs at
the same time. The subcarrier resources in an MC-CDMA
system within a spreading block are allocated to different
users. Some users might not need a handover as they are
(a) in a stable position or (b) away from the cell border. In
both cases these users are effected by intercell interference
as their resource is also allocated in the neighboring cell. To
separate the different demands of the users, users with sim-
ilar demands are combined within time-frequency units, for
example, chunks, in an OFDM frame. The requested param-
eters of the users combined in these chunks are similar, like a
common pilot grid. The spectrum for the users could then
be shared between two cells within a chunk by defining a
broadcast region. By this the affected users of the two cells
would reduce their effective spectrum in half. This would be
a price to pay avoiding intercellular interference. Intercellu-
lar interference could be tackled by intercellular interference
cancellation techniques at complexity costs for all mobile
users. Smart BSs could in addition try to balance the needed
transmit power by risking an increase of intercellular inter-
ference also in neighboring cells. The approach presented in
the following avoids intercellular interference by defining the
effected area as a broadcast region and applying transmit di-
versity schemes for a cellular system, like cyclic delay diver-
sity and STBCs. Part of the ineluctable loss of spectrum ef-
ficiency are compensated by exploiting additional diversity
gains on the physical layer, avoiding the need of high com-
plex intercellular cancellation techniques and decreasing the
overall intercellular interference in the cellular network for
the common good.
In the following, two transmit diversity techniques are
in the focus. The first is based on the cyclic delay diversity
(CDD) technique which increases the frequency diversity of
the received signal and requires no change at the receiver to
4 EURASIP Journal on Wireless Communications and Networking
··· IFFT 1/√M
Front end of a transmitter
Cyclic
prefix
Cyclic
prefix
Cyclic
prefix
δcyc
1
δcyc
M−1.
.
.
Cyclic delay diversity extension
Figure 4: Principle of cyclic delay diversity.
exploit the diversity. The other technique applies the Alam-
outi scheme which flattens the frequency selectivity of the re-
ceived signal and requires an additional decoding process at
the mobile.
3.1. Cellular cyclic delay diversity (C-CDD)
The concept of cyclic delay diversity to a multicarrier-based
system, that is, MC-CDMA, is briefly introduced in this sec-
tion. Later on, the CDD concept will lead to an application
to a cellular environment, namely, cellular CDD (C-CDD). A
detailed description of CDD can be found in [16]. The idea
of CDD is to increase the frequency selectivity, that is, to de-
crease the coherence bandwidth of the system. The additional
diversity is exploited by the FEC and for MC-CDMA also by
the spreading code. This will lead to a better error perfor-
mance in a cyclic prefix-based system. The CDD principle is
shown in Figure 4. An OFDM modulated signal is transmit-
ted over Mantennas, whereas the particular signals only dif-
fer in an antenna specific cyclic shift δcyc
m. MC-CDMA modu-
lated signals are obtained from a precedent coding, modula-
tion, spreading, and framing part; see also Section 2.1.Before
inserting a cyclic prefix as guard interval, the time domain
OFDM symbol (cf. (1)) is shifted cyclically, which results in
the signal
xl−δcyc
mmod NFFT =1
NFFT
NFFT−1
i=0
e−j(2π/NFFT )iδcyc
m·Xi·ej(2π/NFFT )il .
(7)
The antenna specific TX-signal is given by
x(m)
l=1
√M·xl−δcyc
mmod NFFT ,(8)
where the signal is normalized by 1/√Mto keep the average
transmission power independent of the number of transmit
antennas. The time domain signal including the guard inter-
valisobtainedforl=−NGI,...,NFFT −1. To avoid ISI, the
guard interval length NGI has to be larger than the maximum
channel delay τmax. Since CDD is done before the guard in-
terval insertion in the OFDM symbol, CDD does not increase
the τmax in the sense of ISI occurrence. Therefore, the length
of the guard interval for CDD does not depend on the cyclic
delays δcyc
m,whereδcyc
misgiveninsamples.
On the receiver side and represented in the frequency do-
main (cf. (2)), the cyclic shift can be assigned formally to the
channel transfer function, and therefore, the overall CTF
Hl,i=1
√M
M−1
m=0
e−j(2π/NFFT )δcyc
m·i·H(m)
l,i(9)
is observed. As long as the effective maximum delay τ
max of
the resulting channel
τ
max =τmax +max
mδcyc
m(10)
does not intensively exceed NGI, there is no configuration and
additional knowledge at the receiver needed. If τ
max NGI,
the pilot grid and also the channel estimation process has to
be modified [17]. For example, this can be circumvented by
using differential modulation [18].
The CDD principle can be applied in a cellular environ-
ment by using adjacent BSs. This leads to the cellular cyclic
delay diversity (C-CDD) scheme. C-CDD takes advantage
of the aforementioned resulting available resources from the
neighboring BSs. The main goal is to increase performance
by avoiding interference and increasing diversity at the most
critical areas.
For C-CDD the interfering BS also transmits a copy of
the users’ signal as the desired BS to the designated MT lo-
cated in the broadcast area. Additionally, a cyclic shift δcyc
nis
inserted to this signal, see Figure 5. Therefore, the overall de-
lay in respect to the signal of the desired BS in the cellular
system can be expressed by
δn=δdn+δcyc
n, (11)
where δ(dn) represents the natural delay of the signal de-
pending on distance dn. At the MT the received signal can
be described by
Yl,i=X(0)
l,iαd0H(0)
l,ie−j(2π/NFFT)δ0·i+αd1H(1)
l,ie−j(2π/NFFT)δ1·i.
(12)
The transmission from the BSs must ensure that the recep-
tion of both signals are within the guard interval. Further-
more, at the MT the superimposed statistical independent
Rayleigh distributed channel coefficients from the different
BSs sum up again in a Rayleigh distributed channel coeffi-
cient. The usage of cyclic shifts prevents the occurrence of ad-
ditional ISI. For C-CDD no additional configurations at the
MT for exploiting the increased transmit diversity are neces-
sary.
Finally, the C-CDD technique inherently provides an-
other transmit diversity technique. If no cyclic shift δcyc
nis in-
troduced, the signals from the different BSs may arrive at the
desired MT with different delays δ(dn). These delays can be
also seen as delay diversity (DD) [5] for the transmitted MC-
CDMA signal or as macrodiversity [19] at the MT. Therefore,
an inherent transmit diversity, namely, cellular delay diver-
sity (C-DD), is introduced if the adjacent BSs just transmit
the same desired signal at the same time to the designated
MT. The C-CDD techniques can be also easily extended to
more than 2 BSs.
Simon Plass et al. 5
Desired cell
··· IFFT GI d0
2r
d0d1
δ1
GI−1FFT ···
Mobile terminal
GI δcyc
1IFFT ···
Interfering cell
Figure 5: Cellular MC-CDMA system with cellular cyclic delay diversity (C-CDD).
3.2. Cellular Alamouti technique (CAT)
In this section, we introduce the concept of transmit diversity
by using the space-time block codes (STBCs) from orthogo-
nal designs [7], namely, the Alamouti technique. We apply
this scheme to the aforementioned cellular scenario. These
STBCs are based on the theory of (generalized) orthogonal
designs for both real- and complex-valued signal constella-
tions. The complex-valued STBCs can be described by a ma-
trix
B=
←space →
⎛
⎜
⎜
⎝
b0,0 ··· b0,NBS−1
.
.
.....
.
.
bl−1,0 ··· bl−1,NBS−1
⎞
⎟
⎟
⎠↑
time
↓
, (13)
where land NBS are the STBC length and the number of BS
(we assume a single TX-antenna for each BS), respectively.
The simplest case is the Alamouti code [20],
B=x0x1
−x∗
1x∗
0.(14)
The respective assignment for the Alamouti-STBC to the kth
block of chips containing data from one or more users is ob-
tained:
y(k)=y(k)
0
y(k)∗
1
=h(0,k)h(1,k)
h(1,k)∗−h(0,k)∗·x0
x1+n(k)
0
n(k)∗
1.
(15)
y(k)is obtained from the received complex values y(k)
ior their
conjugate complex y(k)∗
iat the receiver. At the receiver, the
vector
y(k)is multiplied from left by the Hermitian of matrix
H(k). The fading between the different fading coefficients is
assumed to be quasistatic. We obtain the (weighted) STBC
information symbols
x=H(k)H·
y(k)=H(k)H·H(k)
x+H(k)H·
n(k)
=H(k)H·
n(k)+
x·1
i=0
h(i,k)
2,(16)
corrupted by noise. For STBCs from orthogonal designs,
MIMO channel estimation at the receiver is mandatory, that
is, h(n,k),n=0, ...,NBS −1, k=0, ...,K−1, must be
MC-CDMA
symbols of BS 0
Time
Other active users BS 0Desired chunk
.
.
..
.
.
X0,1
X0,0
−X∗
1,1
−X∗
1,0
MT 0 MT 1
MC-CDMA
symbols of BS 1
Time
Other active users BS 1Desired chunk
.
.
..
.
.
X0,1
X0,0
X∗
1,1
X∗
1,0
Figure 6: MC-CDMA symbol design for CAT for 2 MTs.
estimated. Disjoint pilot symbol sets for the TX-antenna
branches can guarantee a separate channel estimation for
each BS [8]. Since the correlation of the subcarrier fading
coefficients in time direction is decreasing with increasing
Doppler spread—that is, the quasistationarity assumption of
the fading is incrementally violated—the performance of this
STBC class will suffer from higher Doppler frequencies. Later
we will see that this is not necessarily true as the stationarity
of the fading could also be detrimental in case of burst errors
in fading channels.
Figure 6 shows two mobile users sojourning at the cell
borders. Both users data is spread within one spreading block
and transmitted by the cellular Alamouti technique using
two base stations. The base stations exploit information from
a feedback link that the two MTs are in a similar location in
the cellular network. By this both MTs are served simultane-
ously avoiding any interference between each other and ex-
ploiting the additional diversity gain.
3.3. R´
esum´
e for C-CDD and CAT
Radio resource management works perfectly if all informa-
tion about the mobile users, like the channel state informa-
tion, is available at the transmitter [21]. This is especially true
if the RRM could be intelligently managed by a single genie
manager. As this will be very unlikely the described schemes
C-CDD and CAT offer an improved performance especially
6 EURASIP Journal on Wireless Communications and Networking
at the critical cell border without the need of any informa-
tion about the channel state information on the transmitter
side. The main goal is to increase performance by avoiding
interference and increasing diversity at the most critical en-
vironment. In this case, the term C/I is misleading (cf. (6)),
as there is no I(interference). On the other hand, it describes
the ratio of the power from the desired base station and the
other base station. This ratio also indicates where the mo-
bile user is in respect to the base stations. For C/I =0dB
the MT is directly between the two BSs, for C/I >0 dB the
MT is closer to the desired BS, and for C/I <0 dB the MT is
closer to the adjacent BS. Since the signals of the neighbor-
ing BSs for the desired users are not seen as interference, the
MMSE equalizer coefficients of (3) need no modification as
in the intercellular interfering case [22]. Therefore, the trans-
mit diversity techniques require no knowledge about the in-
tercellular interference at the MT. By using C-CDD or CAT
the critical cell border area can be also seen as a broadcast
scenario with a multiple access channel.
For the cellular transmit diversity concepts C-CDD and
CAT, each involved BS has to transmit additionally the sig-
nal of the adjacent cell; and therefore, a higher amount of
resources are allocated at each BS. Furthermore, due to the
higher RL in each cell the multiple-access interference (MAI)
for an MC-CDMA system is increased. There will be always
atradeoffbetween the increasing MAI and the increasing di-
versity due to C-CDD or CAT.
Since the desired signal is broadcasted by more than one
BS, both schemes can reduce the transmit signal power, and
therefore, the overall intercellular interference. Using MC-
CDMA for the cellular diversity techniques the same spread-
ing code set has to be applied at the involved BSs for the de-
sired signal which allows simple receivers at the MT with-
out multiuser detection processes/algorithms. Furthermore,
a separation between the inner part of the cells and the
broadcast area can be achieved by an overlaying scrambling
code on the signal which can be also used for synchronization
issues as in UMTS [4].
Additionally, if a single MT or more MTs are aware that
they are at the cell border, they could already ask for the C-
CDD or CAT procedure on the first hand. This would ease
the handover procedure and would guarantee a reliable soft
handover.
We should point out two main differences between C-
CDD and CAT. For C-CDD no changes at the receiver are
needed, there exists no rate loss for higher number of trans-
mit antennas, and there are no requirements regarding con-
stant channel properties over several subcarriers or sym-
bols and transmit antenna numbers. This is an advantage
over already established diversity techniques [7] and CAT.
The Alamouti scheme-based technique CAT should provide
a better performance due to the coherent combination of the
two transmitted signals [23].
4. RESULTING CHANNEL CHARACTERISTICS
FOR C-CDD
The geographical influence of the MT for CAT has a symmet-
ric behavior. In contrast, C-CDD is influenced by the posi-
tion of the served MT. Due to δcyc
0=δcyc
1and the relation in
(11), the resulting performance regarding the MT position
of C-CDD should have an asymmetric characteristic. Since
the influence of C-CDD on the system can be observed at
the receiver as a change of the channel conditions, we will
investigate in the following this modified channel in terms
of its channel transfer functions and fading correlation in
time and frequency direction. These correlation characteris-
tics also describe the corresponding single transmit antenna
channel seen at the MT for C-CDD.
The frequency domain fading processes for different
propagation paths are uncorrelated in the assumed qua-
sistatic channel. Since the number of subcarriers is larger
than the number of propagation paths, there exists correla-
tion between the subcarriers in the frequency domain. The
received signal at the receiver in C-CDD can be represented
by
Yl,i=Xl,i·
NBS−1
n=0
e−j(2π/NFFT )iδnαdnH(n)
l,i
H
l,i
+Nl,i.(17)
Since the interest is based on the fading and signal character-
istics observed at the receiver, the AWGN term Nl,iis skipped
for notational convenience. The expectation
Rl1,l2,i1,i2=EH
l1,i1·H∗
l2,i2(18)
yields the correlation properties of the frequency domain
channel fading. Due to the path propagations α(dn)and
the resulting power variations, we have to normalize the
channel transfer functions H(n)
l,iby the multiplication factor
1/NBS−1
n=0α2(dn) which is included for Rn(l,i).
The fading correlation properties can be divided in three
cases. The first represents the power, the second investigates
the correlation properties between the OFDM symbols (time
direction), and the third examines the correlation properties
between the subcarriers (frequency direction).
Case 1. Since we assume uncorrelated subcarriers the auto-
correlation of the CTF (l1=l2=l,i1=i2=i)is
R(l,i)=
NBS−1
n=0
e−j(2π/NFFT )iδn·e+j(2π/NFFT)iδn
=1
α2dn
·EH(n)
l,i·H(n)∗
l,i
E{|H(n)
l,i|2}=1
=
NBS−1
n=0
α2dn,
(19)
and the normalized power is
Rn(l,i)=
NBS−1
n=0
α2dnE⎧
⎪
⎪
⎪
⎪
⎨
⎪
⎪
⎪
⎪
⎩
H(n)
l,i
NBS−1
n=0α2dn
2⎫
⎪
⎪
⎪
⎪
⎬
⎪
⎪
⎪
⎪
⎭=1.(20)
Simon Plass et al. 7
60
40
20
0
Sub-carrier
0
200
400
600
Distance (m)
0
0.2
0.4
0.6
0.8
1
Correlation factor ρ
Figure 7: Characteristic of correlation factor ρover the subcarriers
depending on the distance d0.
Case 2. The correlation in time direction is given by
l1=l2,i1=i2=i. Since the channels from the BSs are i.i.d.
stochastic processes, E{H(n)
l1,i·H(n)∗
l2,i}=E{Hl1,i·H∗
l2,i}and
Rl1=l2,i=EHl1,iH∗
l2,iNBS−1
n=0
α2dn,
Rnl1=l2,i=E$Hl1,iH∗
l2,i
NBS−1
n=0α2dn%NBS−1
n=0
α2dn
=EHl1,iH∗
l2,i.
(21)
We see that in time direction, the correlation properties of
the resulting channel are independent of the MT position.
Case 3. In frequency direction (l1=l2=l,i1=i2) the corre-
lation properties are given by
Rl,i1=i2=EHl,i1H∗
l,i2·
NBS−1
n=0
α2dne−j(2π/NFFT )(i1−i2)δn
C-CDD component
.
(22)
For large dn(α(dn) gets small) the influence of the C-CDD
component vanishes. And there is no beneficial increase of
the frequency diversity close to a BS anymore. The normal-
ized correlation properties yield
Rnl,i1=i2=EHl,i1H∗
l,i2
·1
NBS−1
n=0α2dn·
NBS−1
n=0
α2dne−j(2π/NFFT )(i1−i2)δn
correlation factor ρ
.
(23)
The correlation factor ρis directly influenced by the C-
CDD component and determines the overall channel corre-
lation properties in frequency direction. Figure 7 shows the
characteristics of ρfor an exemplary system with NFFT =64,
γ=3.5, NBS =2, r=300 m, δcyc
0=0, and δcyc
1=7. One
0 102030405060
Sub-carrier
0.6
0.7
0.8
0.9
1
Correlation factor ρ
d0=334 m
d0=335 m
d0=336 m
Figure 8: Correlation characteristics over the subcarriers for d0=
[334 m, 335 m, 336 m].
0102030405060
Delay
1e−04
1e−03
1e−02
1e−01
BER
0
0.5
1
1.5
2
2.5
SNR gain (dB)
SNR gain at BER =1e−03
C-CDD, C/I =0dB
Figure 9: BER and SNR gains versus the cyclic delay at the cell bor-
der (C/I =0dB).
sample of the delay represents 320 microseconds or approx-
imately 10 m, respectively. In the cell border area (200 m <
d0<400 m), C-CDD increases the frequency diversity by
decorrelating the subcarriers. As mentioned before, there is
less decorrelation the closer the MT is to a BS.
A closer look on the area is given in Figure 8 where the in-
herent delay and the added cyclic delay are compensated, that
is, for d0=335 m the overall delay is δ1=δ(265 m) + δcyc
1=
−70 m + 70 m =0 (cf. (11)). The plot represents exemplar-
ily three positions of the MT (d0=[334 m, 335 m, 336 m])
and shows explicitly the degradation of the correlation prop-
erties over all subcarriers due to the nonexisting delay in the
system. These analyses verify the asymmetric and δcyc depen-
dent characteristics of C-CDD.
8 EURASIP Journal on Wireless Communications and Networking
Tab l e 1: Parameters of the cellular transmission systems.
Bandwidth B100.0MHz
No. of subcarriers Nc1664
FFT length NFFT 2048
Guard interval length NGI 128
Sample duration Tsamp 10.0ns
Frame length Nframe 16
No. of active users Nu{1, ...,8}
Spreading lengh L8
Modulation — 4-QAM, 16-QAM
Interleaving C-CDD — 2D
Interleaving CAT — 1D, 2D
Channel coding — CC (561, 753)oct
Channel coding rate R1/2
Channel model — IEEE 802.11n Model C
Velocity — 0 mph, 40 mph
−10 0 10 20 30
C/I (dB)
1e−04
1e−03
1e−02
1e−01
BER
w/o TX diversity, fully loaded
w/o TX diversity, half loaded
C-DD, halved TX power
C-CDD, halved TX power
C-DD
C-CDD
Figure 10: BER versus C/I for an SNR of 5 dB using no transmit
diversity technique, C-DD, and C-CDD for different scenarios.
5. SIMULATION RESULTS
The simulation environment is based on the parameter as-
sumptions of the IST-project WINNER for next genera-
tion mobile communications system [24]. The used chan-
nel model is the 14 taps IEEE 802.11n channel model C with
γ=3.5andτmax =200 nanoseconds. This model represents
a large open space (indoor and outdoor) with non-light-of-
sight conditions with a cell radius of r=300 m. The trans-
mission system is based on a carrier frequency of 5 GHz, a
bandwidth of 100 MHz, and an FFT length of Nc=2048.
One OFDM symbol length (excluding the GI) is 20.48 mi-
croseconds and the GI is set to 0.8 microseconds (corre-
sponding to 80 samples). The spreading length Lis set to
8. The number of active users can be up to 8 depending on
the used RL. 4-QAM is used throughout all simulations and
for throughput performances 16-QAM is additionally inves-
tigated. For the simulations, the signal-to-noise ratio (SNR)
is set to 5 dB and perfect channel knowledge at the receiver
is assumed. Furthermore, a (561, 753)8convolutional code
with rate R=1/2 was selected as channel code. Each MT
moves with an average velocity of 40 mph (only for compar-
ison to see the effect of natural time diversity) or is static.
As described in Section 3, users with similar demands at the
cell border are combined within time-frequency units. We
assume i.i.d. channels with equal stochastic properties from
each BS to the MT. If not stated otherwise, a fully loaded sys-
tem is simulated for the transmit diversity techniques, and
therefore, their performances can be seen as upper bounds.
All simulation parameters are summarized in Tabl e 1. In the
following, we separate the simulation results in three blocks.
First, we discuss the performances of CDD; then, the simula-
tion results of CAT are debated; and finally, the influence of
the MAI to both systems and the throughput of both systems
is investigated.
5.1. C-CDD performance
Figure 9 shows the influence of the cyclic delay δcyc
1to the
bit-error rate (BER) and the SNR gain at the cell border
(C/I =0 dB) for C-CDD. At the cell border there is no in-
fluence due to C-DD, that is, (δ1=0). Two characteristics
of the performance can be highlighted. First, there is no per-
formance gain for δcyc
1=0 due to the missing C-CDD. Sec-
ondly, the best performance can be achieved for an existing
higher cyclic shift which reflects the results in [25]. The SNR
gain performance for a target BER of 10−3depicts also the
influence of the increased cyclic delay. For higher delays the
performance saturates at a gain of about 2 dB.
The performances of the applied C-DD and C-CDD
methods are compared in Figure 10 with the reference sys-
tem using no transmit (TX) diversity technique. For the
reference system both BSs are transmitting independently
Simon Plass et al. 9
−10 0 10 20 30
C/I (dB)
1e−05
1e−04
1e−03
1e−02
1e−01
BER
w/o TX diversity, fully loaded
w/o TX diversity, half loaded
CAT, halved TX power, 0 mph
CAT, 0 mph, 2D interleaving
CAT, 0 mph
CAT, 40 mph
Figure 11: BER versus C/I for an SNR of 5 dB using no transmit
diversity and CAT for different scenarios.
00.25 0.50.75 1
Resource load
1e−04
1e−03
1e−02
1e−01
BER
C-CDD, C/I =10 dB
CAT, C/I =10 dB
C-CDD, C/I =0dB
CAT, C/I =0dB
Figure 12: Influence of the MAI to the BER performance for vary-
ing resource loads at the cell border and the inner part of the cell.
their separate MC-CDMA signal. From Figure 9, we choose
δcyc
1=30 samples and this cyclic delay is chosen through-
out all following simulations. The reference system is half
(RL =0.5) and fully loaded (RL =1.0). We observe a
large performance gain in the close-by area of the cell bor-
der (C/I =−10 dB, ..., 10 dB) for the new proposed diversity
techniques C-DD and C-CDD. Furthermore, C-CDD en-
ables an additional substantial performances gain at the cell
border. The C-DD performance degrades for C/I =0dBbe-
cause δ=0 and no transmit diversity is available. The same
effect can be seen for C-CDD at C/I =−4.6dB (δ1=−30,
δcyc
1=30 ⇒δ=0); see also Section 4. Since both BSs in
C-DD and C-CDD transmit the signal with the same power
−10 0 10 20 30
C/I (dB)
0
20
40
60
80
100
Max throughput per user (%)
C-CDD, 4-QAM
C-CDD, halved TX power, 4-QAM
w/o TX diversity, RL =0.5, 4-QAM
w/o TX diversity, RL =1, 4-QAM
C-CDD, 16-QAM
w/o TX diversity, RL =0.5, 16-QAM
w/o TX diversity, RL =1, 16-QAM
Figure 13: Throughput per user for 4-QAM versus C/I using no
transmit diversity or C-CDD with full and halved transmit power.
as the single BS in the reference system, the received signal
power at the MT is doubled. Therefore, the BER performance
of C-DD and C-CDD at δ=0 is still better than the refer-
ence system performance. For higher C/I ratios, that is, in the
inner cell, the C-DD and C-CDD transmit techniques lack
the diversity from the other BS and additionally degrade due
to the double load in each cell. Thus, the MT has to cope
with the double MAI. The loss due to the MAI can be di-
rectly seen by comparing the transmit diversity performance
with the half-loaded reference system. The fully loaded ref-
erence system has the same MAI as the C-CDD system, and
therefore, the performances merge for high C/I ratios. To es-
tablish a more detailed understanding we analyze the C-CDD
with halved transmit power. For this scenario, the total desig-
nated received power at the MT is equal to the conventional
MC-CDMAsystem.Thereisstillaperformancegaindueto
the exploited transmit diversity for C/I <5dB. The perfor-
mance characteristics are the same for halved and full trans-
mit power. The benefit of the halved transmit power is a re-
duction of the intercellular interference for the neighboring
cells. In the case of varying channel models in the adjacent
cells, the performance characteristics will be the same but not
symmetric anymore. This is also valid for the following CAT
performances.
5.2. CAT performance
Figure 11 shows the performances of the applied CAT in the
cellular system for different scenarios. If not stated otherwise,
the systems are using a 1D interleaving. In contrast to the
conventional system, the BER can be dramatically improved
at the cell border. By using the CAT, the MT exploits the addi-
tional transmit diversity where the maximum is given at the
10 EURASIP Journal on Wireless Communications and Networking
−10 0 10 20 30
C/I (dB)
0
20
40
60
80
100
Max throughput per user (%)
CAT, 4-QAM
CAT, halved TX power, 4-QAM
w/o TX diversity, RL =0.5, 4-QAM
w/o TX diversity, RL =1, 4-QAM
CAT, 16-QAM
w/o TX diversity, RL =0.5, 16-QAM
w/o TX diversity, RL =1, 16-QAM
Figure 14: Throughput per user for 4-QAM and 16-QAM versus
C/I using no transmit diversity or CAT with full and halved transmit
power.
cell border. If the MT moves with higher velocity (40 mph),
the correlation of the subcarrier fading coefficients in time
direction decreases. This incremental violation of the qua-
sistationarity assumption of the fading is profitable compen-
sated by the channel code. The total violation of the afore-
mentioned constraint of CAT (cf. Section 3.2)isachievedby
a fully interleaved (2D) MC-CDMA frame. There is a large
performance degradation compared to the CAT performance
with a noninterleaved frame. Nevertheless, a residual trans-
mit diversity exists, the MT benefits at the cell border, and
the performance is improved. The applied CAT is not only
robust for varying MT velocities but also for non-quasistatic
channel characteristics. Similar to C-CDD, thereis still a per-
formance gain due to the exploited transmit diversity for
C/I <5 dB in the case of halved transmit powers at both BSs.
5.3. MAI and throughput performance of
C-CDD and CAT
The influence of the MAI is shown in Figure 12. The BER
performance versus the resource load of the systems is pre-
sented. Two different positions of the MT are chosen: di-
rectly at the cell border (C/I =0 dB) and closer to one BS
(C/I =10 dB). Both transmit diversity schemes suffer from
the increased MAI for higher resource loads which is in the
nature of the used MC-CDMA system. CAT is not influenced
by the MAI as much as C-CDD for both scenarios. Both per-
formances merge for C/I =10 dB because the influence of
the transmit diversity techniques is highly reduced in the in-
ner part of the cell.
Since we assume the total number of subcarriers is
equally distributed to the maximum number of users per cell,
each user has a maximum throughput of ηmax. The through-
put ηof the system, by using the probability P(n) of the first
correct MC-CDMA frame transmission after n−1failedre-
transmissions, is given by
η=∞
n=0
ηmax
n+1P(n)≥ηmax(1 −FER).(24)
A lower bound of the system is given by the right-hand side
of (24) by only considering n=0 and the frame-error rate
(FER). Figures 13 and 14 illustrate this lower bound for dif-
ferent modulations in the case of C-CDD and CAT.
C-CDD in Figure 13 outperforms the conventional sys-
tem at the cell border for all scenarios. Due to the almost van-
ishing performance for 16-QAM with halved transmit power
for an SNR of 5 dB, we do not display this performance curve.
For 4-QAM and C-CDD, a reliable throughput along the cell
border is achieved. Since C-CDD with halved transmit power
still outperforms the conventional system, it is possible to de-
crease the intercellular interference.
The same performance characteristics as in C-CDD re-
garding the throughput can be seen in Figure 14 for applying
the transmit diversity technique CAT. Due to the combina-
tion of two signals in the Alamouti scheme, CAT can pro-
vide a higher throughput than C-CDD in the cell border area.
The CAT can almost achieve the maximum possible through-
put in the cell border area. For both transmit diversity tech-
niques, power and/or modulation adaptation from the BSs
opens the possibility for the MT to request a higher through-
put in the critical cell border area. All these characteristics
can be utilized by soft handover concepts.
6. CONCLUSIONS
This paper handles the application of transmit diversity tech-
niques to a cellular MC-CDMA-based environment. Ad-
dressing transmit diversity by using different base stations for
the desired signal to a mobile terminal enhances the macro-
diversity in a cellular system. Analyses and simulation re-
sults show that the introduced cellular cyclic delay diversity
(C-CDD) and cellular Alamouti technique (CAT) are capa-
ble of improving the performance at the severe cell borders.
Furthermore, the techniques reduce the overall intercellu-
lar interference. Therefore, it is desirable to use C-CDD and
CAT in the outer part of the cells, depending on available re-
sources in adjacent cells. The introduced transmit diversity
techniques can be utilized for more reliable soft handover
concepts.
ACKNOWLEDGMENTS
This work has been performed in the framework of the IST
Project IST-4-027756 WINNER, which is partly funded by
the European Union. The authors would like to acknowledge
the contributions of their colleagues. The material in this pa-
per was presented in part at the IEEE 64th Vehicular Technol-
ogy Conference, Montr´
eal, Canada, September 25–28, 2006.
Simon Plass et al. 11
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Photographȱ©ȱTurismeȱdeȱBarcelonaȱ/ȱJ.ȱTrullàs
Preliminaryȱcallȱforȱpapers
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nineteenth in a series of conferences promoted by the European Association for
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EUSIPCOȬ2011 will focus on key aspects of signal processing theory and
li ti
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HonoraryȱChair
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GeneralȱChair
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JinhongȱYuanȱ(UNSWȬAustralia)
TamasȱSziranyiȱ(SZTAKIȱȬHungary)
RichȱSternȱ(CMUȬUSA)
RicardoȱL.ȱdeȱQueirozȱȱ(UNBȬBrazil)
Webpage:ȱwww.eusipco2011.org
P
roposa
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f
orȱspec
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ȱsess
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15
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ecȱ
2010
Proposalsȱforȱtutorials 18ȱFeb 2011
Electronicȱsubmissionȱofȱfullȱpapers 21ȱFeb 2011
Notificationȱofȱacceptance 23ȱMay 2011
SubmissionȱofȱcameraȬreadyȱpapers 6ȱJun 2011
