Microwave and Optical Technology Letters

Published by Wiley
Online ISSN: 1098-2760
Print ISSN: 0895-2477
Discipline: Communication Technology
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Aims and scope

Microwave and Optical Technology Letters provides quick publication (3 to 6 month turnaround) of the most recent findings and achievements in high frequency technology, from RF to optical spectrum. The journal publishes original short papers and letters on theoretical, applied, and system results in the following areas: RF, Microwave, and Millimeter Waves; Antennas and Propagation; Submillimeter-Wave and Infrared Technology; Optical Engineering.

Recent publications
  • Antonio M. de Oliveira NetoAntonio M. de Oliveira Neto
  • João F. JustoJoão F. Justo
  • Wesley BeccaroWesley Beccaro
  • Alexandre M. de OliveiraAlexandre M. de Oliveira
This work presents a technique to produce radio frequency (RF) and microwave devices with relative permittivity (εr ${\varepsilon }_{{\rm{r}}}$) variations in specific locations. It is based on additive manufacturing, using two or more filaments, with different permittivities, which are introduced into the print head and mixed with designed proportions. While other techniques can vary the dielectric permittivity using air cavities along the object, our proposal allows varying permittivity without cavities, which provides devices with more stable properties. As an application of this technique in RF devices, a dielectric planar lens, with varying permittivity, was designed and built to work along with a planar Vivaldi antenna. The antenna plus lens system was experimentally measured, indicating gains of up to 1.98 dBi when compared to the system without the lens. It was also demonstrated that the lens has ultra‐wideband properties since it presented an increase in gain when compared to the reference one from 1.5 to 3.0 GHz.
Diagram of the proposed MMIC high‐power PIN limiter
(A) The principle layout of the C‐band MMIC high‐power PIN limiter. (B) Schematic diagram of the limiter. (C) Scanning electron microscopy of heterostructured MMIC limiter.
Microwave performances of the MMIC limiter. (A) Measurement setup. (B) VSWR of the input port. (C) VSWR of the input port. (D) Variation of insertion loss in frequency band.
(A) Comparisons of S‐parameters of C‐band MMIC limiter. (B) Comparisons of VSWR of C‐band MMIC limiter.
(A) The leakage power curves at 5.5 GHz under different ambient temperatures (e.g., 25°C, 75°C, and 125°C). (B) The change curve of insertion loss in small signal state after high‐power input.
  • Zhou DaiZhou Dai
  • Chunyu LiChunyu Li
  • Junda YanJunda Yan
  • [...]
  • Dazhi DingDazhi Ding
This paper designs a mesa‐structured monolithic microwave integrated circuit (MMIC) limiter working in C‐band. The quasi‐vertical Si‐based PIN diodes are heterogeneously integrated on the SiC substrate with high thermal conductivity, such that the parasitic parameters are greatly reduced and the power capacity is enhanced. According to the measured results, a maximum insertion loss of 1.1 dB and a minimum return loss of 17.69 dB for both input and output terminals are realized to cover the 5–6 GHz frequency range. The flat leakage of 23 dBm is achieved for a 150 W continuous wave and 370 W pulse input signal level. We believe the proposed MMIC limiter provides an effective tool that can be used in the transmit‐receive modules working in C‐band.
M × N planar array with element spacing dx, dy in x, y direction.
Three‐dimensional shaped beams generated from multiple beams of unequal power. (A) Peak angle and power required for a stepped pattern; (B) Typical cosine antenna element pattern; (C) Ideal scanned beam peaks at null angles; (D) Scanned peak power with the influence of antenna factor; (E) Peak power recalculated for element pattern; (F) A three‐level stepped beam generated from a 16 × 16 element array; (G) ϕ = 0° 2D plot with the beam peaks; (H) ϕ = 90° 2D plot.
Contour pattern generation using multiple beam peaks on a 16 × 16 element array. (A) Required input contour image; (B) Contour image mapped to coverage area with power levels represented by color scale on the null location p, q; (C) Selected peaks and their locations with power ratio; (D) Realized contour pattern; (E) Amplitude weights for the contour pattern on 16 × 16 array; (F) Phase weights; (G) Contour map with a null zone on a 32 × 32 element array; (H) Phi cut at 70° angles of the blocked region and phi cut 135° shows the flat region.
Pattern measurement. (A) The antenna pattern measurement setup; (B) Phased array antenna stackup using high‐frequency laminates and prepregs; (C) The 256 element antenna array mounted on a heat sink.
Shaped beam pattern measurement. (A) Desired flat beam pattern; (B) Desired step beam pattern; (C) Calculated peaks and their power level for a flat beam; (D) Measured scanned beam power levels; (E) Peak power modified for the scanned peak level; (F) Measured flat beam pattern for ϕ = 0° angle; (G) Modified peak values for a stepped beam pattern with 6 dB power difference; (H) Measured 2D plots at ϕ = 0° of the stepped beam pattern.
  • Sankaravel PrasadSankaravel Prasad
  • Murugappan MeenakshiMurugappan Meenakshi
  • Patnam Hanumantha RaoPatnam Hanumantha Rao
A multiple‐beam synthesis method is proposed to generate three‐dimensional (3D) shaped and contour patterns from a planar phased array antenna. Three‐dimensional radiation patterns are synthesized primarily with optimum beam positions and variable mainbeam power levels of multiple beams, followed by calculating the resultant element weights. Contour patterns are synthesized from multiple beam peaks by mapping the individual beam power levels to their corresponding 3 dB beamwidth coverage areas at each designated location on the contour. The far‐field contour shape affected by the influence of the element pattern, mutual coupling, and scan loss is compensated by the peak power of the scanned beam patterns. The proposed synthesis supports dynamically changeable contour shapes. Flat‐topped, stepped 3D patterns and arbitrary contour shapes with shaped null zones inside the contour are simulated and implemented on a 16 × 16 element 5G mmWave planar phased array. Measured radiation patterns are compared with simulations and presented.
The schematic and photograph of the implemented power amplifier
(A) Simulated load impedance trajectories at CG plane on Smith chart with impedance conditions and, (B) frequency responses of real and imaginary parts at fo, 2fo, and 3fo
Simulation results of the implemented power amplifier (PA): (A) drain efficiency (DE), (B) gain and phase curves with and without LC tuned circuits. Meanwhile, the black dashed and solid lines show the gain and phase properties after the application of LC‐tuned elements for two different values with a margin of ±0.2, respectively, (C) Lower/upper sideband third‐order intermodulation distortion (IMD3) response before and after the addition of LC tuned circuits.
Implemented power amplifier (PA) continuous wave (CW) measurement performances at 2.4 GHz as a function of output power with respect to, (A) amplitude‐to‐amplitude (AM‐AM) and amplitude‐to‐phase (AM‐PM). (B) Normalized gain and phase response for a 5 MHz 6.9 dB 64‐QAM peak‐to‐average power ratio (PAPR) signal.
(A) Measured efficiency with drain efficiency (DE) at −40 dBc displayed. (B) Measured IMD3 performance of the power amplifier (PA) versus output power under two‐tone test with the tone spacing of 5 MHz. (C) Measured adjacent channel leakage ratio (ACLR) under 5 MHz modulated signal with 34.5 dBm average output power.
An optimal balance between high efficiency and high linearity is one of the main performance metrics of modern base stations to handle the deep compression of the power amplifier (PA) module. In principle, the plethora of amplitude‐to‐amplitude (AM‐AM) and amplitude‐to‐phase (AM‐PM) distortions are issues worth exploring. Therefore, this paper presents a linearized harmonic‐tuned PA that operates at 2.4 GHz. The presented PA utilizes a compact input matching network (IMN) and output MN (OMN) with shunt‐connected tunable resonant circuits augmented by the stabilization network, whose presence greatly reduces the transistor parasitics and high‐order effects. Hence, a joint embodiment of each other gives optimal fundamental impedance matching alongside harmonic terminations and precise AM/PM waveform properties. The measurement results have shown that the fabricated PA exhibits peak saturated output power of 40.9 dBm and peak drain efficiency (DE) of 67%. In addition, the AM‐AM and AM‐PM curves under continuous‐wave excitation yield a gain flatness of 0.5 dB over a 37.5 dBm power range and ±3.5° phase distortion, respectively, when the input power level is swept up to the saturation level of 40 dBm. When driven with a 5 MHz 64‐QAM OFDM signal and 6.9 dB power back‐off, the manufactured PA meets the adjacent channel leakage ratio specification of −30 dBc at an average output power of 34.5 dBm.
  • Ru‐Song LiRu‐Song Li
  • Xiao‐Hai ZhengXiao‐Hai Zheng
  • Hong‐Tao DangHong‐Tao Dang
  • [...]
  • Chuan ZhangChuan Zhang
A first principle calculation is performed on Ce3Al by means of density functional theory combined with dynamical mean‐field theory. We find that j = 5/2, j = 7/2 components are in the metallic and insulating regimes, respectively. Itinerant 4f electrons result in the so‐called virtual charge fluctuations in the context of Kondo model with an average occupancy number of 4f electrons about 1.0. Finally, the so‐called quasiparticle band structure is also discussed for comparison with experimental angle‐resolved photoemission spectrum.
Schematic drawing of the plasmonic photoconductive antenna (PCA), optical image, and SEM images of the fabricated device. (A) Schematic drawing of the plasmonic PCA. (B) Optical image of the PCA device. (C) SEM image of the plasmonic grating located in region B. (D) Enlarged SEM image of the plasmonic grating.
Terahertz (THz) signal generated by the conventional photoconductive antenna (PCA) (blue), the plasmonic PCA (orange). (A) Time‐domain signal (the signal of the conventional PCA is shifted on the vertical axis). (B) The power spectral density of the THz output signal.
FEM simulation result. (A) Distribution of the norm of the time‐averaged Poynting vector near the surface of the device with plasmonic grating (|〈Sgrating〉|). (B) Distribution of the norm of the time‐averaged Poynting vector near the surface of the device without plasmonic grating (|〈SGaAs〉|). (C) Comparison of the norm of the time‐averaged Poynting vector at y = 0, −5, −10, −15, −20 nm, and x = −100 nm to 100 nm. (D) Cumulative absorbed power near the device surface and the enhancement factor (Ae) for absorbance.
  • Gyejung LeeGyejung Lee
  • Byungwoo SonByungwoo Son
  • Do‐Kyeong KoDo‐Kyeong Ko
  • [...]
  • Jae‐Hyung JangJae‐Hyung Jang
A large aperture photoconductive antenna (PCA) with a plasmonic grating structure was fabricated and characterized. The intensity of the terahertz radiation from the plasmonic PCA was enhanced two‐fold compared with that from the conventional PCA without the plasmonic grating structure. By using an electrically isolated nano‐grating structure, the performance enhancement was found to be due to the plasmonic effect other than bias field enhancement. The photo‐absorption enhancement near the GaAs surface and the plasmonic grating interface was investigated to elucidate the terahertz radiation enhancement mechanism in the plasmonic PCA. The highly localized electric field near the nanoscale grating shows optical absorption at the GaAs surface was enhanced. The enhanced optical absorption increased the photo‐generated carrier density at the device surface, leading to enhanced intensity and bandwidth of the terahertz radiation from the plasmonic PCA.
  • Xiaoqian ZhangXiaoqian Zhang
  • Hanshan LiHanshan Li
In the weapon test range, the space projectile explosion position parameter at the terminal trajectory is an important index to evaluate the damage performance of the projectile fuze. Due to the random distribution of the falling dispersion of the projectile, it is difficult to test the projectile explosion position in the long‐range terminal trajectory falling area. To solve this problem, this paper proposes a method for measuring the projectile explosion position based on ultra‐wide imaging with a multi‐lens array. A curved multi‐lens array optical imaging system is designed and the parameter calculation method of the optical system is presented. The relationship between the space projectile explosion position and the center of the sub‐eye lens and the relationship between the center of the sub‐eye lens and the imaging point of the image sensor is established according to the imaging principle of the optical system. The object image relations of any seven adjacent sub‐eye lenses of the curved multi‐lens array optical imaging system are combined and the calculation model of the space projectile explosion position is deduced. The experimental results verify that the designed curved multi‐lens array optical imaging system and the calculation model of the projectile explosion position proposed in this paper are reasonable.
  • Xiaolin ZhangXiaolin Zhang
  • Xiaoyang YuXiaoyang Yu
  • Tianqi QiTianqi Qi
  • [...]
  • Yuqiang YangYuqiang Yang
In this paper, a doubly passively Q‐switched (PQS) Tm:YAP laser was experimentally realized by using two WS2‐based saturable absorbers (SAs) for the first time. Under the doubly PQS operation, the pulse width of 0.93 µs at 1989.3 nm was obtained with an average output power (AOP) of 391 mW and peak power of 4.04 W. While for PQS Tm:YAP laser with single WS2‐based SA, a pulse width of 1.63 µs and an AOP of 676 mW was achieved with a peak power of 3.4 W. The experiment results indicate a conclusion that the doubly PQS technology is an efficient method to further compress the pulse width, enhance the stability of the pulses train, and improve the AOP for the Tm:YAP laser.
  • Pengye SongPengye Song
  • Jianing TaoJianing Tao
  • Yangyang LiYangyang Li
  • [...]
  • Hui LiuHui Liu
Polarization‐duplexing of a ytterbium‐doped fiber mode‐locked laser in all‐normal‐dispersion with an intracavity birefringent crystal is demonstrated. The laser outputs two asynchronous pulse trains with slightly different repetition rates, orthogonally crossed polarization, and colinear beams. The optical spectra are highly overlapped. A tuning of the repetition rate difference ranging from 3.7 to 8.6 kHz is realized by adjusting the birefringence in fiber. This laser offers a platform for exploring the compact fiber dual‐comb lasers with tunable repetition rate differences.
Schematic diagram of array structure and antenna element phase control. (A) 3‐D view, (B) top view of the array, (C) front view of the array, and (D) back view of the array. Top and side views of the current distribution of the antenna array at different times. (E) t = 0, top view, (F) t = 0, side view at φ = 0° (G) t = T/4, top view, and (H) t = T/4, side view at φ = 0°.
Top and side views of the antenna element. (A) Top view and (B) side view. (C) Comparison of the S‐parameter simulated and measured the results of the element.
Simulation results of radiation patterns and axial ratios corresponding to different R2. (A), (B) at R2 = 47 mm, (C), (D) at R2 = 49.5 mm, and (E), (F) at R2 = 52 mm. When R2 = 49.2 mm, the (G) radiation pattern contour plot and (H) axial ratio contour plot.
(A) Photograph of the fabricated antenna array. (B) Environment of the measurement. Fabricated (C) power divider and (D) coupler. The comparison of the S‐parameter simulated and measured results of the power divider and coupler. (E) Power dividers. (F) Through‐end. (G) Coupled‐end. (H) Through‐end and coupled‐end phases. The comparison of the simulated and measured results of the radiation pattern at (I) φ = 0° (xz‐plane), (J) φ = 90° (yz‐plane). (The solid line is the simulated results, and the dashed line is the measured results.) The comparison of the simulated and measured results of the axial ratio at (K) φ = 0° (xz‐plane), (L) φ = 90° (yz‐plane).
(A) Photograph of the fabricated antenna array. (B) Environment of the measurement. Fabricated (C) power divider and (D) coupler. The comparison of the S‐parameter simulated and measured results of the power divider and coupler. (E) Power dividers. (F) Through‐end. (G) Coupled‐end. (H) Through‐end and coupled‐end phases. The comparison of the simulated and measured results of the radiation pattern at (I) φ = 0° (xz‐plane), (J) φ = 90° (yz‐plane). (The solid line is the simulated results, and the dashed line is the measured results.) The comparison of the simulated and measured results of the axial ratio at (K) φ = 0° (xz‐plane), (L) φ = 90° (yz‐plane).
  • Yu LuoYu Luo
  • Guangying ZhaoGuangying Zhao
  • Ningning YanNingning Yan
  • [...]
  • Fanyi MengFanyi Meng
This study presents the phase setting method for a spherical conformal antenna array to achieve a wide beamwidth circularly polarized radiation pattern. A dodecahedral spherical array was fabricated and measured to validate the phase setting method. The array adopts a regular dodecahedron spatial distribution structure composed of 11 dual‐polarization elements. By exciting 11 elements simultaneously with designed phase distributions, the proposed spherical array can achieve a wide beamwidth of more than 210° with circular polarization by selecting the radius of the dodecahedron. In addition, the power division network of the array is designed and processed, and the feed network is simplified by rotating the antenna element port to control the phase. Measurements agree well with the simulations. The measured results show that the proposed spherical antenna array achieves a half‐power beamwidth of more than 210° and an axial‐ratio beamwidth of larger than 240°.
A novel resonant‐type group delay controller based on third‐order resonant topology with tunable central frequency and coupling coefficients is proposed. The coupling structure is realized by coupled line resonators loaded with two shunt capacitors for coupling coefficient tuning, and a series capacitor for resonant frequency tuning, then the group delay and working frequency can be tuned. The structure has the advantage of a large range of tunable coupling coefficients and breaks through the group delay tunable range of traditional delay circuits. The circuit is designed, simulated, and measured. The measurement shows that the group delay can be tuned from 5 to 20 ns with a tunable central frequency in the range of 0.8–1.05 GHz, and minimum ± 0.5 ns group delay bandwidth and −3 dB magnitude bandwidth of 12 MHz.
In this letter, A wideband filtering phase shifter is proposed based on vertically installed planar (VIP) circuit structures, where the VIP circuit is used to realize a tight coupling. The even‐odd mode analysis is adopted to analyze the whole structure. Then, a 45° filtering phase shifter is designed to verify the validity of the design methodology. The simulated and measured results demonstrate that the proposed phase shifter has low insertion losses (<0.5 dB) and excellent return losses (<−15 dB). The 3°‐phase shift bandwidth is more than 64.31%.
Flowchart of DE algorithm. DE, differential evolution; MPC, multipass cell.
(A) Spot pattern based on DE algorithm simulation and (B) seven‐ring spot pattern obtained by simulation. DE, differential evolution.
(A) MPC physical map based on DE algorithm and (B) spot pattern based on DE algorithm experiment. DE, differential evolution; MPC, multipass cell.
(A) Block diagram of the TDLAS system and (B) verification of effective optical path of the developed MPC. DAQ, data acquisition; MPC, multipass cell; PC, personal computer; TDLAS, tunable diode laser absorption spectroscopy.
(A) Allan deviation of 1000 ppm methane measured by MPC designed based on DE algorithm. (B) Allan deviation of 1000 ppm methane measured by seven‐ring spot pattern MPC. DE, differential evolution; MPC, multipass cell.
To solve the contradiction between high performance and portability of tunable diode laser absorption spectroscopy (TDLAS) systems, we propose a multipass cell (MPC) design method based on differential evolution (DE) algorithm, and realize intelligent optimization design of MPC with small volume and long optical path. MPC is the core component of the TDLAS system. The optical path of MPC determines the detection accuracy of the TDLAS system. The optical path is affected by the position and angle of incident laser, mirror spacing, and other parameters. The DE algorithm is used to optimize the parameters that affect the performance of MPC, and a compact MPC with a spherical mirror diameter of 25.4 mm, a laser reflection number of 183, an optical path of 5.98 m, and a mirror spacing of 31.7 mm is designed. Under the same spherical mirror condition, the detection limit of MPC designed based on the DE algorithm is increased from 0.91 to 0.76 ppm compared with the MPC commonly used seven‐ring spot pattern. This has a guiding and exemplary role in the design and development of high space utilization MPC.
Adaptive genetic algorithm (AGA) optimization process
The structure of the basic and AGA‐optimized sensors: (A) 3D structure of the basic sensor, (B) side view of the basic sensor, (C) top surface of the basic sensor, (D) bottom surface of the basic sensor. (E) Single, (F) double, and (G) the details of the copper patch.
Simulation results of the three structures. (A) S21 for three kinds of structures and electric field distribution of (B) basic structure, (C) single structure, and (D) double structure.
The devices and the measurement results of the three biosensors (A) the fabricated devices, (B) the measurement setup, (C) measurement results of basic structure, (D) measurement results of single structure, (E) measurement results of double structure, and (F) the fitting lines of the sensors.
This paper presents three devices of biosensors for glucose concentrations. A resonator based on the defected ground structure (DGS) was designed and the optimizations for the single square and both squares of the DGS via adaptive genetic algorithm were applied to enhance the performance of the sensors including the Q‐factor and the electric field distribution. The fabricated devices of the best optimized one exhibited an enhanced Q‐factor of 442 and a sensitivity of 142.2 MHz/mgml⁻¹, which is more than 2.37 times than the basic structure.
This paper presents a modified structure of the Vivaldi antenna based on an FR4 dielectric substrate of 0.8 mm thickness and 110 mm × 96 mm dimensions for ultra‐wideband (UWB) detection systems. The modified Vivaldi antenna has two major novelties. (1) Bottle‐shaped slots are etched on both sides of the classic Vivaldi antenna to extend the surface current path length and concentrate electromagnetic energy. (2) Rectangular metal strips are implemented in front of the exponential slot to guide the UWB radiation waveform in the desired direction. The modified Vivaldi antenna is evaluated using the CST 2020 software and measured using the vector network analyzer ranging from 3 to 8 GHz. The obtained results show that a maximum peak gain of nearly 9 dBi at 4 GHz can be achieved, the radiation efficiency is improved to 80%, the main lobe level is increased, and the side lobe level is decreased. To further demonstrate the performance of the proposed Vivaldi antenna, a UWB imaging experiment is designed. The imaging results show that the imaging system using the proposed antennas can obtain images with good resolution, which indicates that the antenna has good radiation characteristics and can be used for UWB detection applications.
Proposed antenna configuration. (A) Antenna geometry with a superstrate layer (side view), (B) parametric definition of the proposed antenna (T1 and ε1 ${\varepsilon }_{1}$ = substrate thickness and permittivity, T2 and ε2 ${\varepsilon }_{2}$ = superstrate thickness and permittivity, ε1 ${\varepsilon }_{1}$ = 3.55, ε2 ${\varepsilon }_{2}$ = 10.7, εVAC ${\varepsilon }_{{VAC}}$ = 1), and (C) schematic model of the proposed antenna.
S‐Parameters of the proposed antenna with varying LSUP and LGND (LSUP = LGND). (A) Reflection coefficient |S11|, (B) port‐to‐port isolation (|S21|), and (C) Phase difference between the output ports of HC.
Radiation patterns (@phi = 0⁰) of the proposed antenna with and without (patch only) superstrate integration at various frequencies with varying LSUP and LGND (LSUP = LGND). (A) 8.2 GHz, (B) 8.5 GHz, and (C) 9 GHz.
Gain and axial ratio results of the proposed antenna at NADIR. (A) Gain and (B) axial ratio.
In this article, a dual circularly polarized high gain superstrate integrated X‐band antenna is designed and analyzed for CubeSat applications. The proposed antenna consists of a metallic patch, multilayer space qualified printed circuit boards, and a dielectric superstrate with a high relative permittivity on top of the patch to establish a suitable resonance condition for high‐gain operation. The driven patch is fed by a 3‐dB hybrid coupler for the generation of dual (left hand circular polarization and right hand circular polarization) circular polarization. The antenna exhibits an impedance bandwidth of >18.5% (7.75–9.4 GHz) with a peak gain of >16 dBiC. The achieved antenna performance makes it suitable for CubeSat missions at X‐band.
The topology of the proposed tri‐band impedance transformer
The design flow chart of the proposed tri‐band impedance transformer
The simulated |S11 | of example A and example C
Photograph of the fabricated tri‐band impedance transformer
The simulated and measured |S11| of example B
A compact tri‐band impedance transformer that can match the frequency‐dependent complex loads is presented in this letter. The proposed transformer is cascade of a dual‐band conductance transforming network, a dual‐band susceptance stub network and a dual‐to‐tri‐band transformer. There are four topologies of the transformer are available according to the different loads at the first frequency and the realizability of the transformer. All the parameters are acquired by solving the closed form design equations and a detailed design process is provided. The proposed design method is demonstrated through two examples, and a prototype is fabricated. Good agreement between the simulated and measured results is obtained. Compared with the existing works, the transformer not only can match the frequency‐dependent complex loads, but also obtains wide bandwidths and has a very short electrical length. In summary, the proposed impedance transformer is very suitable for the design of tri‐band active circuits and systems.
(A) Schematic diagram of the proposed filter. (B) Even‐mode equivalent circuit. (C) Odd‐mode equivalent circuit.
(A) The imaginary part Im(S11) and the real part Re(S11) of S11 of the proposed filter. (B) Comparison of |S11| of multimode structures with and without capacitive loading stub. (C) The S‐parameter of the traditional multimode structure.
(A) Influence of capacitor C1 on harmonic suppression, (B) Influence of Ze3/Zo3 on harmonic suppression, (C) Influence of capacitor C2 on harmonic suppression, (D) Influence of Ze4/Zo4 on harmonic suppression.
Physical structure of the proposed filter. Parameters of the proposed filter are designed as: Wm = 3.05, W1 = 0.33, W2 = 2.6, W3 = 2.35, W4 = 4.43, Wc = 0.3, L11 = 8.59, L12 = 13.08, L13 = 9.29, L14 = 15.96, L21 = 14.29, L22 = 18.3, L3 = 8.35, L4 = 7.7, Lc = 2.33, Ls1 = 1.4, Ls2 = 2, S1 = 0.25, S2 = 1, S3 = 1, S4 = 0.24, Sc1 = 0.5, Sc2 = 0.25, Viad = 0.3 (all in unit: mm).
Photograph, simulated and measured S‐parameters of fabricated wideband BPF with ultra‐stopband and high selectivity. BPF, bandpass filter.
In this paper, a miniaturized ultra‐wideband filter with ultra‐wide stopband and high selectivity using a capacitive loading stub in a multimode structure is proposed. The second, third, and fourth harmonics of the proposed filter are all suppressed on account of using a capacitive loading stub, which makes it have the characteristic of an ultra‐wide stopband. In addition, there are two transmission zeros located at the upper stopband close to the passband, which improves the selectivity of the proposed filter. The proposed filter is fabricated and measured, and the result shows the filter with the center frequency f0 1.185 GHz has 151% 3‐dB fractional bandwidth. Its upper stopband is extended to 9.18 GHz (7.75 f0) and the 20‐dB roll off rate is 212.5 dB/GHz. What's more, the size of the proposed filter is 0.175 λg × 0.18 λg, where λg is the guided wavelength at the center frequency f0.
In this paper, a miniaturized multiple‐input, and multiple‐output (MIMO) antenna array is proposed based on quarter‐mode substrate‐integrated waveguide. The decoupling structure of composite neutralization lines (NLs) is used to improve the isolation of the MIMO antenna array. Each NL comprises a simple metal strip attached to each of the two antenna radiators to create an extra coupling path that opposes the primitive antenna coupling to reduce mutual coupling. Eventually, the 2 × 2 MIMO antenna array is fabricated and implemented to verify the design. The measured impedance bandwidth of S11 less than −10 dB ranges from 3.48 to 3.52 GHz, and the peak gain of the element is 4.87 dBi, the mutual coupling levels between the orthogonal and the parallel antenna elements are −19.2, −22.6, and −17.9 dB, respectively, making it a suitable candidate for MIMO systems.
Configuration of the proposed substrate integrated waveguide horn antenna (A) top view, (B) bottom view, and proposed unit cell configuration (C) perspective view, (D) top view, and (E) bottom view.
Retrieved constitutive parameters of the proposed unit cell (A) permittivity, (B) permeability, (C) dispersion diagram and the impedance and electric field distributions of the substrate integrated waveguide horn with/without loading the bilayer metamaterials (D) Re(z), (E) Im(Z), (E) without/(G) with loading the bilayer metamaterials.
The reflection coefficients of the substrate integrated waveguide horn antenna with different structural parameters (A) with h, (B) with CL4, (C) with CL1, (D) with CL2, and the fabricated model (E) top view and (F) bottom view.
Simulated and measured reflection coefficients (A) and far‐field radiation patterns (B), gains and radiation efficiencies (C) of the proposed antenna
In this paper, a wideband planar substrate integrated waveguide (SIW) horn antenna based on surface wave excitation through loading metamaterials is designed. Strong coupling effect is realized between the employed bilayer metamaterial unit with the top layer as a π‐shaped‐slotted patch while the bottom layer as the complementary cross‐shaped‐slotted patch. The equivalent permittivity of the extended substrate in front of the planar horn aperture can be effectively reduced to unity and impedance mismatching for conventional planar SIW horn antenna can be reduced in a wide bandwidth. Results indicate that the working frequency of the antenna is 12.3–15.8 GHz with the relative impedance bandwidth of 24.6%.
Schematic diagram of the proposed multiwavelength QCLs. QCL, quantum cascade laser.
Modeling results of the heat dissipation module
Picture of the integrated source consisting of four QCLs in C‐mount package. QCL, quantum cascade laser.
Spectrums of the multiwavelength QCLs. QCL, quantum cascade laser.
Wavelength as functions of temperature and current
Integration of multiwavelength distributed feedback quantum cascade lasers (QCLs) in C‐mount packages is demonstrated in this paper, which is applicable to portable gas sensors. The thermal management is performed by modeling and experimentally verifying. The QCLs with the wavelength of 4492, 5459, 6224, and 9640 nm are integrated. The wavelength tuning range and optical power of each QCL are no smaller than 6.5 nm and 65 mW, respectively. The dimensions of the integrated sources are 40 mm × 40 mm × 80 mm $40\unicode{x0200A}\text{mm}\times 40\unicode{x0200A}\text{mm}\times 80\unicode{x0200A}\text{mm}$.
In this study, a simple and efficient dual‐port single‐loop antenna unit, intended for full metal‐frame terminals, is presented as the first of its kind. The proposed multiple‐input and multiple‐output (MIMO) antenna unit is constructed within a loop resonator, inserted between the metal frame and the ground plane without any slits or cuttings. Two voltage ports are utilized to simultaneously excite the shared loop resonator as radiators; however, the strong mutual coupling is induced from one port to another because of the shared loop resonator. Herein, a simple and efficient decoupling capacitor is loaded at the center of the loop resonator, decoupling and decorrelating the two‐port loop resonator. In this way, a dual‐port single‐loop antenna unit is accomplished for 5G MIMO applications. Two units are constructed, forming 4 × 4 MIMO antennas, which are validated in both simulation and measurement.
The proposed dual‐mode Gap waveguide (GW) cavity, Hgap = 0.5 mm, Hpin = 3 mm, La = 11.2 mm, and Ll = 12.3 mm. (A) 3D view. (B) Top view.
(A) The bottom plate. (B) The Qe of the degenerate modes versus the rotation angle of the feeding slot. (C) The coupling slot length versus the electric and magnetic coupling strength (single slot).
Topology of the filters, S/L is the source and load of the filters. (A) The dual‐mode filter. (B) The dual‐band filter. The dotted lines indicate the non‐resonating paths. The middle layer. (C) The dual‐mode filter. La = 11.3 mm, Ll = 11.85 mm, dx = 0.8 mm, dy = 0.7 mm, px = 0.95 mm, py = 0.7 mm, Lc = 8 mm, Wc = 0.7 mm, d = 4.3 mm, Lf = 7.12 mm, and Wf = 1.3 mm. (d) The dual‐band filter. La = 10.45 mm, Ll = 12.59 mm, Lc1 = 7.85 mm, Wc1 = 0.7 mm, d1 = 3.82 mm, Lc2 = 10.15 mm, Wc2 = 1.2 mm, d2 = 3.15 mm, Lf = 7.12 mm, Wf = 1.82 mm, and θ = 40 °.
E‐field of the Gap waveguide (GW) cavity for the dual‐band filter. (A) TE101, 24.6 GHz. (B) TE102, 28.3 GHz. (C) TE201, 31.5 GHz. (D) TE301, 35 GHz. (E) Center plate surface current at 30.98 GHz.
(A) 3D view of the filters. (B) The fabricated filters. The simulated and measured filter responses. (C) The dual‐mode filter. (D) The dual‐band filter.
The design of compact dual‐mode/band filters based on gap waveguide and 3D printing technology is presented in this paper. The spatial power divider is designed to arrange the external coupling. And the multi‐slot middle plate is devised to manage the coupling between the dual‐mode cavities, introducing the non‐resonating modes to produce additional transmission zeros and improving the selectivity. Moreover, the dual‐mode/band filters structure are built up and optimized for 3D printing technology. Finally, the fabricated filters are measured to verify the design.
Experimental setup of the simultaneous pulse generation and microwave spectroscopy
3D model of the TEM cell with the resonator
(A) Temporal and (B) spectral bipolar pulse profiles generated by an FW generator excited with 350 fs pulses. FW, frozen wave.
(A) Experimental setup used for twofold pulses with a 2‐cm transmission line and 4 kV input voltage. Electric pulse activated by (B) F1a, (C) F1b, and (D) F2 optical beams; (E) twofold electric pulse obtained through coherent combination of F1a, F1b, and F2; and (F) spectrum of the twofold pulse.
Measured and numerically calculated │S21│values of the TEM cell accommodating the resonator compared to measurements through microwave spectroscopy using the generator connected to the TEM cell
In this paper, we present the temporal and spectral shaping of kilovolt picosecond electrical pulses obtained by using the linear switching process with femtosecond laser source. The frozen wave generator integrating up to four photoconductive semiconductor switches is activated by varying the optical pulse amplitude and time delay. Electric pulses with rise time and duration as short as 60 and 100 ps, respectively are used to implement a microwave spectroscopy experiment. This optoelectronic device is synchronous with the optical pulses which can be used for optical spectroscopy, after nonlinear frequency conversion. This single‐shot microwave spectroscopy technique was used to characterize the resonances of a passive dielectric resonator. Pulse shaping integrating coherent wave combining is also demonstrated.
Diagram of antenna element GND, ground of the antenna
Photograph of the fabricated four‐element antenna array with end launch connector for measurement: (A) top view and (B) back view
Measured and simulated S11 of the fabricated antenna array
Measured and simulated realized gain of the antenna array
Measured radiation patterns in E‐plane (xoy‐plane) and H‐plane (xoz‐plane). (A). 6.11 GHz, E‐plane (xoy‐plane). (B). 6.11 GHz, H‐plane (xoz‐plane). (C). 8.6 GHz, E‐plane (xoy‐plane). (D). 8.6 GHz, H‐plane (xoz‐plane)
A planar antenna array with wide frequency band and wide end‐fire radiation beam is proposed in this letter. The antenna element is composed of periodic structure, and its feeding position deviates from the center of symmetry, so end‐fire radiation is obtained. The measurement results show that the antenna array operates in the frequency range of 6.11−10.38 GHz with voltage standing wave ratio less than 2, and the corresponding impedance bandwidth is 51.7%. It exhibits a wide radiation beam of more than 80° at end‐fire direction and achieves realized gain of more than 8 dB in measured band.
In this paper low profile, low cost, and highly efficient antenna for ultra‐wideband (UWB) applications is presented. A novel shape of recta‐tri substrate integrated waveguide antenna is designed to achieve a wideband response and high efficiency. The proposed antenna is printed on FR4 substrate with a thickness of 1.6 mm. The overall antenna dimension is 30 × 35 mm2. The frequency of the antenna bandwidth ranges from 3.64 to 14.17 GHz with a peak gain of 7.7 dB. The field distribution of the proposed antenna is observed. The fabricated antenna results are tested and compared, which well agree with the simulated results. The proposed antenna is suitable for Wi‐Fi, ISM, and UWB applications.
In this paper, a temperature sensing probe is designed using two cascaded Fabry‐Perot interferometers (FPIs). FPI1 is made by splicing single‐mode optical fiber (SMF)‐no‐core optical fiber‐capillary‐SMF in turn, and then drilling a micro‐hole on the capillary wall of FPI1 with femtosecond laser, and injecting dimethyl silicone oil (DSO) into the capillary through the micro‐hole with immersion method. FPI2 is made by splicing SMF‐capillary‐SMF sequentially. Due to the thermal sensitivity of DSO, the temperature sensitivity of a single FPI1 reaches −0.3674 nm/°C. When the free spectral ranges of FPI1 and FPI2 are close, their cascaded sensor will produce a Vernier effect for improving the sensitivity. The experimental results show that the sensitivity of the sensing probe is as high as 1.8456 nm/°C. In addition, the sensor has a compact structure and can be used flexibly in confined space temperature measurement.
In this paper, a novel dual‐band multilayer frequency selective surface (FSS) with high selectivity and miniaturization characteristics is proposed. The FSS consists of a top and bottom resonant layer and an intermediate coupling layer. The top and bottom resonant layers provide dual‐band passbands, and the cross‐coupling between the three layers makes the structure produce four transmission zeros (TZs) distributed two‐by‐two on both sides of the passband to improve the sharp roll‐off characteristics of the structure. Use bent structures in the top and bottom layers of the cell design to achieve miniaturization of the elements. FSS provides stable performance in incident angles from 0° to 45° with different polarizations of incident waves (TE and TM wave). The lower band of FSS works at 9.1 GHz and the higher band works at 10.26 GHz, the cell size of FSS is 0.157 λ 0 × 0.157 λ 0 × 0.015 λ 0 $0.157{\lambda }_{0}\times 0.157{\lambda }_{0}\times 0.015{\lambda }_{0}$, FSS samples were made and tested, and the test results fit well with the simulation results.
Equal 18‐way power divider, (A) global view and layer structure and (B) partial view
Equivalent model. (A) Power division network equivalent model and (B) amplitude equalization network
Comparison of the (A) loss and (B) characteristic impedance of microstrip line, stripline, and half‐air filled stripline
Amplitude imbalance with varied Φ0
Simulated and measured results of the proposed power divider. (A) Input return loss and insertion loss. (B) Phase imbalance and magnitude imbalance.
An equal 18‐way power divider, using a half‐air‐filled stripline technique, is presented in this letter. An amplitude equalization network is designed to mitigate the amplitude imbalance of the multiplexed planar/quasi‐planar power‐division network. Optimized transmission paths are obtained based on a combination of 1‐6‐3 power division and a multilayer structure, which effectively reducing transmission losses. To validate the design, this power divider, which is used to feed a sparse antenna array, is designed, optimized, fabricated, and measured. The measured return loss was better than 18 dB, and the measured insertion loss was better than 1.55 dB over the range 7–10 GHz. The amplitude imbalance is better than 0.4 dB, and the phase imbalance is better than 7.3°. The measured results are in good agreement with the simulation results.
Response diagram of the proposed cascaded filter
The proposed bandpass filter (BPF). (A) 3‐D view of the proposed BPF based on substrate integrated suspended line (SISL). (B) Schematic of the proposed BPF.
Diagram of the proposed bandpass filter (BPF)'s core circuit. (A) Top view of Substrate 3. (B) Bottom view of Substrate 3.
Photographs of the fabricated bandpass filter (BPF). (A) Manufactured details of the proposed BPF's core circuit. The SMD components are listed on the right side of the figure. (B) Photograph of the fabricated BPF. (C) Self‐packaged BPF.
Measured results for the fabricated bandpass filter (BPF). (A) S‐parameters. (B) Group delay.
This letter proposes an active‐passive cascaded bandpass filter based on the substrate‐integrated suspended line platform for the first time. By cascading an active‐RC highpass circuit with a passive‐LC lowpass filter circuit, the proposed filter achieves a GHz bandwidth, an extremely low cut‐off frequency, and a controllable in‐band gain. It has the advantages of compact size, DC‐offset cancellation, and self‐packaging. As proof of concept, a sixth‐order bandpass filter was designed and fabricated. The measured 3‐dB bandwidth is from 88 kHz to 1.1 GHz with a 5.8 dB in‐band gain, and the circuit area is only 0.021 × 0.022 λg, where λg is the guided wavelength at 0.55 GHz.
Changing situation of drain efficiency varies with φ
Output matching circuit (A) showing the circuit topology and (B) the results of the output matching circuit.
Input matching circuit (A) showing the circuit topology and (B) the results of input matching circuit.
Microwave power amplifier with the lumped element (A) showing the circuit topology and (B) photograph of the amplifier.
Measured and simulated results of microwave power amplifier (A) in the range of 2–3 GHz band (B) at 2.3 GHz.
In this paper, a hybrid continuous inverse class‐F high‐efficiency power amplifier based on phase shift analysis is proposed. A broadband and high‐efficiency microwave power amplifier are designed by adopting load pull, source pull technique, and low‐pass filter matching structure. The measured results show that the drain efficiency ranges from 67.5% to 77.8%, the saturated output power ranges from 40.2 to 42.4 dBm, and the saturated power gain ranges from 10.2 to 12.4 dB in the working frequency band from 2 to 3 GHz. The power amplifier proposed in this paper fits the current broadband microwave communication technology and plays an important role in the transmission of high data flow signals by wireless transmitters.
Hybrid frequency selective rasorber (FSRs) composed of hybrid lossy layers and bandpass frequency selective surfaces (FSSs) are proposed in this study, which works as an absorber with absorption bands of 3.34–11.95 GHz and 17.44–24.29 GHz providing the fractional bandwidths of 112% and 33% while reserving a transparent window from 13.75 to 15.32 GHz. The proposed FSR is designed by stacking a hybrid lossy layer and band‐pass FSS. One of the lossy layers is composed of a λ/4 transmission line resonator in the middle of the lumped resistors' loaded bent dipole, which realizes an absorption‐transmission (A–T) frequency response, and the other is constructed by four stripe‐type parallel inductor and capacitor resonators connected to a lumped resistors' loaded circular ring, which realizes T–A frequency response. The bandpass FSS is a three‐tier structure in which the top and the bottom layers are composed of periodical circle patches and the middle layer is composed of circle slots. To explicate the working mechanism of the hybrid FSR, an equivalent circuit model is established, meanwhile, the surface current loss is analyzed. A fabricated prototype of the proposed FSR is assembled and measured to test the simulated frequency response.
The schematic diagram of the two oxygen detection system. (A) Faraday modulation spectroscopy‐based system, (B) wavelength modulation spectroscopy‐based system. I, current driver signal; T, temperature control signal
The HITRAN simulated absorption spectrum of oxygen from 760.7 to761.1 nm (red curve) and the wavelength tuning curve of the distributed feedback laser (blue curve).
(A) The 1f signal amplitudes at different O2 concentration levels in Faraday modulation spectroscopy. (B) The comparison of the measured O2 concentration and the normal concentration.
(A) Time series measurements of 20% O2. (B) Allan deviation plots for the O2 measurements.
Allan deviation plots for the O2 measurements based on three different techniques: direct absorption spectroscopy (DAS), wavelength modulation spectroscopy (WMS) and Faraday modulation spectroscopy (FAMOS).
An oxygen sensor system was developed based on Faraday modulation spectroscopy (FAMOS). In the system, a distributed feedback laser near 760.9 nm was used as the light source and a 30‐cm long gas cell was employed for gas absorption. The first harmonic was extracted by a commercial lock in amplifier for the derivation of concentration. Oxygen measurement experiments were conducted the performance of the sensor system was obtained. In addition, oxygen sensor system with wavelength modulation spectroscopy (WMS) scheme was developed with the same light source and gas cell for a direct comparison. Based on Allan deviation analysis, a detection limit of 0.186% for the WMS scheme and a detection limit of 0.108% for the FAMOS scheme were acquired at an averaging time of 1 s. The precisions can be further improved to 0.032% and 0.03% at an averaging time of 15 s, respectively.
Configuration of the second‐order filter consisting of two traditional quarter‐mode substrate integrated waveguide cavities (A) without coupling coplanar waveguide (CPW) and (B) with coupling CPW. (C) The simulated S‐parameters of these two filters; the simulated electric field distribution at the center frequency of these two filters (D) without coupling CPW and (E) with coupling CPW; the simulated magnetic field distribution at the center frequency of these two filters (F) without coupling CPW and (G) with coupling CPW.
(A) Configuration of the proposed filter. (B) Coupling schematic topology. (C) The simulated electric field distribution at the center frequency of Passband I and Passband II. (D) Frequency of two resonant frequencies in a single resonator versus l2. (E) Analysis of achievable frequency ratio of the proposed dual‐band filter.
(A) Extracted external Q factor of Passband I vs. ls with wn as a variable. (B) Extracted external Q factor of Passband II vs. ls with wn as a variable. (C) Extracted coupling coefficients K12I vs. d1 and l3. (D) Extracted coupling coefficients K12II vs. d1 and l3. (E) Extracted coupling coefficients K12I and K12II vs. w3.
(A) The simulated S‐parameters of a single resonator. (B) The simulated S‐parameters with the different length of l3. (C) The simulated S‐parameters with the different length of w3. (D) The simulated, and measured S‐parameters of the fabricated filter.
A novel miniaturized dual‐band quarter‐mode substrate integrated waveguide (QMSIW) bandpass filter with high selectivity is proposed in this letter. Two passbands are generated by the fundamental mode and the first higher order mode of the spiral‐stub‐loaded QMSIW resonator, respectively. Miniaturization is achieved by loading stubs and applying quarter‐mode cavities. High selectivity can be realized by introducing two transmission zeros (TZs) between the two passbands. One TZ is generated by adding coplanar waveguide coupling between the two resonators, the other TZ is synthesized by coupling between the modes of the resonator. The proposed dual‐band filter has the characteristics of miniaturization, high selectivity, and low loss, which is suitable for satellite communication related applications.
Laser‐induced plasma spectroscopy (LIPS, also LIBS) is a promising technique for the challenging issues associated with the real time and in‐situ monitoring of the elements in aerosol particulate matter. A prototype of Aero‐LIPS had been set up with the techniques of aerosol beam focusing, enhanced plasma emission collector, and conditional data filter to demonstrate the potential application of air pollution composition monitoring. It can identify more than 40 elements from aerosols and continuously monitor 20 elements with a time interval of 10 min. In the field test of an Asian dust event, the major elements, such as Ca, Mg, Al, Si, Cl, P, and S, were real‐time monitored, which weighted 77.9% part of the particulate matter. The evolutions of temporal elemental concentrations went well along with the particle matter concentration. In the test, several persistent lines of U and Th have been identified from dusted air, although their concentrations range in the level of nano‐grams per cubic meter. Further work is ongoing to directly determine the detection limit of U and Th to confirm the potential application for nuclear emission safeguard.
FPGA application logic. ADC, analog‐to‐digital converter; FPGA, Field Programmable Gate Array.
(A) Programmable delay unit block diagram and (B) data conversion system of target simulator. ADC, analog‐to‐digital converter; DAC, digital‐to‐analog converter; FPGA, Field Programmable Gate Array; PLL, phase locked loop.
Architecture of variable fractional delay
(A) Range versus delay produced by simulator, (B) magnitude response of fractional filters, (C) delay response of fractional delay filter, and (D) filter simulation result
Variable fractional delay filter FPGA simulation. FPGA, Field Programmable Gate Array.
In this paper, digital parallelization and fractional delay‐based novel methods are proposed for the realization of high bandwidth, high‐resolution Ku‐band radar target simulator with 2.5 GHz intermediate frequency. High bandwidth waveform from radar is sampled by high‐speed Analog to Digital Converter, and samples are parallelized in Field Programmable Gate Array (FPGA) to work at the nominal clock frequency. In Digital RF Memory‐based target simulator, for finer range resolution, the FPGA clock frequency needs to be increased, which leads to increased system design complexity. The finer range resolution is accomplished without altering the system clock frequency using variable fractional delay filters and the digital parallelization methodology is proposed in this paper. The maximum target range that can be simulated is 20 km. As a result, memory requirements, computational complexity, and power dissipation are reduced. Finally, simulation and implementation results are presented.
Fiber‐coupled atomic magnetometer (AM) based on a spin‐exchange relaxation‐free (SERF) regime has been a promising candidate for cryogenic devices for traditional magnetoencephalography (MEG) systems, as it has a flexible structure and a small lateral section for biomedical applications. The principles, key technologies, and applications of fiber‐coupled AM based on the atomic spin effect are reviewed in this paper. The principle of fiber‐coupled AM is introduced, the characteristics of conventional and fiber‐coupled AMs are compared, and the development history and current situation are reviewed. In addition, according to the demands for MEG applications, the key technologies of fiber‐coupled AM are summarized, and the bottleneck problems and future development directions for weak magnetic measurement are analyzed.
(A) Front and side views of the proposed radiation element (dimensions in mm), (B) simulated reflection coefficient changing wd ${w}_{d}$ when s $s$ = 0 mm, (C) simulated reflection coefficient changing s $s$ when wd ${w}_{d}$ = 0.5 mm, (D) simulated antenna gain in the azimuth and elevation planes.
(A) Simplified schematic of a parallel feed network for dual‐band operation, (B) BCML‐based dual‐band parallel feed network (wS1=0.9 ${w}_{S1}=0.9$, lS1=8.8 ${l}_{S1}=8.8$, wS2=1.15 ${w}_{S2}=1.15$, lS2=11 ${l}_{S2}=11$, wo=1.55 ${w}_{o}=1.55$, lS3=25 ${l}_{S3}=25$, all in mm), and (C) simulated S‐parameters.
(A) geometry of the proposed printed collinear dipole array antenna, (B) simulated reflection coefficients and boresight gain of the proposed antenna, and simulated radiation patterns at (C) 2.45 GHz, (D) 5.2 GHz, (E) 5.5 GHz, and (F) 5.8 GHz.
Simulated reflection coefficients of the proposed antenna for spacing p $p$
(A) photograph of the fabricated collinear dipole array antenna without housing, (B) one with housing, (C) measured reflection coefficients and gains of the proposed antenna, and the measured radiation pattern at (D) 2.4 GHz, (E) 5.2 GHz, (F) 5.5 GHz and (G) 5.8 GHz.
A dual‐band printed dipole array antenna for a wireless local area network (WLAN) application is presented. A radiation element consists of four pairs of printed dipoles, and two radiation elements are collinearly arrayed. A broadside coupling microstrip line‐based two‐sectional impedance transformer is used to feed the radiation elements in parallel. The proposed dual‐band antenna is designed to cover the dual‐frequency bands (2.4–2.48 and 5.15–5.825 GHz) which are the lower‐ and upper‐frequency bands for WLAN operation. The experimental results support the proposed design theory and also exhibit the gain values of 3.89–4.88 and 5.79–7.18 dBi at the lower‐ and upper‐bands, respectively, maintaining the omnidirectional radiation pattern. The measured radiation efficiencies at each frequency points, 2.4, 5.2, and 5.5 GHz are 87.39%, 75.43%, 69.96%, and 73.07% $ \% $, respectively. In addition, for mass production, the effect from a low‐cost flexible coaxial cable, and polyethylene housing on the antenna performance is also investigated in this letter.
This paper observed the performance of the coated optical microbottle resonator (MBR) for the liquid ethanol concentration sensor. The MBR is prepared in three sizes by a technique known as “soften‐and‐compress” from silica fiber SMF‐28 with parameters named stem diameter, bottle length, and bottle diameter. The MBR is then coated with PMMA by the “drop‐casting” technique used as an ethanol sensor. The MBR‐PMMAs were characterized by microfiber diameter and managed to have a Q‐factor. The ethanol liquid concentration used is from 10% to 100% ppm. The sensitivity, linearity, repeatability, stability, and wavelength shift results were then used to determine the performance of MBR‐PMMAs as an ethanol sensor. The most significant size of MBR‐PMMA, known as “C,” is defined as an excellent ethanol liquid concentration sensor. Additionally, the resonator's size is determined and may influence its ability to perform well as a sensor.
Novel broadband circularly polarized (CP) electromagnetic band‐gap (EBG) antenna array was proposed in this paper. The antenna's radiating element is utilized by a microstrip‐fed circular patch, radial slot, and three rectangular notches. The proposed antenna elements are fed by feeding network comprised form four hybrid couplers and microstrip lines with appropriate lengths to support desired phase difference at output ports. Radiating patches are sequentially rotated to enhance CP property and form a 4 × 4 array. Finally, the novel scheme of electromagnetic structure (EBG) was inserted among radiating elements to decrease the mutual coupling. Using these structures results in the improvement of antenna bandwidth to cover the entire C‐band. In addition, CP purity is increased at the implicational band. Measured results indicate that the array has an impedance bandwidth of 4 GHz covering the C‐band totally and 3 dB axial‐ratio bandwidth of 2.6 GHz that is between 4.9 and 7.5 GHz (~42%). The antenna array has a peak gain of 8dBi at 6 GHz.
Two light source layout modes. The layout of the (A) circular and (B) polygonal with 17 LEDs.
(A–D) Illuminance and SNR of circular layout and polygonal layout.
PSO algorithm optimizes LED position results. (A) Polygonal layout after PSO, (B) convergence of illuminance uniformity, and (C) convergence of Q.
Experimental test and data comparison. (A) Experimental test of LED light source layout, (B) illuminance distribution during simulation, and (C) illuminance distribution during the experiment.
This paper proposes a novel polygonal layout based on rectangular plane stereo space. The illumination and signal‐to‐noise ratio of polygonal layout and circular layout is simulated in a 5 m × 4 m × 3 m room; The simulation results show that: compared with the illuminance uniformity of the circular layout of 0.67, the illuminance uniformity of the polygonal layout reaches 0.81, which is improved by 20.90%. The position and power of the polygonal layout are further optimized respectively by particle swarm optimization (PSO), and the illuminance uniformity and signal‐to‐noise ratio quality factors (Q) of the polygonal layout can be increased from 0.81 and 19.31 to 0.88 and 37.38 after optimization, respectively. The illuminance uniformity increased by 8.64%, and the Q increased by 93.58%. The experimental system of polygonal layout optimized by PSO is built, by comparing the simulation and experimental results, the average illuminance error of the simulation and experiment is 10.58%, the illuminance uniformity error is 5.68%, and the illuminance distribution trend is consistent.
Configuration of the proposed multiband antenna and photo of the fabricated samples. (A) The whole structure of the proposed antennas, designed geometry and fabricated prototype. The flat structure of the antenna before the folded (B) top view and (C) bottom view.
Simulated and measured performance of the antenna. (A) The reflection coefficient of the multiband antenna, (B) current distribution at different frequencies (the current distribution at 3.1 GHz is plotted when the multisection tapered slot antenna (MS‐TSA) is excited, (C) reflection and isolation coefficients from Port 1 and Port 2 at the lower frequency band, and (D) reflection coefficients from Port 2 at the mm‐wave band.
The radiation patterns of the antenna (A) at lower frequency bands 0.9, 2, and 3.1 GHz, (B) at upper‐frequency bands at 27, 28, and 29 GHz.
The simulated and measured (A) realized gain and (B) radiation efficiency at the lower band for monopole antenna and multisection tapered slot antenna (MS‐TSA) and for upper band (MS‐TSA).
With the advanced communications systems, the integration of 5G bands, including sub‐6‐GHz and millimeter‐wave (mm‐wave) bands with the existing lower bands of 3G/4G has become an essential need for the future mobile handsets. This article proposes an integration between a large‐frequency‐ratio dual‐band tapered slot antenna with a multisection meander monopole antenna on a handset board. The tapered slot Vivaldi antenna (TSVA) functions as a resonant open‐ended slot at low frequencies (sub‐6 GHz) while also acting as a high‐gain Vivaldi antenna at higher frequencies (mm‐wave 28 GHz). The modification of the slot geometry, that is, generating multisections within the slot, is shown to improve the impedance bandwidth. Furthermore, the meander‐line monopole antenna covers the bands of the existing mobile generation (3G/4G) at 0.8, 1.8, 2.1, and 2.3 GHz. Most importantly, the compact size of the proposed antenna is realized by utilizing the side wall of a mobile board.
The typical switched phase shifter topology. (A) Switched L/C type. (B) T‐type. (C) π‐type. (D) High‐pass/low‐pass networks. (E) The general applicability of these four total structures.
(A) The conventional switched L/C phase‐shifting cell. (B) The equivalent circuit at reference state. (C) The equivalent circuit at phase‐shifting state.
(A) The improved switched L/C phase‐shifting cell. (B) Small inductors replaced by microstrip lines in the phase‐shifting cell. (C) The equivalent circuit at reference state. (D) The equivalent circuit at phase‐shifting state.
Simulation results of the switched L/C phase‐shifting cells before and after improved
(A) The simplified schematic of the proposed 6‐bit phase shifter. (B) The micrograph of the proposed 6‐bit phase shifter. (C) The measured input reflection coefficient. (D) The measured output reflection coefficient. (E) Simulated and measured IL and the rms phase error. (F) Measured the relative phase shift of each state. IL, insertion loss; rms, root‐mean‐square.
A Ku wideband accurate compact 6‐bit digital phase shifter is designed in this letter using three different switched phase shift structures: the switched L/C structure, T‐type structure, and high‐pass/low‐pass network. To improve the bandwidth and the gain flatness, an improved switched L/C structure is used and the systematic cascade sequence of the phase shifter is optimized. The improved switched L/C structure solves the problem that conventional L/C networks are hypersensitive to the coupling capacitance at high‐frequency input. As a result, the improved L/C structure can achieve wideband and better s‐parameters. The proposed 6‐bit switched phase shifter is implemented in 0.15‐μm GaAs pHEMT process. The measured root‐mean‐square (rms) phase error is less than 2.8° at 12–18 GHz, and the average insertion loss (IL) is 6.5–6.9 dB. The chip size of the proposed 6‐bit digital phase shifter is 3.1 × 1 mm², including all RF and dc pads, which is very suitable for radar systems.
This research proposed a deep neural network (DNN) model employing a multilayer feedforward artificial neural network to characterize the relative permittivity and loss tangent of a solid sample in a broad frequency range from 1 to 10 GHz. The method exploited a grounded coplanar waveguide as a measurement fixture, and a vast amount of data was obtained from full‐wave simulations. The latter was used to train the proposed DNN model. We performed parametric studies to examine optimal DNN hyperparameters and improve the efficiency of the material property retrieval process. The proposed model was validated by retrieving the relative permittivity and loss tangent of a known substrate. The results show good agreement with the known reference values with a slight error of 1.2%.
Band pass filters are realized by coupled microelectromechanical resonators and their bandwidth is determined by the ratio of coupling beam stiffness to resonator beam stiffness at the coupling location. The coupling beam is connected to the resonator beam in such a way that the resonator beam length and coupling beam width are in the same axis. Practically, it is not possible for the coupling beam to make contact with the resonator beam at a single point. But presently, coupled resonators are designed for specific bandwidth by considering resonator beam stiffness at any point within the contact area between the coupling beam and resonator beam. Due to the finite width of the coupling beam, resonator stiffness at the coupling location is multivalued. It causes additional velocity coupling. So, the bandwidth calculated using single‐point stiffness on the resonator beam will be far from the bandwidth of the actually coupled resonator. For the first time, this finite‐width effect is taken into account. The novelty of this work is the consideration of this finite width effect of coupling beam and cancellation of additional velocity coupling to minimize error in bandwidth estimation. This is achieved by a combination of λ/4 and 3λ/4 coupling beams. In this work, the coupled resonator is modeled using an electrical equivalent circuit.
The configuration and photograph of the proposed antenna: (A) top view; (B) bottom view; (C) fabrication antenna top view; (D) fabrication antenna bottom view
Design evolution and optimization of the designed antenna: (A) the structure of Ant. 1, Ant. 2, Ant. 3, Ant. 4, Ant. 5, and the proposed antenna; (B) simulated S11; (C) simulated S21; (D), simulated current distribution at 3.5 GHz; (E) simulated current distribution at 5 GHz; (F) simulated reflection coefficients of L6; (G) simulated reflection coefficients of L9
The simulated and measured S‐parameters results of the designed antenna: (A) S11; (B) S21
Radiation patterns of the proposed antenna: (A) 3.5 GHz E‐plane; (B) 3.5 GHz H‐plane; (C) 5 GHz E‐plane; (D) 5 GHz H‐plane
Calculated parameters of the proposed antenna: (A) ECC; (B) DG; (C) ME; (D) TARC. DG, diversity gain; ECC, envelope correlation coefficient; ME, multiplexing efficiency; TARC, total active reflection coefficient.
A minimized planar dual‐band multiple‐input multiple‐output antenna with high efficiency for the fifth‐generation applications is proposed in this paper. The designed antenna is composed of two identical radiating elements and a defected ground structure (DGS). Each radiating element is made up of a circular ring, a U‐shaped microstrip, a T‐shaped microstrip, and an L‐shaped microstrip. A rectangular microstrip with a small square stub is connected to the DGS to improve the isolation. The size of the designed antenna element is 24 mm × 15 mm. The measured reflection coefficients are less than ∗10 dB in the 5G New Radio n77/n78/n79 (3.3–4.2 and 4.8–4.96 GHz). The total peak gain and efficiency are greater than 3.5 dBi and 82%, respectively. The values of calculated envelope correlation coefficient are lower than 0.015. The proposed dual‐band MIMO antenna is suitable for 5 G applications.
(A) Top image of the proposed antenna. The total size is 36 × 0 mm². (B) Bottom view of the proposed structure with double T‐shaped strips connected with the ground region. (C) The E‐field of PW:1 is 8.04 × 10² V/m. (D) The E‐Field of PW:2 is 2.18 × 10³ V/m. (E) The E‐field of PW:3 is 2.44 × 10³ V/m. (F) The E‐field of PW:4 is 1.67 × 10³ V/m. (G) The E‐field of PW:5 is 1.05 × 10³ V/m.
(A–E) Top and bottom views of PW:1 to PW:5. All designs are energized by the side ports.
Analysis of simulated and observed reflectance coefficients for the five presented designs. (A) PW:1 denotes the lowest S11 of −11.51 dB and two frequency bands. (B) PW:2 has a minimum S11 of −19.85 dB and four frequency bands. (C) PW:3 has a minimum S11 of −19.06 dB and one frequency band. (D) PW:4 has a minimum S11 of −34.54 dB and three frequency bands. (E) PW:5 has a minimum S11 of −19.65 dB and three frequency bands.
Transmittance and reflectance responses for various designs. (A–E) Peak isolation values for PW:1 to PW:5 are 28, 62, 25, 28, and 26 dB, respectively.
The three‐dimensional, two‐dimensional, and normalized directivity plot. (A) 3D view of PW:1. (B) Simulated and measured directivity of PW:1. (C) Normalized directivity of PW:1. (D) 3D view of PW:2. (E) Simulated and measured directivity of PW:2. (F) Normalized directivity of PW:2. (G) 3D view of PW:3. (H) Simulated and measured the directivity of PW:3. (I) Normalized directivity of PW:3. (J) 3D view of PW:4. (K) Simulated and measured directivity of PW:4. (L) Normalized directivity of PW:4. (M) 3D view of PW:5. (N) Simulated and measured directivity of PW:5. (O) Normalized directivity of PW:5.
The presented article represents the rudimentary‐shaped effective 1 × 2 MIMO antenna structure. The five variation types in the ground region shape are analyzed. The structure's total dimensions are 40 × 36 × 1.6 mm3. For efficiently manufacturing, the low‐profile substrate FR4 is employed, and the reflectance response of a simulated and modelled design is compared. The placement of two T‐shaped slits in the ground area aids in achieving a wider bandwidth, multiband response, excellent Gain, and directivity. Overall, the PW:4 has a minimum reflectance coefficient of −34.54 dB, a broad bandwidth of 3.38 GHz, a peak gain of 13.02 dB, maximum narrow band isolation of 62 dB, and a maximum normalized directivity of 156°. The Envelop Correlation Coefficient, Channel Capacity Loss, Total Active Reflection Coefficient, and Diversity Gain for MIMO design are also examined. These findings indicate the provided design's behavior as per the standard requirements.
(A) Top and (B) bottom configurations of the proposed split‐ring resonator‐based sensor
Equivalent circuit of a circular split‐ring resonator
Resonance shift measurement when MUT is applied
Resonance shifts for different dielectric constants
Resonance shift versus dielectric constant
This paper presents the development of a metamaterial‐based microwave sensor for dielectric constant measurements. The sensor is constructed with a circular split‐ring resonator on one side and a coplanar waveguide transmission line on the other side. The proposed sensor has a size of 35 mm × 35 mm × 1.27 mm and resonates at 3.31 GHz. The sensor is fabricated on a RO3210 substrate with a dielectric constant of 10.2 and a loss tangent of 0.003. Both the simulation and experimental measurements show a linear relationship between the shift of the resonance frequency and the dielectric constants of the test samples, with a coefficient of determination R2 above 0.99. The dielectric constant of an unknown material can therefore be easily estimated using the inverse linear regression model. To validate the performance of the sensor, three known substrate materials, namely, Rogers RT6002, FR4, and Rogers TMM10, were used as test samples. The measured dielectric constants for these three materials were 2.8045, 4.3775, and 8.8719, respectively. The results agree closely with those indicated in the data sheets, with discrepancies of less than 5%.
(A) Layer configuration of the proposed double spiral antenna (DSA) (all dimensions in mm), (B) Unit cell of artificial magnetic conductor (AMC) structure (top layer), (C) frequency selective surface (FSS) structure (bottom layer), and (D) simulation setup of a unit cell
(A) Fabricated prototypes, (B) reflection phase and magnitude curve of proposed artificial magnetic conductor (AMC), (C) reflection phase of frequency selective surface (FSS) structure, (D) fabricated applicator and applicator with phantom, and (E) proposed applicator with tumor embedded in phantom (all dimensions in mm), measurement setup of the proposed applicator with the phantom.
Reflection coefficient curve for different parameters optimization, (A) for spiral arm width, (B) for aperture slot length, (C) for aperture slot width, (D) for the length of feed line, (E) for artificial magnetic conductor (AMC) slot length, and (F) for AMC slot width.
(A) Measurement setup for S‐parmeter, (B) variation of S‐parameter for different configurations, and (C) field pattern and electric field distribution within phantom.
Normalized specific absorption rate (SAR) for different configurations along (A) x‐ and y‐axis, (B) along z‐axis, (C) temperature distribution, and (D) temperature profile for different applicators.
A compact focused metamaterial‐based applicator is investigated for hyperthermia cancer treatment in this letter. The metamaterial‐based applicator consists of a double spiral antenna (DSA), an artificial magnetic conductor (AMC) as a reflector, and a frequency selective surface (FSS) as a lens. An AMC structure (64 × 64 mm2) is placed in the back of the antenna at a 5 mm distance that converges the bidirectional field pattern of the antenna into the directional field pattern. Further, to increase penetration inside the tissue, an FSS structure (64 × 64 mm2) is placed at the top of DSA at a distance of 4 mm. The FSS directs the energy toward the tumor location and provides a uniform heating pattern. The field and temperature distribution are numerically obtained for the proposed applicator in the heterogeneous phantom (layers of skin, fat, and muscles), and the performance of the applicator is evaluated in terms of the specific absorption rate, penetration depth, and effective field size. The performance of the applicator is also analyzed for a deeply placed (at 16 mm from the skin surface) irregular‐shaped tumor. The temperature profile calculates 44°C at an input power of 2.5 W. The results show that the applicator is capable of heating sufficiently deep‐seated tumors.
This study proposed a method for distinguishing the baseband signal for each frequency in two‐tone radar systems using the difference in the number of peaks of the envelope detected signal. The proposed method is based on the characteristic that the baseband signal for each frequency exhibits a different number of peaks, depending on the operating frequency, at a fixed sampling frequency in data acquisition. The phase difference between the transmitting and receiving signals for each frequency can be obtained using envelope detection of the baseband signals in two‐tone radar systems, and the proposed method can effectively match the phase difference and operating frequency without using additional controls and synchronization. This simplifies the procedure to estimate the absolute range up to the maximum unambiguous range, corresponding to a phase difference of 2π. Measurement results demonstrate that the absolute range, using a two‐tone radar with a frequency spacing of 150 MHz, can detect the distance with an accuracy of 97.8% up to the maximum unambiguous range using the proposed method.
Journal metrics
$2,950 / £2,000 / €2,450
Article Processing Charges (APC)
Acceptance rate
42 days
Submission to first decision
1.311 (2021)
Journal Impact Factor™
3 (2021)
Top-cited authors
W.C. Chew
  • University of Illinois, Urbana-Champaign
Stephen Gedney
  • University of Colorado
Raj Mittra
  • Pennsylvania State University
Saou-Wen Su
  • Asus Tek Computer Inc. (Asus)
Akhlesh Lakhtakia
  • Pennsylvania State University