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Eye diagram after 2 km. (a) PAM-2 at 30 Gb/s, (b) PAM-2 at 50 Gb/s, (c) PAM-4 at 60 Gb/s, (d) PAM-4 at 80 Gb/s, (e) PAM-4 at 100 Gb/s, and (f) PAM-4 at 112 Gb/s. 

Eye diagram after 2 km. (a) PAM-2 at 30 Gb/s, (b) PAM-2 at 50 Gb/s, (c) PAM-4 at 60 Gb/s, (d) PAM-4 at 80 Gb/s, (e) PAM-4 at 100 Gb/s, and (f) PAM-4 at 112 Gb/s. 

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We present a silicon photonic traveling-wave Mach–Zehnder modulator operating near 1550 nm with a 3-dB bandwidth of 35 GHz. A detailed analysis of traveling-wave electrode impedance, microwave loss, and phase velocity is presented. Small- and large-signal characterization of the device validates the design methodology. We further investigate the pe...

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... length by 5 GHz [12]. Fig. 8(d) demonstrates the extracted characteristic impedance of the TW-MZM from S-parameters under different bias voltages. The measured characteristic impedance has excel- lent agreement with the calculated results presented in Fig. 3(b), with less than 4% variations from the calculated values. The predicted impedance dispersion at higher frequencies is clearly observed in the measured results; however, over the 50 GHz frequency range, the impedance only varies by 2 . Moreover the EE 6.4 dB and the EO 3 dB bandwidths shown are very close, indicating that the performance of the modulator is not limited by velocity mismatch [12]; this validates the design methodology outlined in Section 2. The observed EE S 21 value of 21 GHz is lower than the calculated value of 25 GHz. This can be due to the neglected radiation losses and higher actual surface resistances. Next, we examine the large signal performance of the modulator using an Agilent wide band oscilloscope. At the input of the modulator a tunable laser with maximum output power of 14 dBm is used. The modulator is operated at the quadrature point using the tunable laser. At this wavelength, the modulated signal power is À 3.8 dBm and a 3 V DC bias voltage is applied using a RF probe to reverse bias the PN junctions. A 10 31 À 1 pseudorandom bit sequence (PRBS) signal generated by a SHF pulse pattern generator is amplified using a wide band microwave amplifier and attenuated by passive RF attenuators to obtain a 4.6 V pp driving signal This signal is then applied to the modulator using a high-frequency RF probe. Fig. 9 shows eye diagrams for 30, 40, 50, and 55 Gb/s with extinction ratios of 11.58, 7.59, 5.35, and 4.30 and SNR of 10.31, 6.55, 4.32, and 3.18, respectively. Eye diagrams, shown in Fig. 9, provide a visual qualitative presentation of the performance of the device. In order to quantitatively evaluate the performance of the modulator, bit error rate (BER) test of the system is performed. To do this, the input laser power to the modulator is set to 14.5 dBm. The output of the modulator is fed to an AC coupled Picometrix PD þ TIA receiver which is then connected to a SHF bit error tester. An error-free ð BER G 10 À 12 Þ operation up to 45 Gb/s is obtained with received power of À 3.5 dBm, limited by the bandwidth of the PD þ TIA receiver. The small optical insertion loss, together with the high optical extinction ratio of the transmission spectra and the high E-O bandwidth make this modulator an ideal transmitter for multilevel modulation formats. In this section we compare the performance of the modulator with PAM-2 and PAM-4 modulation formats over different lengths of fiber to reach a 100G Ethernet transport rate. At the input of the modulator, the same tunable laser with 14 dBm output power is used. The RF driving signal is generated using an AC coupled 8-bit Digital to Analog Converter (DAC) operating at 70 GSa/s. Use of a DAC allows us to apply digital signal processing (DSP) at the transmitter side. Four processes are applied to the waveform. First the symbol stream is up- sampled from one sample per symbol to 70 = R B , where ð R B Þ is the desired symbol rate. Next a root raised cosine pulse shaping filter is applied. Thirdly, to equalize the spacing between modulated optical power levels, the nonlinearity of the power transfer function of the TW-MZM is com- pensated by applying an arcsin function to the waveform. Finally the frequency response of DAC, RF amplifier and TW-MZM cascade is pre-compensated by applying an inverse response function. An amplifier is used to amplify the DAC output to 2.2 V pp which is then applied to the modulator using RF probes. The modulated signal is propagated through 1, 2, and 5 km of Corning SMF 28e þ fiber. The PD þ TIA is used for opto-electrical (O-E) conversion before an Agilent real time oscilloscope serving as an 8-bit Analog to Digital converter (ADC) sampling at 80 GSa/s. At the receiver side, the digital signal processing is performed offline. First the signal is resampled from ADC rate of 80 GSa/s to twice the symbol rate R B . Next a matched filter defined at 2 samples per symbol is applied to the signal. The stream of samples is then fil- tered by a linear FIR filter. To recover the transmitter's clock and to apply symbol decision at the correct sampling instant, a digital clock recovery algorithm is implemented [7]. The output symbols are then used for error counting and to calculate the signal to noise ratio (SNR) and quality factor of the system. The DSP applied at the transmitter and receiver sides is discussed in detail in [7]. Fig. 10 illustrates the block diagram of the transmission system explained above. We present the system performance qualitatively using eye diagrams and quantitatively by measuring BER and SNR. BER measurement is done by error counting. For PAM-N formats, SNR is defined as the ratio of the average signal power over average noise power. Fig. 11 shows eye diagrams for PAM-2 and PAM-4 formats at different baud rate after propagating through 2 km of fiber. In currently deployed metro and long haul fiber optic transmission systems, Forward Error Correction (FEC) is used to significantly lower the BER. Based on OT4U standard [2], a client payload of 100 Gb/s is transmitted at line rate of 112 Gb/s, which includes 6.7% (FEC) over head. A BER measurement below the pre-BER threshold of 4 : 4 Â 10 À 3 results in an output BER G 10 À 15 , viewed as error free transmission in the context of optical transmission. In this paper we assume FEC encoding and decoding at the transmitter and receiver side. All eye diagrams are obtained after receiver DSP. A successful 100 Gb/s PAM-4 post-FEC error-free transmission through 2 km of fiber is achieved in all cases. After 5 km, for the same bit rate of fiber the pre-FEC BER is measured at 4 Â 10 À 3 which is slightly lower than FEC threshold of 4 : 4 Â 10 À 3 . For PAM-2, a maximum of 64 Gb/s transmission was achieved at pre- FEC BER of 1 : 31 Â 10 À 4 which was limited by DAC's bandwidth. Fig. 12 illustrates SNR and BER at various bit rates. We observe that, as the bit rate increases, the SNR and BER performance of the system degrades. The DAC has a 3 dB bandwidth of 15 GHz; however, using Nyquist sampling theory, the DAC can generate frequencies up to 35 GHz when sampling at 70 GSa/s. At higher bit rates the signal has higher frequency content. After digital compensation of the frequency response of the DAC, a signal of larger spectral content will have reduced V pp swing out of the DAC, worsening the RF signal quality. At the receiver side, the same large bandwidth signal will integrate more inband noise power, further deteriorating the SNR. The cumulative effect of transmitter and receiver signal worsening as the symbol rate increases is observed in Fig. 12. The low insertion loss of the device allowed the full operation of the modulator without the need for optical amplifiers, which further differentiates this work from other modulators presented in the literature [5], [7], [12]. In this paper, we present the design and characterization of a low voltage silicon photonic traveling wave modulator. A thorough analysis of the implemented traveling wave electrode is presented. It is shown that using a CPS geometry, it is possible to minimize the mismatch between the microwave phase velocity and optical group velocity by careful design. As a result, the main bandwidth limiting factor is determined to be the microwave loss. A 3 dB electro-optic bandwidth of 35 GHz under 3 V DC bias is demonstrated. We further investigate the performance of the device in a short reach transmission system. By applying digital signal processing on transmitter and the receiver side we obtain a successful 112 Gb/s transmission of 4 level pulse amplitude modulation over 5 km of SMF below pre-FEC hard decision threshold of 4 : 4 Â 10 À 3 . We present an alternative solution to 4 Â 25 Gb/s WDM transmission systems. We demonstrate that higher modulation formats such as PAM-4, together with digital signal processing can be used to achieve 100 Gb/s transmission on a single wavelength. The authors gratefully acknowledge CMC Microsystems for enabling fabrication and providing access to simulation and CAD ...

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... Néanmoins, l'ajout de tels convertisseurs dans les émetteurs-récepteurs hauts-débits peut représenter un coût énergétique et économique important [81]. Plusieurs de ces démonstrations reposent sur la structure communément appelée series-push-pull MZM ou SPP-MZM [91,92,94]. Cette dénomination désigne un agencement particulier des régions dopées d'un MZM, où deux régions dopées de chaque bras sont réunies en une seule (par exemple les deux régions P sont mises en commun, Fig. 3.5 [92]). ...
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