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Abstract— As integrated circuit technology advances,

implantable biotelemetry devices will be able to handle more

complex functions. Their flexibility and performance can be

increased by incorporating multi-rate and multi-modulation

schemes. The Class-E transmitter has been a primary choice so

far in implantable devices as it provides low-power design with

high-power efficiency. This paper investigates the design

procedures of class-E amplifier for different modulation

schemes in biotelemetry systems. A Class-E transmitter circuit

that produces OOK, FSK and PSK modulated signals has been

designed, optimized and analyzed in terms of a second order

system for general implantable electronics.

I. INTRODUCTION

lthough the Class-E amplifier is commonly used for

wireless power and data transfer in biomedical

telemetry systems, it has mostly been optimized for

power signals (pure sinusoidal signals) in literature

[14],[7][8]. The focus of this paper lies in applying the

Class-E amplifier to different modulated signals in

biomedical telemetry devices. The paper also shows that a

class-E system can be analyzed better with the second order

system parameters.

Two transmission standards which are of particular

interest are the Medical Implant Communication Service

(MICS) standard and 27 MHz ISM standard. The MICS

standard has recently been allocated by international

communication authorities to provide small size, low-power,

faster data transfer and a longer communication range for

future implantable devices. The standard specifies a

frequency range between 402MHz and 405MHz, with a

maximum bandwidth of 300kHz at any one time, and a

maximum transmission power of 25µW [9][10]. The ISM

standard involves a number of allowable frequencies, one of

which is 27 MHz. This frequency was of particular interest

due to the fact that some of the recent implantable designs

have been targeting some frequencies around this value to

accommodate a higher data rate transmission [5][6]. Thus,

transmitter circuit given in herein is being implemented for

these two medical bands.

II. ANALYSIS OF CLASS-E AMPLIFIER

The reason why the Class-E amplifier is used in

implantable biotelemetry is because of the fact that when

used as a modulator, it eliminates the need for a mixer,

which is a power hungry block and used in many

Manuscript received April 16, 2007. This work was supported in part by

the Australian Research Council (ARC) under Discovery Projects Grant.

A. N. Laskovski and M. R. Yuce are with the School of Electrical

Engineering and Computer Science, University of Newcastle, Callaghan,

NSW 2308, Australia (e-mail: mehmet.yuce@newcastle.edu.au ).

Fig. 1 The Class E Amplifier

conventional modulators. Fig. 1 shows a generic schematic

of a class-E oscillator. The MOS device (i.e. M1 transistor)

acts as a switch and is driven by a digital signal at the carrier

frequency fc. The capacitor C2 connected in series with L acts

as a DC blocking capacitor and its value should be small

enough to be nearly resonant with L at the carrier frequency

fc [7]. For an inductive link, the inductor L is used as a

physical coil to transfer data and power to the secondary side

through a magnetic coupling. The resistor R represents the

parasitic resistor of the inductor and the reflected resistance

from the loading of the secondary site. When an antenna is

used, this load resistor will be the impedance of the antenna.

The transmitter power can be variable by changing the power

supply VDD or the resistor at the load R. Thus any required

power level specified by medical standards can easily be

adjusted using these components.

The load network, characterized by C1, C2 and L converts

the digital input signal into a sinusoidal output with a zero

DC offset. By considering the load network as a second

order system, it is possible to gain an understanding of how

to manipulate the output of the circuit to achieve as high an

output as possible with a less power consumption. The

transfer function of the load network can be calculated as:

( )

s

2

2

1

LCL

R

ss

L

R

s

VH

v

v

DD

in

out

++

==

(1)

where vin is the voltage across the capacitor C1.

A digital square wave is represented as a series of positive

and negative step inputs, and after conducting an inverse

Laplace transform, the output signal for one half period is

determined to be:

Analysis of Class-E Amplifier with Mixed Data Modulation

for Biotelemetry

Anthony N. Laskovski and Mehmet R. Yuce, Member IEEE

A

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Fi g. 2. A low-power transmitter design using Class-E amplifier for PSK, FSK and OOK signals

(

2

1

ζ−

)

2

2

1 sin)(

ζω

t

ω

ζω−

−=

n

t

n

DDout

e RCVtv

n

(2)

2

1

LC

n=ω

and

L

CR

2

2

=

ζ

.

where ωn is the natural frequency and ζ is the damping

factor. The output signal and therefore efficiency may be

manipulated by selecting a particular damping coefficient (ζ)

and damped frequency (ω = 2πfc) where

Simulation results based on the effect of these parameters

will be presented in Section IV.

According to [8], the efficiency of the Class E amplifier is

the RMS power seen at R divided by the input DC power, as

shown in (4). It assumes that the output voltage is sinusoidal.

Using (2), the efficiency can be given by (5).

2

1 ζ−ωω=

n

(3)

DDDD

2

o

I RV

V

2

2

=η

(4)

t

DD

DD

n

e

LCRC CRLC

RC

2

2

I

V

ζω −

η

−+

=

22

2

2

2

44

2

(5)

Since the RMS voltage of PSK and FSK signals are the same

as that of a sinusoidal signal, (5) has also been used to

determine the efficiency of the PSK and FSK modulated

signals. In the case of OOK, the efficiency is simply halved.

III. A CLASS-E TRANSMITTER WITH MIXED MODULATED

SIGNALS

In this section, we present a transmitter circuit using a

Class-E amplifier that provides a low-power implementation

of three different modulated signals: On-Off Keying (OOK),

Phase-Shift Keying (PSK) and Frequency Shift Keying

(FSK). The PSK modulated signal is generated by applying

the carrier and the data signals to the gate of transistor M1

through an exclusive-OR as shown in Fig 2. The data is

directly applied to M3 for an OOK modulated signal. FSK

can be achieved by generating a variable frequency signal at

the input of the Class-E amplifier with a digital voltage

controlled oscillator. Note also that the circuit does not need

to be altered to allow different data rates (multi-rate

transmission).

IV. CIRCUIT SIMULATIONS

Simulations were conducted on the basic class-E amplifier

of Fig.1 in order to determine the optimal circuit parameters

for the proceeding modulation simulations. The circuit is

simulated using Berkeley BSIM4v4 spice models. Two

parameters were examined closely based on the theory of

second order systems. The first parameter to be considered

was the damping coefficient ζ∈(0,1). A transfer function

tuned to 403MHz with a load resistance of 500Ω was

simulated in MATLAB under varying conditions, and step

response plots of two of the outermost coefficients are

presented in Fig. 3. The higher curve represents a damping

coefficient of 0.9, meaning that the signal is over-damped.

The lower curve is the result of a damping coefficient of 0.1,

meaning that it was under-damped. This explains why the

under-damped signal oscillated for several cycles of its

oscillation frequency. It is evident that the damping

coefficient makes a considerable difference to the output

signal of the Class E circuit.

Fig. 3. A Step response with damping ratios of 0.1 and 0.9 applying to the

higher and lower curve respectively.

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This effect was further investigated through a series of

circuit simulations in SABER with a digital clock input

signal. The load resistance R was kept constant at 500Ω, as

was the forced oscillation frequency of 403MHz. The

relationship between the damping coefficient and circuit

efficiency is presented in Fig. 4, showing that the lower the

damping coefficient, the higher the efficiency of the circuit.

As an under-damped system with periodic step inputs,

superposition occurs cycles after each change in the digital

carrier signal. Due to the fact that there is no change in

phase, the oscillations constructively superimpose to

increase the overall output power of the signal and therefore

higher circuit efficiency at lower values of ζ.

The next parameter considered was the output resistance R

and the MOSFET channel lengths and widths. R was varied

while ζ was kept constant at 0.1 and the forced oscillating

frequency at 403MHz. The simulations were conducted with

three different transistor settings. From the results plotted in

Fig 5, it is evident that the efficiency of the class E circuit

peaks according to the relationship between R and the

parameters of the transistors in use. In reality R will vary

according to transmission conditions, however the circuit

can be optimized for a particular value of R which is likely

to be the most common load resistance to the class-E

transmitter.

Fig. 4. The effect of the damping factor ζ on efficiency (η) with R =500Ω,

VDD= 3V, lchannel=500nm, widthp-channel=20µm, widthn-channel=10µm.

Fig. 5. The effect of the load resistance R on efficiency η with a constant ζ

of 0.1 and VDD of 3V and varying the scale of the MOSFET channel widths,

widthp-channel/widthn-channel while keeping the channel length 500nm.

It was mentioned in section II that C1 was approximately the

same as C2, and that C2 serves the purpose of blocking DC

volts from entering the output. A number of simulations

have been conducted to show a relationship between C2 and

C1. C1 was kept constant at 0.157 pF and C2 was varied as

shown in Fig. 6. It may be observed that when C2 is close to

the value of C1 a maximum transfer of energy takes place.

This is because the voltage at the junction between C2 and L

approaches zero. As C2 exceeds C1 the efficiency drops and

then plateaus to a constant value, meaning that C2 needs to

be the same value as C1 (which is in resonance with L) to

successfully drive the average voltage across C1 to 0V at the

highest frequency. Fig. 6 also shows the variation of C1

while keeping C2 constant. This plot shows the efficiency

rapidly decrease when C1 is increased to be higher than C2.

This occurs because as C1 exceeds the value of C2, C2 is no

longer large enough to drive the average voltage between L

and C2 down to zero, meaning that the excess DC energy is

dissipated through R, which is detrimental to the efficiency

of the circuit.

Fig . 6. Variations in C1 and C2 with VDD=3V, ζ=0.1, R=500Ω, L=982nH,

MOSFET channel length 500nm, widthp-chan/widthn-chan=20µm/10µm. a) The

effect of varying C1 while keeping a constant C2 at 0.157pF. b) The effect

of varying C2 while keeping a constant C1 at 0.157pF.

An important factor in complying with the MICS standard

is for the transmission signal to remain below 25µW. Thus,

the effect of VDD was investigated. It is evident from Fig. 7

that for lower values of VDD, circuit efficiency is highly

dependant on the input power, but as VDD exceeds a certain

value, its value becomes less significant. By comparing Fig.

5 with Fig. 7 it is evident that balancing VDD and R is

Fig. 7. The effect of the damping factor VDD on efficiency (η) with R

=500Ω, ζ= 0.1 (C1=C2=0.157pF, L=982nH), lchannel(n,p)=500nm, widthp-

channel=20µm, widthn-channel=10µm.

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important in controlling the efficiency of the circuit and also

the output power such that it complies with the MICS

standard.

It was predicted in Section II that PSK would have an

almost identical efficiency to an un-modulated signal. The

Class E transmitter given in Fig. 2 was simulated in PSK and

OOK (plots shown in Fig. 8 and 9 respectively) modes to

indicate the efficiency of each method.

The simulation results show that the efficiency of the un-

modulated circuit (with VDD = 3V, ζ= 0.1 and R =500 Ω) is

66.9%, while the efficiency of the PSK circuit is 25.9%. The

difference may be explained by the presence of an XOR

gate, which comprises 20 additional transistors. The

efficiency of the OOK signal was simulated to be 32.7%,

which is just less than half the efficiency of the un-

modulated circuit. Considering the fact that the XOR was

not in operation during the OOK or the un-modulated

circuit, the results are close to what was expected. An FSK

signal was simulated with the two signal frequencies

269MHz and 403MHz. The efficiency of this circuit was

29%. As shown in Fig 10, there is a noticeable difference

between the two frequencies; however the circuit has been

optimized for 403MHz. This is the reason why the efficiency

is not similar to the un-modulated value of 66.9%.

The transmitter consumes 75 µW at 0.8 VDD (with the

output signal, Pout = 2.25 µW) and 2.2 mW at 3V (Pout =

941µW) when operating at the MICS band. To meet the

MICS regulation for the transmitted power, the VDD should

be kept low. In the case of 27 MHz ISM, the power

consumptions 9.16 µW (Pout = 0.243µw), 18.5 µw (Pout =

0.592µW), 479 µW (Pout = 44µW), 4.22 mW (Pout =

300µW) for supply voltages of 0.8V, 1V, 3V and 5 V

respectively. The damping factor was ζ = 0.08 and data rate

was 300 Kbps.

V. CONCLUSION

A data transmitter using class-E oscillator for telemetry

has been optimized with a second order system. A circuit has

been designed to verify the effect of the damping factor on

the operation of Class E. A transmitter circuit is designed

based on the recently available MICS band. Generation of

modulations schemes such as OOK, FSK and PSK have

been discussed. When modeling the load network of the

amplifier as a second order system, several parameters were

noticed to make a considerable difference to the output of

the circuit.

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[5]

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[8] F. H. Raab, "Effects of circuit variations on the class E tuned power

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[9] FCC Rules and Regulations, “MICS Band Plan,” Table of Frequency

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[10] Australian Communications Authority, Radio Frequency Planning

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[6]

[7]

Fig. 8. A simulation plot of the output of the PSK circuit (403MHz)

Fig. 9. A simulation plot of the output of the OOK modulated circuit.

a)

b)

Fig. 10. Output plots of a FSK signal. a) Time transient signal. b)

Frequency spectrum