Design of an LC-VCO with One Octave Tuning Range
Andreas Kämpe and Håkan Olsson
Radio Electronics LECS, Department Microelectronics and Information Technology, KTH
Electrum 229, 164 40 Kista
Abstract - This paper presents the design of a wideband,
fully integrated LC-VCO. The architecture is fully
differential and has a tuning range from 1.2 GHz to 2.6
GHz. The phase-noise varies within the tuning range
from -138 dBc/Hz to -128 dBc/Hz at 1 MHz frequency
offset. The VCO is implemented in a 0.18µm CMOS
process using a 1.8 V supply. The circuit, including the
bias, consumes only 3.8 mW at 2.6GHz and 8.5mW at 1.2
As more and more wireless standards, such as WLAN,
DVB and UMTS. are introduced, an elegant solution
would be a multi-standard transceiver . Therefore
there is a need for extremely wideband circuit blocks for
the RF front-end. The VCO is a key building block in
frequency synthesizers. A challenge is to design a
VCO with a wide tuning range maintaining a low phase-
noise and power consumption. The design is further
complicated by the lack of high quality monolithic
inductors and the small capacitance variation of the
varactors for low control voltage, limitated by the
Oscillators without LC-tanks such as ring oscillators
can achieve a very wide tuning range but they suffer
from very high phase noise or high power consumption
, , . On the contrary a fully integrated LC-VCO
can be made with a low phase-noise and with relatively
low power consumption, but they usually suffer from a
narrow tuning range , .
In this paper, a fully integrated LC-VCO with a tuning
range over one octave is presented. It also exhibits low
phase-noise and low power consumption. The large
tuning range is achieved by the use of an array of
For RF transceivers, the LC-type oscillator is superior
in phase noise due to the band pass filtering of the LC
resonator. Harmonics are attenuated and any sideband
noise is reduced.
The VCO’s output frequency is tuned by on-chip
varactors. These varactors should have low parasitic
capacitance and wide tuning range to cope with
process variations. For an LC-VCO to achieve a tuning
range of one-octave, it would require a capacitance
tuning of two octaves, due to the square dependency
of the frequency to capacitance:
Where fmin and fmax denote the highest and lowest
oscillation frequency, tuned by a varactor with a
capacitance that can be varied from Cmin to Cmax. The
tuning capacitor has to have a Cmax/Cmin ratio even
larger than 4 to compensate for the capacitive
parasitics Cp of the negative resistance and the
inductor. Designing an on chip varactor with this large
Cmax/Cmin ratio in a low voltage CMOS process is not
easy, and would result in a large varactor sensitivity
(VCO gain). This is not recommended, since it would
degrade the phase noise performance of the VCO. Low
frequency noise and interference reaching the varactor
would phase-modulate the VCO and be up-converted
to the carrier frequency increasing the phase noise.
3 THE CAPACITOR ARRAY
Achieving a large Cmax/Cmin ratio while having a small
VCO gain can instead be solved by using an array of
switched capacitors as shown in Fig. 1.
Fig. 1: Switched capacitor array.
The switched capacitors are used as band selectors or
as coarse tuning. For fine tuning, a varactor is used.
The switches consists of NMOS transistors due to
their higher transconductance, but there is a tradeoff
with the transistor size, between loss and capacitive
load. This translates into either a reduced power
consumption or an increased tuning range.
For small losses, the drain source resistance (RDS(ON))
should be reduced
transconductance. Thus a wide transistor with
minimum gate length and a large overdrive (Vgs-Vt)
should be used. For a small capacitive load, the Cgs and
Cgd have to be minimized, requiring a narrow transistor
with minimum gate length. The capacitor array is
shown in Fig. 2
by maximizing the
Fig. 2: Capacitor array.
The capacitors on both sides of drain and source are
used for band switching, but they also act as coupling
capacitors isolating the biasing voltage from the
negative resistance. The drain and source are biased
via resistors. When the switch is on, the biasing is set
to 0V and the gate to 1.8 V. This maximizes the
overdrive resulting in a reduced RDS(ON). When the
switch is off the bias is set to 1.8 V and the gate is at
0V. This reduces the voltage dependent Cgs and Cgd
capacitance by 20%. The increased overdrive makes it
possible to use smaller transistors which reduce the
capacitive load without increasing the losses.
The oscillation amplitude is determined by the negative
resistance and the load impedance of the LC tank. At
resonance the LC tank has an impedance
Thus the oscillation amplitude increases with the
oscillation frequency. If the tuning range is large, e.g.
one octave, the oscillation amplitude will vary
significantly between fmax and fmin. This requires an
adjustable negative resistance, and is achieved by
changing the biasing
transconductance of gmn and gmp in the negative
resistance. As the control voltage and thereby the
frequency is increased, the biasing current is
current, affecting the
decreased. This bias control guaranties startup, and
ensures constant oscillation amplitude independent of
the oscillation frequency.
4 VCO ARCHITECTURE
All the blocks in the VCO (inductor, varactor, cap-
array, negative resistance) are fully differential to
reduce the sensitivity to power supply variations and
substrate interference. Fig. 3 shows the block diagram
of the VCO. The negative resistance consists of a
cross-coupled complementary structure of n and p-
channel-transistors.The oscillation frequency f0 is
controlled by the LC-tank. The array of capacitors is
switched in or out in discrete frequency steps, while
the varactor is used for fine tuning.
Fig. 3: Block diagram.
The cross-coupled complementary structure with n-
MOS and p-MOS transistors was chosen due to its
differential operation, large output swing and low
phase-noise for a given current. An (n&p-core)
operated in the current-limited region  can achieve
the same oscillation amplitude but with less current
than an n-core structure. The varactor consists of four
accumulation-mode transistors in an anti-parallel
configuration, shown in fig. 4. This enables differential
tuning. The complete varactor has a Cmax/Cmin ratio of
Fig. 4: MOS varactor
5 THE INDUCTOR
In an on chip LC-oscillator the inductor is the dominant
source of loss, but is compensated by the negative
resistance. The Q of an inductor can be increased by
using a differential coil instead of two single coils. The
coupling factor increases the inductance but with
unaffected series resistance.
A fully differential inductor was designed. It has a
diameter of 340 µm and consists of three turns, see Fig.
Fig. 5: Inductor layout.
The inductor is designed by stacking the three top
metal layers M6, M5 and M4 on top of each other.
They are then all connected in parallel to minimize the
series resistance, thereby reducing the phase noise,
The disadvantage of this triple layer inductor is the
reduced tuning-range. The metal layers M5, M4 and
lower are closer to the substrate which increases the
The inductor was designed using Electromagnetic
(EM) simulators such as ASITIC  and ADS. The
geometry and size was optimized using ASITIC, then
fine tuned and simulated with ADS. Simulations
(shown in Fig. 6.) resulted in an inductance around 3.6
nH Between 1.0 and 3.0 GHz. The Q varies from 10.5 to
188.8.131.52 1.82.02.2 2.4 2.6 2.81.03.0
Fig. 6: Inductor performance.
The simulated S-parameter data was fitted to a lumped
model of a transmission line, shown in Fig 7.
Fig. 7: Inductor model.
The extraction of the simulated data to this model
resulted in less than 2% error from 1.2 GHz to 3GHz.
The VCO was implemented in a 0.18µm CMOS process
and verified in simulations using Cadence SpectreRF.
This Resulted in a tuning range from 1.2 GHz to 2.6
GHz, shown in Fig. 8.
000 001010 011
Frequency of oscillation
Differential control voltage (V)
Fig. 8: tuning range.
100 101 110111
The phase-noise at 1 MHz frequency offset varies from
-138 dBc/Hz at 1.2 GHz, to -128 dBc/Hz at 2.6 GHz, The
circuit including the bias, consumes only 3.8 mW at 2.6
GHz versus 8.5mW at 1.2 GHz. The current
consumption of the VCO at 2.6 GHz is 2.1 mA with a
core current of 1.4 mA. The biasing circuitry adds 0.7
To compare the performance of various VCO’s, a
common approach is to use a figure of merit (FOM),
FOM normalizes the phase noise to offset frequency,
oscillation frequency and power consumption PVCO.
This results in a FOM of -190 dBc/Hz for this design. In
The table below, some VCOs from litterature are listed.
Our design has an overall very good performance
expressed in FOM and superior if the wide tuning
range is taken in account.
 0.13 SOI
* Quadrature VCO
In this paper we have presented a low power, low-
phase noise VCO having a tuning-range over one
octave (1.2 to 2.6 GHz). The VCO is completely
differential (even the tuning is differential). The VCO is
implemented in a 0.18µm CMOS process using a 1.8 V
supply. Simulation at 2.6 GHz oscillation frequency,
showed a phase noise of -128 dBc/Hz at 1 MHz
frequency offset. The VCO, including the bias,
consumes only 3.8 mW.
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