Vol. 119 (2011)
ACTA PHYSICA POLONICA A
Physical Aspects of Microwave and Radar Applications
Antenna Beam Broadening in Multifunction
Phased Array Radar
R. Fatemi Mofrad∗and R.A. Sadeghzadeh
Electrical & Computer Engineering Department, K.N. Toosi University of Technology, Tehran, Iran
A phased array antenna is designed for multifunction phased array radar simulation test bed.
element pattern, mutual coupling between elements, phase quantization, amplitude and phase error and elements
failure rates on array pattern are discussed.Target angle measurement and side lobe cancelling, in order to
reduce jamming power through side lobes, is illustrated in this antenna. Also antenna beam width is broadened
with different methods and compared with narrow beam characteristics.
ening factors, beam broadening may lead to a better coverage and power efficiency relative to narrow beam antenna.
It is shown that, for special broad-
PACS: 84.40.Ba, 84.40.Xb, 84.40.Ua
Phased array antennas have matured rapidly in recent
years and this technology is set to become the norm in
complex and advanced radar systems.
steer the radar beam electronically allows a combination
of functions, such as tracking, surveillance and weapon
guidance, which were traditionally performed by dedi-
cated individual radars. This new type of radar is called
multifunction array radar (MFAR). In these radars, ef-
fect of different antenna beam characteristics (e.g. beam
width) or tracking algorithms on the overall radar per-
formance need to be done based on realistic simulations,
because these sophisticated radars cannot be tested com-
pletely in real world. These realistic simulations should
have two main properties: first to include different as-
pects of real operational scenarios facing MFAR as much
and accurate as possible. The second is that these sim-
ulations should provide the facility to model different
part of a MFAR in order to evaluate the performance
of each section in the radar as a whole system (to con-
sider the interaction between subsystems). MFAR simu-
lation test bed is a software tool for MFAR designers to
design and evaluate the performance of such kind of so-
phisticated radars . In the MFAR simulation test bed,
active phased array radar, with specification in Table I,
is considered as a pilot for different radar resource man-
agement, target tracking and beam forming algorithms
comparison and development. In this simulation test bed,
transmitting and receiving chain, antenna structure and
signal processing algorithms are fixed. User may write his
or her own radar resource management, target tracking
and beam forming algorithms and after defining appro-
priate operational scenarios, assess results of the designed
algorithms. The most important part of this simulation
test bed is a phased array antenna. That is an active
The ability to
∗corresponding author; e-mail: firstname.lastname@example.org
phased array with about 5000 elements. It is assumed
that digital beam forming is possible at element level and
so designer may design appropriate beam forming algo-
rithms and evaluate the results on radar performances.
angle tracking accuracy
antenna scanning range in az. and el.
antenna tilt angle
sum pattern side lobe level
difference pattern side lobe level
number of T/R modules
In this paper, a narrow pencil beam with capability to
measure azimuth and elevation angles of target and side
lobe cancelling in the presence of jamming is designed
for this simulation test bed. Effect of element pattern,
phase quantization, amplitude and phase error, elements
failure rates on array pattern and mutual coupling be-
tween elements, are presented. The narrow beam width
is usually designed to meet tracking requirements of a
MFAR (resolution and accuracy). Search of a large area
by this narrow beam becomes too time consuming. In
these occasions antenna beam broadening is useful.
In this paper antenna beam width is broadened and
compared with narrow beam characteristics. It is shown
that, for special broadening factors, beam broadening
may lead to a better coverage and power efficiency rela-
tive to narrow beam antenna.
R. Fatemi Mofrad, R.A. Sadeghzadeh
2. Planar array structure
For scanning in elevation and azimuth planar array
should be used. Details of planar array theory are given
in available texts [2–4]. All the antenna elements have a
certain element pattern e(θ, ϕ). This is multiplied with
the array factor resulting in the final antenna pattern
E(θ, ϕ) (field strength):
E(θ,ϕ) = e(θ,ϕ)f(θ,ϕ).
Array antennas often use rectangular or triangular
placement of elements. Usually an array with a trian-
gular grating, particularly an equiangular one, is pre-
ferred. This arrangement reduces the number of elements
by nearly 13% and increases the area associated to each
element . Designed array has a rectangular aperture
with 80 elements in azimuth and 64 elements in elevation
direction, so an asymmetrical pattern in H and E planes
The maximum distance between radiators of a scan-
ning antenna array is related to the maximum angle of
deviation of the pattern. For ±45◦electronic steering,
distance between elements is designed to be equal to
0.58λ which will not generate grating lobes. With this se-
lection, dimension of aperture will be: 4.64 m × 3.71 m.
There are many weighting window types with different
properties. In the Taylor tapering there is a better trade-
off between decrease in the side lobe level and broadening
the main beam . Amplitude weighting, at T/R module
level leads to more complication of the modules (control-
lable attenuators and phase shifters will be needed).
Array pattern with and without Taylor taper-
For difference pattern generation, the aperture can be
derived with the Bayliss tapering . The Bayliss ta-
pering is applied at output of 160 sub-arrays each with
4 × 8 elements . The Bayliss amplitude tapering at
sub-array level will increase side lobe level of difference
pattern relative to sum pattern. Usually target detection
and acquisition is performed by sum pattern, so higher
side lobe level in the difference pattern would not cause
false alarm due to clutter or other unwanted targets.
The radiated pattern of antenna array with and with-
out amplitude tapering is shown in Fig. 1. With taper-
ing, side lobe level will decrease by about 16 dB. In the
transmission mode the aperture of antenna array should
operate with uniform amplitude illumination. In this way
maximum possible power of transmitter modules will be
derived with maximum efficiency.
3. Angle measurement
In this antenna, monopulse likelihood function is used
for direction estimation of target . In likelihood es-
timation there is a tapering to form difference pattern.
This tapering equals to the coordinates of element which
are distributed symmetrically around axis. This tapering
in comparison to the Bayliss tapering produces sharper
difference pattern and increases accuracy of angle estima-
tion but with higher side lobes. By multiplying Bayliss
tapering with this tapering, sharpness of difference pat-
tern will decrease but lower side lobes will be produced.
The characteristics of this mixed tapering are a tradeoff
between selection of the Bayliss and likelihood estimation
tapering. An example of direction estimation is shown in
Fig. 2 for beam direction at 35◦in azimuth and elevation.
Figure 3 depicts standard deviation of target direction es-
(SNR = 10).
Direction estimation for beam direction at 35◦
tion (in degree) as a function of signal to noise ratio
(SNR) (in dB).
Standard deviation of target direction estima-
Antenna Beam Broadening in Multifunction Phased Array Radar
timation as a function of SNR. These results well match
with similar results in .
4. Digital phase shifter
Usually digital phase shifters, characterized by the
number of bits M, are used for beam steering. M de-
versus scan angle (in degree) with discrete phase shifter.
Increase of side lobe level ratio (SLLR) (in dB)
termines the residual phase error within the interval:
∆ϕ = ±π/2M. The main effect of discrete phase shift-
ing is on side lobes. In Fig. 4 simulation results show
increase in side lobe level (SLL) with 4 type of discrete
phase shifters. These results well match with those in .
According to these results, 6 bits phase shifter has satis-
5. Element pattern effect
Parameter of the designed microstrip patch antenna
at 3 GHz are [6, 7]:
L = W = 0.39λ0,
d = 0.06λ0.
In Eq. (2), d is height of substrate, εris dielectric con-
stant of substrate and L and W are dimensions of patch.
Half power beam width of element is 80 degree, so there is
3 dB loss of effective radiated power (sum of gain and ra-
diated power) at ±40 degrees steering angles. The effect
of element pattern loss together with beam broadening
loss due to beam steering is depicted in Fig. 5. This result
well matches with those in .
6. 20% failure rate of elements
When 20% of elements accidentally turn off, side lobe
level will increase. Figure 6 compares array pattern in
this condition with main array pattern. Simulation re-
sults show that in the worse condition there is 22 dB
increase in side lobe level of sum pattern and 14 dB in
difference. The effect of failure on the gain and beam
direction is negligible.
(in dB) with scan angle (in degree) because of both el-
ement pattern and beam broadening in E-plane.
Decrease of effective radiated power (ERP)
gree) for 20% failure of elements (160 subarray).
Array pattern (in dB) versus scan angle (in de-
7. Amplitude and phase error
Simulation results show that with 40% amplitude tol-
erance there is 2 dB increase in side lobe level. This
error is independent of steering angle. Also it has been
shown that destructive effect of phase error is more than
amplitude error. Its main effects are loss in the effective
radiated power and increase in side lobe level. According
to simulation results, maximum tolerable phase error is
about 0.5 ≈ 0.25 rad.
Decrease of ERP (in dB) with increase of phase
R. Fatemi Mofrad, R.A. Sadeghzadeh
scan angle (in degree) with increase of phase tolerance.
Increase in main beam ripples (in dB) versus
Figure 7 shows loss of maximum effective power with
increase in phase error. Phase error causes ripples in
main beam and with increasing in phase error these rip-
ples may be seen clearly.
This effect is shown in Fig. 8. In this figure decrease
of effective radiated power is also shown. This power
distributes in side lobes. Side lobes that are far from
bore side are higher in comparison with pattern that has
no phase error.
8. Side lobe canceller
The Widrow–Hoff least mean square (LMS), side lobe
canceller (SLC) algorithm, which is a closed loop digi-
tal algorithm described by [2, 9] was implemented in the
simulation test bed. The benefit of using the SLC can be
measured by jammer cancellation ratio (CR), defined as
the ratio of the output noise power with and without the
SLC. For instance the CR value obtained in this simula-
tion test bed with one channel SLC is about 30 dB for a
jammer at 14.5◦azimuth angle.
tern (continuous line) with one auxiliary antenna for
jammer incoming at 14.5◦azimuth.
Adopted pattern (dotted line) and main pat-
Figure 9 shows adapted antenna pattern obtained by
this LMS algorithm. The gain margin of auxiliary SLC
antenna with respect to the side lobe gain of the radar
antenna in the jammer direction is an important param-
eter. A large value of the gain margin in the steady-state
of an adaptive SLC would be desirable. However, in the
transient state of the SLC a low value of gain margin
would be advisable. A compromise value is around 10 dB
for the gain margin. In Fig. 9 half power beam width of
auxiliary antenna cover ±15◦around bore sight and so
its gain will be 16 dB that is 6 dB more than side lobe
9. Mutual coupling effect
Microstrip patch is a main candidate to be used as
the elements of integrated phased arrays. Such arrays
may include active devices on the same or on a parallel
substrate, integrated monolithically or in a hybrid fash-
ion. Design of these arrays depends on understanding of
the effects of substrate thickness, dielectric constant, and
grid spacing on the scan performance of the beam. The
scan performance means the active reflection coefficient
magnitude, with the array matched at broadside and is
directly related to the active element pattern. For an ar-
ray of printed patch antenna, scan blindness is possible
whenever the wave number coincides with the propaga-
tion constant of a surface wave on the structure. Scan
blindness will occur if the following three conditions are
1) The propagation constant equals surface wave prop-
2) The grid spacings dx, dy are such that the equality
of propagation constant in (1) occurs for values of u, v
in real space.
3) The pole of TM (TE) surface wave in (1) is not
cancelled by a zero value of kx (ky).
Mathematically, condition (1) can be expressed as
In Eq. (3) λ0 is the free space wavelength. Physically,
the cancellation condition referred to in (3) means that
the polarization of the array is such that the particular
TM or TE surface wave cannot be excited.
Figure 10 shows blind point versus space between el-
ements of designed arrays (with patch length and width
L = W = 0.39λ0, substrate thickness d = 0.06λ0, and
substrate permittivity εr = 2.1).
sition moves toward broadside by increase in the space
between elements. With dx = dy = 0.58λ, the array
has a scan blindness in the E-plane at 44.76◦.
dx = dy = 0.51λ blind spot moves to 70◦that is a better
The blind spot po-
Antenna Beam Broadening in Multifunction Phased Array Radar
Blind angle (in degree) versus space between
0.51λ, dy = 0.51.
Reflection coefficient versus scan angle dx =
choice for an array with ±45◦scanning angle. In this
case the magnitude of the reflection coefficient at 45◦is
about 0.3 and then the loss of power gain will be only
Figure 11 shows the magnitude of the reflection coef-
ficient of a patch in designed array with spacings dx =
dy = 0.5λ. These results well match with those in .
10. Beam broadening
As was said before, in the transmission mode the aper-
ture of antenna array should operate with uniform ampli-
tude illumination. In this way maximum possible power
of transmitter modules will be derived with maximum
efficiency. So transmit beam broadening should be done
with uniform amplitude and phase only tapering. Broad-
ening factor is defined as broadened 3 dB beam width to
narrow 3 dB beam width. In the MFAR simulation test
bed, a broadening factor of 4 in elevation angle is desired
so broadening should be done by linear array with 64 ele-
ments. Gradient search algorithm (GSA)  is used for
broadening of the linear array.
In Fig. 12 pattern of broadened and narrow beams are
presented and compared.
Figure 13 shows required phase of elements for beam
broadening. This phase is calculated by GSA. A good
measure of comparison between two patterns, is beam
Fig. 12. Narrow and broadened beam patterns.
Phaseof elementsin broadenedbeam
power efficiency which is defined as the ratio of trans-
mitted power in the 3 dB main lobe of broadened beam
to narrow beam. With beam broadening, it is possible
to radiate more power into space so a beam power ef-
ficiency more than one is achieved. Beam efficiency for
three linear arrays:
— a linear array with 64 elements and λ = 0.1,
— a linear array with 64 elements and λ = 1,
— a linear array with 80 elements and λ = 1
is presented in Fig. 14 for broadening factors up to 15.
As is clear, for broadening factors more than 2.5, beam
power efficiency becomes more than 1 (0 dB).
With beam broadening, space coverage is also in-
creased. In Fig. 15 from  space coverage of a narrow
beam of width equal to 1 is π/4. This coverage for a 1×4
broadened beam is 3 + π/4. There is a coverage differ-
ence of 3 − 3π/4 between coverage of two beams. In the
other word, broadened beam approximately cover 20%
(0.6438/π) of environment. This improvement in cover-
age is for none overlapped beams. For overlapped beams
this improvement will increase appropriately.