Integrated Submm Wave Receiver:
Development and Applications
Valery P. Koshelets, Manfred Birk, Dick Boersma, Johannes Dercksen,
Pavel Dmitriev, Andrey B. Ermakov, Lyudmila V. Filippenko, Hans Golstein,
Ruud W.M. Hoogeveen, Leo de Jong, Andrey V. Khudchenko,
Nickolay V. Kinev, Oleg S. Kiselev, Pavel V. Kudryashov, Bart van Kuik,
Arno de Lange, Gert de Lange, Irina L. Lapitsky, Sergey I. Pripolzin,
Joris van Rantwijk, Avri M. Selig, Alexander S. Sobolev,
Mikhail Yu Torgashin, Vladimir L. Vaks, Ed de Vries, Georg Wagner,
and Pavel A. Yagoubov
Abstract A superconducting integrated receiver (SIR) comprises in a single chip
a planar antenna combined with a superconductor-insulator-superconductor (SIS)
mixer, a superconducting Flux Flow Oscillator (FFO) acting as a Local Oscillator
(LO) and a second SIS harmonic mixer (HM) for the FFO phase locking. In
V.P. Koshelets (?) ? A.B. Ermakov ? A.V. Khudchenko ? N.V. Kinev ? O.S. Kiselev
Kotel’nikov Institute of Radio Engineering and Electronics, Russian Academy of Science,
Mokhovaya st. 11/7, 125009, Moscow, Russia
SRON Netherlands Institute for Space Research, 9700 AV, Groningen, The Netherlands
e-mail: email@example.com; Khudchenko@hitech.cplire.ru
P. Dmitriev ? P.V. Kudryashov ? A.S. Sobolev ? M. Yu Torgashin
Kotel’nikov Institute of Radio Engineering and Electronics, Russian Academy of Science,
Mokhovaya st. 11/7, 125009, Moscow, Russia
D. Boersma ? J. Dercksen ? L.V. Filippenko ? H. Golstein ? R.W.M. Hoogeveen ? L. de Jong ?
B. van Kuik ? A. de Lange ? G. de Lange ? I.L. Lapitsky ? J. van Rantwijk ? A.M. Selig ?
Ed de Vries
SRON Netherlands Institute for Space Research, 9700 AV, Groningen, The Netherlands
M. Birk ? G. Wagner
DLR German Aerospace Centre, Remote Sensing Technology Institute, 82234, Wessling,
S.I. Pripolzin ? V.L. Vaks
Institute for Physics of Microstructure, Russian Academy of Science, Ulyanova 46, GSP-105,
Nizhny Novgorod, Russia
SRON Netherlands Institute for Space Research, 9700 AV, Groningen, The Netherlands
European Organization for Astronomical Research in the Southern Hemisphere (ESO),
Karl-Schwarzschild-Strasse 2, 85748, Garching bei M¨ unchen, Germany
A. Sidorenko (ed.), Fundamentals of Superconducting Nanoelectronics,
NanoScience and Technology, DOI 10.1007/978-3-642-20158-5 10,
© Springer-Verlag Berlin Heidelberg 2011
264V.P. Koshelets et al.
this report, an overview of the SIR and FFO developments and optimizations is
presented. Improving on the fully Nb-based SIR we have developed and studied
Nb–AlN–NbN circuits, which exhibit an extended operation frequency range. Con-
tinuous tuningof the phase locked frequencyhas been experimentallydemonstrated
at any frequency in the range 350–750GHz. The FFO free-running linewidth has
been measured between 1 and 5MHz, which allows to phase lock up to 97% of
the emitted FFO power. The output power of the FFO is sufficient to pump the
matched SIS mixer. Therefore, it is concluded that the Nb–AlN–NbN FFOs are
mature enough for practical applications.
These achievements enabled the development of a 480–650GHz integrated
receiverforthe atmospheric-researchinstrumentTErahertzand submillimeterLImb
Sounder (TELIS). This balloon-borne instrument is a three-channel supercon-
ducting heterodyne spectrometer for the detection of spectral emission lines of
stratospheric trace gases that have their rotational transitions at THz frequencies.
One of the channels is based on the SIR technology. We demonstrate for the
first time the capabilities of the SIR technology for heterodyne spectroscopy in
general, and atmospheric limb sounding in particular. We also show that the
application of SIR technology is not limited to laboratory environments, but that
it is well suited for remote operation under harsh environmental conditions. Light
weight and low power consumption combined with broadbandoperation and nearly
quantum limited sensitivity make the SIR a perfect candidate for future airborne
and space-borne missions. The noise temperature of the SIR was measured to be
as low as 120K in double sideband operation, with an intermediate frequency
band of 4–8GHz. The spectral resolution is well below 1MHz, confirmed by
our measurements. Remote control of the SIR under flight conditions has been
demonstrated in a successful balloon flight in Kiruna, Sweden.
Capability of the SIR for high-resolution spectroscopy has been successfully
proven also in a laboratory environment by gas cell measurements. The possibility
to use SIR devices for the medical analysis of exhaled air will be discussed. Many
medically relevant gases have spectral lines in the sub-terahertz range and can be
detectedbyanSIR-basedspectrometer.TheSIR canbeconsideredas anoperational
device, ready for many applications.
A Superconducting Integrated Receiver (SIR) [1, 2] was proposed more than
10 years ago and has since then been developed up to the point of practical
applications [3–5]. Our approach consists in developing a single chip heterodyne
receiver, which is smaller and less complex than traditional devices. Typically, such
a receiver consists of a number of main components (local oscillator (LO), mixer,
antenna structure, phase lock circuit, etc.), which are usually built as separate units
and are complex (and thus costly). According to our concept (see Fig.10.1), we
have integrated all these components onto one single chip reducing overall system
complexityin changefor increasedon-chipandlithographicfabricationcomplexity.
10Integrated Submm Wave Receiver: Development and Applications 265
Fig. 10.1 Block-diagram of the superconducting integrated receiver
An SIR comprises on one chip all key elements needed for heterodyne detection: a
low-noise superconductor-insulator-superconductor (SIS) mixer with quasi-optical
antenna,a flux-flow oscillator (FFO)  actingas an LO and a secondSIS harmonic
mixer (HM) for the FFO phase locking.The concept of the SIR is very attractive for
many practical applications because of the compactness and the wide tuning range
is limited by the tuning range of the LO, typically 10%–15% for a solid-state
structure and the matching circuitry between the SIS and the FFO. A bandwidth
up to 30%–40% may be achieved with a twin-junction SIS mixer design. Another
potential advantage is the use of arrays of SIR channels within a single cryostat that
could operate at the same or different LO frequencies.
One of the important practical application of the SIR is TErahertz and sub-
millimeter LImb Sounder (TELIS) [5, 9, 10] – a three-channel balloon-borne
heterodyne spectrometer for atmospheric research developed in a collaboration
of four institutes: Deutsches Zentrum f¨ ur Luft- und Raumfahrt (DLR), Germany,
Rutherford Appleton Laboratories (RAL), United Kingdom, and SRON – Nether-
lands Institute for Space Research, the Netherlands (in tight collaboration with
Kotel’nikov Institute of Radio Engineering and Electronics, IREE, Moscow). All
three receivers utilize state-of-the-art superconducting heterodyne technology and
operate at 500GHz (by RAL), at 480–650GHz (by SRON C IREE), and at 1.8THz
(by DLR). TELIS is a compact, lightweight instrument capable of providing broad
spectral coverage, high spectral resolution and long flight duration. The TELIS
instrument serves also as a test bed for many novel cryogenic technologies and as a
pathfinder for satellite-based instrumentation.
TELIS is mounted on the same balloon platform as the Fourier transform
spectrometer MIPAS-B , developed by IMK (Institute of Meteorology and
266 V.P. Koshelets et al.
Climate research of the University of Karlsruhe, Germany) and is operated in the
mid-infrared.680–2;400cm?1/. Both instruments observesimultaneously the same
air mass, and together they yield an extensive set of stratospheric constituents that
can be used for detailed analysis of atmospheric chemical models, such as ozone
vertical profiles of ClO, BrO, O3and its rare isotopologues, O2, HCl, HOCl, H2O
and three rare isotopologues, HO2; NO; N2O; NO2; HNO3; CH3Cl, and HCN. In
this paper, the design and technology for the 480–650GHz channel as used in
the flight configuration are presented in conjunction with test results and the first
preliminary scientific results.
10.2 Flux Flow Oscillators
A Josephson Flux Flow Oscillator (FFO)  has proven [4, 5, 7] to be the most
developed superconducting LO for integration with an SIS mixer in a single-chip
submm-waveSIR [1–5]. TheFFO is a long Josephsontunneljunctionof the overlap
geometry (see Fig.10.2) in which an applied dc magnetic field and a dc bias
current, IB, drive a unidirectional flow of fluxons, each containing one magnetic
flux quantum, ˚0D h=2e ? 2 ? 10?15Wb. Symbol h is Planck’s constant and e is
the dc magnetic field applied to the FFO. According to the Josephson relation, the
junction oscillates with a frequency f D (1=˚0)V (about 483.6GHz/mV) if it is
biased at voltage V . The fluxons repel each other and form a chain that moves
along the junction. The velocity and density of the fluxon chain and thus the power
and frequency of the submm-wave signal emitted from the exit end of the junction
due to the collision with the boundary may be adjusted independently by proper
settings of IBand ICL. The FFO differs from the other members of the Josephson
oscillator family by the need for these two control currents, which in turn provides
the possibility of independent frequency and power tuning.
We experimentally investigated a large number of the FFO designs. The length,
L, and the width, W , of the FFO used in our study are 300–400?m and 4–28?m,
Fig. 10.2 Schematic view of a flux-flow oscillator
10Integrated Submm Wave Receiver: Development and Applications 267
respectively.Thevalueofthecritical currentdensity,JC, is in therange4–8kA=cm2
giving a Josephson penetration depth, ?J ? 6–4?m. The corresponding value of
the specific resistance is Rn ? L ? W is
calculations, we use a typical value of the London penetration depth, ?L? 90nm
for all-Nb junctions, and a junction specific capacitance, Cs ? 0:08pF=?m2. The
activeareaofthe FFO (i.e.the AlOxorthe AlN tunnelbarrier)is usuallyformedas a
long window in the relatively thick .200–250nm/ SiO2insulation layer sandwiched
between the two superconducting films (base and wiring electrodes). The so-called
“idle” region consists of the thick SiO2layer adjacent to the junction (on both sides
of the tunnel region) between the overlapping electrodes. It forms a transmission
line parallel to the FFO (not shown in Fig.10.2). The width of the idle region .WID
2–14?m/ is comparable to the junction width. The idle region must be taken into
account when designing an FFO with the desired properties. In our design, it is
practical to use the flat bottom electrode of the FFO as a control line in which the
current ICLproduces the magnetic field, which mainly is applied perpendicular to
the long side of the junction.
There are a number of important requirements on the FFO properties to make it
suitable for application in the phase locked SIR. Obviously, the FFO should emit
enough power to pump an SIS mixer, taking into account a specially designed
mismatch of about 5–7dB between the FFO and the SIS mixer, introduced to avoid
leakage of the input signal to the LO path. It is a challenge to realize the ultimate
performance of the separate superconducting elements after their integration in
a single-chip device. Implementation of the improved matching circuits and the
submicron junctions for both the SIS and the HM allows delivering optimal FFO
power for their operation.
Even for ultra wideband room-temperaturePLL systems the effective regulation
bandwidth is limited by the length of the cables in the loop (about 10MHz for
typical loop length of two meters). It means that the free-running FFO linewidth
(LW) has to be well below 10MHz to ensure stable FFO phase locking with a
reasonably good spectral ratio (SR) – the ratio between the carrier and total power
emitted by the FFO . For example, only about 50% of the FFO power can be
phase locked by the present PLL system at a free-running FFO LW of 5MHz. A
lowspectralratioresultsina considerableerrorat resolvingthecomplicatedspectral
line shape . Thus, a sufficiently small free-runningFFO LW is vitally important
for the realization of the phase locked SIR for the TELIS.
? 50–25 ? ?m2. For the numerical
Earlier the Nb–AlOx–Nb or Nb–AlN–Nb trilayers were successfully used for the
FFO fabrication.Traditionalall-Nb circuits are being constantlyoptimizedbut there
seems to be a limit for LW optimizations at certain boundary frequencies due
to Josephson self-coupling (JSC) effect  as well as a high frequency limit,
imposed by Nb gap frequency .?700GHz/. That is the reason for novel types
268 V.P. Koshelets et al.
Fig. 10.3 The dependencies of Rj/Rn ratio on critical current density Jc for SIS junctions of
different types fabricated at IREE
of junctions based on materials other than Nb to be developed. We reported on
development of the high quality Nb–AlN–NbN junction production technology
. The implementation of an AlN tunnel barrier in combination with an NbN
top superconducting electrode provides a significant improvement in SIS junction
quality. The gap voltage of the junction VgD 3:7mV. From this value, and the gap
voltage of the Nb film ?Nb=e D 1:4mV, we have estimated the gap voltage of our
NbN film as ?NbN=e D 2:3mV .
The dependency of the ratio of subgap to normal state resistance (Rj/Rn) vs.
critical current density .Jc/ for different types of the Nb-based junctions fabricated
at IREE is presented in Fig.10.3. One can see that the Nb–AlN–NbN junctions are
of very good quality at high current densities, important for implementation in THz
mixers. The same technique was further used to produce complicated integrated
circuits comprising SIS and FFO in one chip.
The use of Nb for top “wiring” layer is preferable due to lower losses of Nb
compared to NbN below 720GHz; furthermore, the matching structures developed
for the all-Nb SIRs can be used directly for the fabrication of receivers with
Nb–AlN–NbN junctions. The general behavior of the new devices is similar to the
all-Nb ones; even the control currents, necessary to provide magnetic bias for FFO,
were nearly the same for the FFOs of similar design.
A family of the Nb–AlN–NbN FFO IVCs measured at different magnetic fields
produced by the integrated control line is presented in Fig.10.4 .L D 300?m;
W D 14?m; WID 10?m/. A single SIS junction with an inductive tuning circuit
is employed as a HM for the LW measurements. The tuning and matching circuits
were designed to provide“uniform” coupling in the frequencyrange 400–700GHz.
10Integrated Submm Wave Receiver: Development and Applications269
Fig. 10.4 IVCs of the Nb–AlN–NbN FFO measured at different magnetic fields produced by the
integrated control line. The color scale shows the level of the DC current rise at the HM induced by
the FFO. Red area marks the region of the FFO parameters where the induced by FFO HM current
exceeds 25% of the Ig. This level is well above the optimal value for an SIS-mixer operation
Measured values of the HM current induced by the FFO oscillations (HM pumping)
are shown in Fig.10.4 by the color scale. The HM pumpingfor each FFO bias point
was measured at constant HM bias voltage of 3mV (pumping is normalized on the
current jump at the gap voltage, Ig D 140?A). From Fig.10.4, one can see that
an FFO can provide large enough power over the wide frequency range: limited
at higher frequencies only by the Nb superconducting gap in transmission line
electrodes (base and wiring layers) and below 400GHz by design of the matching
The Nb–AlN–NbN FFOs behave very similar to all-Nb ones. The feature at
about 600GHz where the curves get denser is a Josephson Self-Coupling (JSC)
boundary voltage. It was first observed for all-Nb FFOs . The JSC effect is
the absorption of the FFO-emitted radiation by the quasi-particles in the cavity
of the long junction. It considerably modifies the FFO properties at the voltages
V ? VJSC D 1=3Vg(VJSCcorresponds to 620GHz for the Nb–AlN–NbN FFO).
Just abovethis voltage,thedifferentialresistanceincreasesconsiderably;thatresults
in an FFO-LW broadening just above this point. This, in turn, makes it difficult
or impossible to phase lock the FFO in that region. For a Nb–AlOx–Nb FFO, the
transition corresponding to VJSCD Vg=3 occurs around 450GHz. So, by using the
Nb–AlN–NbN FFOs we can coverthe frequencygap from 450to 550GHz imposed
by the gap value of all-Nb junctions. The feature in Fig.10.4 around 1mV is very
likely dueto a singularityat the differenceof the superconductinggaps?NbN??Nb.
270V.P. Koshelets et al.
Continuous frequency tuning at frequencies below 600GHz for the Nb?AlN?
NbN FFOs of moderate length is possible, although the damping is not sufficient
to completely suppress the Fiske resonant structure at frequencies below Vg=3.
For short junctions with a small ’ (wave attenuation factor), the distance between
the steps in this resonant regime can be as large, that it is only possible to tune
the FFO at the certain set of frequencies. For a 300–400?m long Nb–AlN–NbN
junction, this is not the case – the quality factor of the resonator formed by a
long Nb–AlN–NbN Josephson junction is not so high at frequencies >350GHz.
Therefore, the resonance steps are slanting and the distance between them is not so
big (see Fig.10.4). This allows us to set any voltage (and any frequency) below
VJSC, but for each voltage only a certain set of currents should be used. So, in
this case we have the regions of forbidden bias-current values, specific for each
voltage below VJSC, instead of the forbidden voltage regions for the Fiske regime
in Nb–AlOx–Nb FFO . Special algorithms have been developed for automatic
working point selection in flight.
InFig.10.5,the typicalcurrent-voltagecharacteristics(IVCs)ofa Nb–AlN–NbN
SIS junctionof an area of about 1?m2is given, both the unpumpedIVC (solid line)
and the IVC when pumped by a Nb–AlN–NbN FFO at different frequencies(dotted
lines). One can see that the FFO provides more than enough power for the mixer
pumping.In this experiment,we use the test circuits with low-loss matchingcircuits
tuned between 400 and 700GHz. Even with the specially introduced 5dB FFO/SIS
mismatch (required for the SIR operation) the FFO delivers enough power for the
SIS mixer operation in the TELIS frequency range of 480–650GHz .
Fig. 10.5 The IVCs of the SIS mixer: unpumped – solid curve, pumped at different frequencies –
dashed and dotted lines
10 Integrated Submm Wave Receiver: Development and Applications 271
Fig. 10.6 The IVCs of the SIS mixer: unpumped – black solid curve, pumped at different FFO
bias currents (different powers) – lines with symbols; FFO frequency D 500GHz
Fig. 10.7 The pump current of the SIS mixer biased at 3mV as a function of the FFO bias current
at the fixed frequency 500GHz (see Figs.10.4 and 10.6)
An important issue for the SIR operation is a possibility to tune the FFO power,
while keeping the FFO frequencyconstant. This is demonstratedin Fig.10.6, where
the IVCs of an SIS mixer are shown, while being pumped at different FFO bias
currents (different powers). The dependence of the SIS pump current on the FFO
bias current is presented in Fig.10.7, showing that the FFO power can be tuned
more than 15dB, while keeping the same frequency by proper adjustment of the
control line current.
272V.P. Koshelets et al.
10.2.2Spectral Properties of the FFO
The FFO LW has been measured in a wide frequency range from 300GHz
up to 750GHz using a well-developed experimental technique . A specially
designed integrated circuit incorporates the FFO junction, the SIS HM and the
microwave matching circuits. Generally, both junctions are fabricated from the
same Nb/AlN/NbN or Nb/AlOx/Nb trilayer. The FFO signal is fed to the SIS
HM together with a 17–20GHz reference signal from a stable synthesizer. The
required power level depends on the parameters of the HM; it is about of 1?W
for a typical junction area of 1?m2. The intermediate frequency (IF) mixer product
.fIFD ˙(fFFO?n?fSYN)at? 400MHzisfirst boostedbyacooledHEMTamplifier
.Tn? 5K; gain D 30dB/ and then by a high-gain room-temperatureamplifier.
To accurately measure the FFO line shape, the IF signal must be time-averaged
by the spectrum analyzer. To remove low-frequency drift and interference from
the bias supplies, temperature drift, etc., we use a narrow bandwidth .<10kHz/
Frequency Discriminator (FD) system with relatively low loop gain for frequency
locking of the FFO. With the FD narrow-band feedback system that stabilizes the
mean frequency of the FFO (but does not affect FFO line shape), we can accurately
measure the free-runningFFO LW, which is determined by the much faster internal
(“natural”) fluctuations (see Fig.10.8).
Fig. 10.8 Spectra of the Nb–AlN–NbN FFO operating at 515.2605GHz (blue dashed line –
frequency locked by FD; red solid line – phase-locked). Linewidth D 1:7MHz; spectral ratio D
10Integrated Submm Wave Receiver: Development and Applications 273
The resulting IF signal is supplied also to the Phase Locking Loop (PLL)
system. The phase-difference signal of the PLL is fed to the FFO control line
current. Wideband operation of the PLL (10–15MHz full width) is obtained by
minimizingthe cableloop length.A part of the IF signal is deliveredto the spectrum
analyzer through a power splitter (see Fig.10.8). All instruments are synchronized
to harmonics of a common 10MHz reference oscillator.
The integrated HM may operate in two different regimes, either as a quasi-
particle mixer (SIS) or as a Josephson mixer. To exclude the noise from the
Josephson super-current fluctuations and thereby realize a pure quasi-particle
regime, the super current has to be suppressed by a relatively large magnetic field.
This requires a special control line placed near the SIS mixer. The quasi-particle
regime of the HM operation can also be realized with sufficient synthesizer power.
It has been shown  that the FFO LW and signal-to-noise ratio are almost the
same for these two regimes, although the phase noise might be somewhat lower in
the quasi-particle mode.
10.2.2.2Dependence of the FFO Linewidth on FFO’ Parameters
Detailed measurements of the FFO LW [18, 19] demonstrate a Lorentzian shape
of the free-running FFO line in a wide frequency range up to 750GHz, both at
higher voltages on the flux flow step (FFS) and at lower voltages in the resonant
regime on the Fiske steps (FSs). This implies that the free-running (“natural”) FFO
LW in all operational regimes is determined by the wideband thermal fluctuations
and the shot noise. This is different from many traditional microwave oscillators,
where the “natural” LW is very small and the observed LW can be attributed mainly
to external fluctuations. It was found [18,19] that the free-running FFO LW, ıf ,
exceeds theoretical estimations made for lumped tunnel Josephson junction. The
expression for the LW dependency on voltage and differential resistances found for
all-Nb FFOs [18,20] is valid for Nb–AlN–NbN junctions as well:
dC K ? RCL
where Si0 is the power density of low frequencycurrent fluctuations, RdBand RdCL
are differential resistances on bias and control line currents, respectively. Note that
ratio RdCL=RdBis constant for fixed FFO bias, so ıf D A(IB) (RdB)2Si0.
Earlier, a so-called Super Fine Resonance Structure (SFRS)  was observed
on the FFO IVCs, resulting in the jumps of the FFO between tiny steps (frequency
spacing is of about 10MHz, see Fig.10.9). The presence of the SFRS prohibits
phase locking at frequencies between the steps. This is unacceptable for practical
applications. Recently, we found that the SFRS is related to interference of the
acoustic waves created by the FFO (generation of the phonons by Josephson
junction, see ). A special technological procedure allows us to eliminate this
interference and to realize continuous FFO-frequency tuning in the SIR, being
274V.P. Koshelets et al.
Fig. 10.9 Down-converted spectra of the FFO: (a) free-running FFO; (b), (c) – the lines show the
maximum FFO signal level recorded in the MaxHold regime of the Spectrum Analyzer (the top
point of curve “a”) on the FFO frequency, measured before (b) and after (c) special Si substrate
Fig. 10.10 Linewidth dependency on frequency for two types of the FFO
vitally important for TELIS project (see Fig.10.9). Details of this study will be
In Fig.10.10, we present a comparative graph of the free-running FFO LW for
two types of the tri-layer. One can see that the LW of Nb–AlN–NbN FFO is twice
as small up to 600GHz. It should be emphasized that due to overlapping FSs
continuous tuning is possible and any desirable frequency can be realized. Several
10Integrated Submm Wave Receiver: Development and Applications 275
“stacked” stars at certain frequenciesfor the NbN FFO mean that the best LW value
can be selected by adjusting FFO bias. Note that the spread in the LW values at
a selected frequency is small and all can actually be applied for measurements.
Each star corresponds to an “allowed” bias current at an FS (as described above in
Sect.2.1). Althoughthe FFO tuning on an FS is complicated,the benefit in LW (and
consequently the spectral ratio) is worth the effort. Linewidths below 3MHz can
be achieved in the whole range between 350 and 610GHz. An abrupt increase of
the FFO LW at some frequencies is caused by the Josephson self-coupling effect.
The JSC (absorption of the FFO-emitted radiation by the quasi-particles in the
cavity of the long junction, see above) considerably modifies the FFO properties
at the voltages V ? VJSC D 1=3Vg (VJSCcorresponds to 620GHz for the
Previous LW measurements have demonstrated[7,23] the essential dependences
of the free-running FFO LW on the FFO voltage, its current density and geometry
of the biasing electrodes. In this report, we summarize the results of the FFO study
and optimization of the FFO layout for both types of FFOs. Recently, it was shown
[4, 7] that the LW decreases considerably with increasing width, W , of the FFO
junction. This is valid for all frequencies of interest, and consequently, the spectral
ratio of the phase locked FFO for wide junctions is better. We have increased the
FFO width up to 28?m, which is more than five times the Josephson penetration
depth ?J. A number of FFOs with the same electrode layout, but different widths
of the FFO junction (W D 4, 8, 12, 16, 20 and 28?m) are fabricated using the
same technological procedure yielding the same junction parameters (normal state
resistance ? area, RnS D 30??m2). The results of the LW measurements of these
circuits at three frequencies are presented in Fig.10.11.
Fig. 10.11 Linewidth of free-running FFOs (left axis) and corresponding spectral ratio for the
phase-locked FFO (right axis) measured at different FFO frequencies as a function of FFO width.
All circuits are fabricated by the same technological procedure .RnS D 30??m2/
276V.P. Koshelets et al.
Even for the largest tested width .W D 28?m/, there is no evidence of deteri-
oration in the FFO behaviour. Furthermore, the power delivered to the SIS mixer
is getting higher and the LW lower at all frequencies. The decrease of the FFO LW
with increasingFFO width is in accordancewith existingtheoreticalmodels andour
expectations. The bias current differential resistance, Rd, decreases approximately
inversely proportional to the bias current IB. Since the FFO LW is proportional
to Rd2?IB, it scales down linearly with the junction width. Of course, one can
expect that the LW decrease will saturate and the FFO performance will deteriorate
with further increase of the width (e.g., due to appearance of transversal modes).
Without a reliable theory, the optimal value of the FFO width has to be determined
experimentally. Note that for a wider FFO the center line of the junction is shifted
away from the edge of the control line (the RdCLgoes down). This may result
in a considerable reduction of extraneous noise from external magnetic fields.
Furthermore, a wider FFO presumably will have a more uniform bias current
distribution . At the present state, the width of the FFO for TELIS is chosen
to be 16?m. This is a tradeoff between LW requirements and technical limitation
on the maximum bias and control line currents (both should not exceed 70mA).
In contrastto variationofthe FFO LW on theFFO width,previousmeasurements
 have demonstrated a considerable increase of the FFO LW with the FFO current
density. This contradicts the simplified consideration: the increase of the FFO
current density (as it is for increase of the FFO width) should result in the increase
of the total FFO bias current, IB, and reduce the FFO differential resistance on the
bias current Rd. Since the FFO LW is proportionalto Rd2?Ib, one should expect the
decrease of the measured FFO free-running LW for larger FFO current density. In
reality, Rddoes not decrease as much as this simple consideration predicts and the
LW increases. On the contrary, a high value of the current density .Jc? 8kA=cm2/
is important for wide-band operation of the SIS-mixer at the submm wave range.
The increase of the FFO LW with current density (as discussed above) creates a
serious problemin the designand developmentof SIR chips. Implementationof two
separate tri-layers with different current densities – one for the SIS mixer (high Jc)
and the other one for the FFO/HM (lower Jc) seems to be a solution. We have
successfully tested and verified this approach for the SIR microcircuits for TELIS.
Improvement of the FFO performance was obtained by enlarging the electrodes
overlapping area, the so-called “idle region”. Larger overlapping presumably
provides a more uniform bias-current distribution, due to reduced inductance of the
overlapping electrodes. Larger overlapping of the FFO electrodes also implies that
theFFO ofthe same widthis shiftedfromthe edgeof thebottomelectrode,resulting
in a considerable decrease of the RdCLvalue. Note that for a wide FFO also some
shift of the FFO center line appears due to increasing of the width. Experimentally,
we found that an idle region WID 10?m is the optimal value for the present FFO
design. Up to now, there is no adequate model that can quantitatively describe
both the processes in the FFO and a self-consistent distribution of the bias current.
Nevertheless, the presented results are very encouraging and these modifications of
the FFO were implemented in the TELIS SIRs.
10Integrated Submm Wave Receiver: Development and Applications 277
To further explore this approach, we have developed different designs of the
“self-shielded” FFO with a large ground plane in the base electrode. Such FFOs
are expected to be less sensitive for variations in the external magnetic field and
have to provide more uniform bias current distribution (since all bias leads are
laying over superconducting shield and have low inductance). Actually, the low-
inductive bias leads provide a possibility of optimal (rather than uniform) current
distribution, “regulated” by the FFO itself. The last feature optimizes the emitted
FFO power. Indeed, the IVCs of all shielded FFOs are much more reproducible;the
power delivered to HM is higher compared to a traditional design. Unfortunately,
the free-running LW for all variants of shielded FFOs with separate bias leads is
much larger than for FFOs of traditional design. It seems that injection of the bias
via separate leads results in some spatial modulation of bias current  despite
the additional triangular elements added for more uniform current injection. On
the contrary, designs that employed three superconducting electrodes provide both
perfect pumping and improved LW, details will be published elsewhere.
10.2.2.3 Spectral Ratio, Phase Noise
As it was mentioned above, the free-runningFFO LW has to be well below 10MHz
to ensure stable FFO phase locking with a reasonably good spectral ratio (SR, the
ratio between the carrier and total FFO power). For example, only about 50% of
the FFO power can be phase locked by the present TELIS PLL system at free-
running FFO LW of 5MHz. A low spectral ratio results in a considerable error
at resolving of the complicated atmospheric line shapes . For the given PLL
system, the value of the SR is fully determined by the free-running FFO LW: these
two quantities are unambiguously related (see Fig.10.12, where data for FFOs of
different designs and types are presented). The theoretical curve, calculated in ,
coincides reasonably well with the experimental data. A possibility to considerably
increase the SR by application of the ultra-wideband cryogenic PLL system has
been recently demonstrated .
An important issue for TELIS operations is the possibility to tune the FFO
frequency and power independently, while providing the same spectral ratio of PL
FFO. The TELIS HM is pumped by a tunable reference frequency in the range
of 19–21GHz from the LO Source Unit (LSU), phase locked to the internal ultra
stable 10MHz Master Oscillator. The HM mixes the FFO signal with the n-th
harmonic of the 19–21GHz reference. The LW and SR of the TELIS FFO are
almost constant over a wide range of FFO bias current at fixed FFO frequency (see
Fig.10.13). From this figure, one can see that the SR is about 50% over the range
of bias current, Ib, 14–30mA, while the pumping level varies from 3:5?A at IbD
14mA up to 81?A at Ib D 30mA. Furthermore, the SR D 34% can be realized
at Ib D 12mA, where the HM pumping is below 0:5?A. It means that at proper
choice of the HM voltage and LSU power even moderate HM pumping by the FFO
is enough for efficient PLL operation (providing sufficient signal-to-noise ratio).
278V.P. Koshelets et al.
Fig. 10.12 Spectral ratio for the phase-locked FFO of different types and designs as a function
of free-running FFO linewidth. Solid line – calculated dependence of the SR on FFO LW for PLL
bandwidth D 10MHz
Fig. 10.13 Dependence of the HM current induced by FFO (HM pumping) and spectral ratio after
FFO phase-locking as a function of FFO bias current. All the data measured at FFO frequency of
To prove the capabilities for high-resolution spectroscopy, line profiles around
625GHz of OCS gas have been successfully measured by the SIR operating in the
DSB regime . The tests were done in a laboratory gas cell setup at a gas pressure
down to 0.2mBar, correspondingto the FWHM LW < 5MHz. It was demonstrated
10Integrated Submm Wave Receiver: Development and Applications 279
that the spectrum recorded by the Digital Auto Correlator (DAC) is a convolution
product of the signal (gas emission lines) with the FFO line spectrum; resolution in
this experiment is limited by DAC back-end. More detailed spectral measurements
data will be presented in the next section.
To investigate the ultimate frequency resolution of the receiver, we have mea-
sured the signal of the synthesizer multiplied by a super-lattice structure . The
signal recorded in these measurements is a convolution of the narrow-bandwidth
(delta-function-like)spectrum of the synthesizer with phase locked spectrum of the
FFO with an accuracy of the used resolution bandwidth of the spectrum analyzer
(30kHz). It was confirmed that the frequency resolution of the receiver is better
The residual phase noise of the phase locked FFO – measured relative to the
reference synthesizer – as a function of the offset from the carrier is plotted in
Fig.10.14. To get the absolute FFO phase noise, one should add the synthesizer
noise multiplied by n2to the residual phase noise of the FFO. Data for the
Rohde&Schwarz R ?SMF100A Microwave Signal Generator with improved phase
noise  are also presented in Fig.10.14, for the case where the FFO, operating
at 450GHz, is locked to the 20th harmonic of the synthesizer, n2D 400. The total
(absolute)FFO phasenoise (solid line in Fig.10.14)is dominatedby the synthesizer
Fig. 10.14 Experimental phase noise of a phase locked FFO at 450GHz. Since the phase noise
of the FFO is measured relative to the 20th harmonic of the synthesizer, the synthesizer noise
, multiplied by a factor 202D 400, should be added to the residual FFO noise to get the total
(absolute) FFO phase noise – solid line
280V.P. Koshelets et al.
noiseforoffsets< 10kHz. Thenoiseat largerfrequencyoffsetis mainlydueto PLL
system. Note that the FFO phase noise is overestimated since no subtraction of the
noise added by the IF amplifier chain was performed;actually at offsets muchlarger
than the PLL regulation bandwidth .>20MHz/ the measured phase noise is mainly
determined by the IF chain.
possible for Nb–AlN–NbNFFOs due to bendingand overlappingof the FSs, so that
any desirable frequency can be realized. A possibility to phase lock the Nb–AlN–
NbN FFO at any frequency in the range 350–750GHz has been experimentally
demonstrated. An optimized design of the FFO for TELIS has been developed
and tested. A free-running LW value from 5 to 1MHz has been measured in the
frequency range 300–750GHz for a “wide” FFO. As a result, the spectral ratio of
the phased locked FFO varies from 50% to 97% correspondingly. The “unlocked”
rest of the total FFO power increases the phase noise and the calibration error. To
ensureremoteoperationofthephaselockedSIR severalproceduresforits automatic
computercontrolhave been developedand tested. New designs of the FFO intended
for further improvement of its parameters are under development, but even at the
present state the Nb–AlN–NbN FFOs are mature enough for practical applications.
10.3.1 TELIS Instrument Design
The front-end of the ballon-borne TELIS instrument for atmospheric research is
common for the three channels on board. It consists of the pointing telescope, a
calibration blackbody, relay and band-separating optics (see Fig.10.15). Details of
Fig. 10.15 Optical lay-out of the TELIS SIR channel
10Integrated Submm Wave Receiver: Development and Applications 281
the optical design can be found in [28–30]. The three mirrors of the dual offset
Cassegrain telescope are mounted on a common frame, rotatable around the optical
axis of the outputbeam.Limb scanningis performedbetweenthe uppertroposphere
(8–10km in the Arctic) to flight altitude (typically 32km) in 1–2km steps. At
the tangent point of the line of sight, the vertical (elevation) resolution is about
2km for an observational frequency of 500GHz, scaling inversely proportional
with frequency. In horizontal (azimuth) direction, the spatial resolution is about a
factor of 2 less due to the anamorphicity of the telescope. This is allowed as the
atmospheric properties within the beam hardly depend on the azimuth.
The radiometric gain of the spectrometers is calibrated once or twice in
every Limb scan using a conical blackbody reference source and a measurement
of the cold sky. For this, a small flip mirror is included between the telescope and
the beam-separatingoptics.By measuringat two up-lookingtelescopepositions,the
impact of the remaining air above the gondola can be assessed.
Simultaneous observation by the receivers is achieved by quasi-optical beam
splitting. First, a wire-grid-basedpolarizing beam splitter is employed to reflect one
linear polarization to the 500GHz channel, the other linear polarization is split by
a dichroic filter between the SIR channel and the THz channel. Subsequently, off-
set mirrors shape and direct the three beams to the cryogenic channels. Inside the
custom designed liquid-helium cooled cryostat, each receiver has dedicated cold
optics, a superconducting mixing element and IF amplifiers.
The very compact 500GHz receiver channel consists of a fixed-tunedwaveguide
SIS mixer, a cryogenic solid-state LO chain and a low-noise IF chain operating
at a relatively high IF .IF D 15–19GHz/ . The 1.8THz channel employs a
type optical interferometer. The mixer is based on a phonon-cooledNbN HEB (Hot
Electron Bolometer) . The 480–650GHz SIR receiver channel is based on a
single-chip SIR, as described in the next section.
The warm optics couples to the SIR channel with a beam that has a waist radius
rangingfrom2 to 3mm, locatedat the cryostat window.The system-pupilis imaged
by two additional mirrors on the silicon elliptical lens; on the back surface of
this lens, the SIR chip is located. The SIR-channel cold-optics is also frequency
independent to fully exploit the wide-band operation of the SIR device.
The amplitude-phase distribution of the near field beam of the SIR cold channel
at 600GHz as measured at the dewar window is shown in Fig.10.16. The beam
waist is measured to be 2.25mm, which is within 1% of the designed value. The
measured Gaussisity of the beam is 92.4%.
The IF processor (located on the main frame of TELIS) converts the amplified
IF output signals of the three receivers to the input frequency range of the digital
autocorrelator. The digital autocorrelator has a bandwidth of 2 ? 2GHz with 2,048
spectral channels. Both the IF processor and the digital autocorrelatorare developed
by Omnisys Instruments AB .
The SIR channel is controlled with a battery-operated ultra low-noise biasing
system. Since noise on the bias lines of the FFO translates in a wider FFO
LW, several precautions, such as decoupling of digital control lines and extensive
282 V.P. Koshelets et al.
Fig. 10.16 The amplitude (top figure) and phase (lower figure) distribution of the near field beam
of the SIR channel. The amplitude is given in units of dB. The distance from the beam waist is
110mm and the frequency is 600GHz
filtering and shielding, are implemented.The SIR bias unit is digitally controlled by
digital autocorrelator, and with the host instrument MIPAS. A radio link provides
real-time two-way contact with the ground segment consisting of a server computer
with three dedicated client computers, coupled through TCP/IP socket connections.
The complete system is dimensioned to have sufficient cooling liquids and battery
power for a 24h flight.
10Integrated Submm Wave Receiver: Development and Applications 283
Fig. 10.17 Photo of the SIR microcircuit with double-slot antenna
10.3.2 SIR Channel Design
A key element of the 480–650GHz channel is the SIR [1–5] that comprises in
one chip (size of 4 ? 4 ? 0:5mm, see Fig.10.17) a low-noise SIS mixer with
quasioptical antenna, a superconducting FFO  acting as an LO and a second SIS
HM for FFO phase locking. Since the free-running LW of the FFO can be up to
10MHz, for spectral applications the FFO has to be locked to an external reference
oscillator employing a phase lock loop(PLL) system. The concept of the SIR looks
very attractive for TELIS due to a wide tuning range of the FFO. In the SIR, the
bandwidth is basically determined by the SIS mixer tuning structure and matching
circuitry between the SIS and FFO; bandwidth up to 30–40% may be achieved with
a twin-junctionSIS mixer design (bothfor double-slotand double-dipoleantennas).
To achievethe requiredinstantaneousbandwidthof 480–650GHz, a twin-SISmixer
with 0:8?m2junctions and new design of the FFO/SIS matching circuitry were
implemented. A microscope photograph of the central part of the SIR chip with
double-dipole antenna is presented in Fig.10.18.
The resolution of the TELIS back-end spectrometer is 2.160MHz, sufficient
to resolve the exact shape of atmospheric lines. The FFO line shape and spectral
stability should ideally be much better than this. However, the free-running LW of
the FFO can be up to 10MHz and thereforea PLL has been developedto phase lock
the FFO to an external reference oscillator [6,14]. For this, a small fraction of the
FFO power is first directed to a so-called HM, placed on the SIR chip. The HM is
pumped by an off-chip LSU, which is a tunable reference frequency in the range of
19–21GHz. The frequency of the LSU is chosen such that the difference frequency
284 V.P. Koshelets et al.
Fig. 10.18 Central part of the SIR chip withdouble-dipole antenna, twinSIS-mixer, and harmonic
mixer for FFO phase-locking
of the nth harmonic of the LSU, generated by the HM, and the FFO is about 4GHz.
This difference signal is then amplified by a cryogenic low-noise HEMT amplifier
and down-converted to 400MHz by using a second reference at 3.6GHz. Finally,
the frequency and phase of this 400MHz signal is compared against yet another
reference frequency of 400MHz and the resulting error signal is fed back to the
FFO. The LSU and the reference signals at 3.6GHz and at 400MHz are all phase
locked to an internal ultra stable 10MHz Master Oscillator.
All components of the SIR microcircuits are fabricated in a high quality
Nb–AlN/NbN tri-layer on a Si substrate . The receiver chip is placed on the
flat back surface of the elliptical silicon lens (forming an integrated lens-antenna)
with accuracy 10?m, determined by the tolerance analysis of the optical system.
As the FFO is very sensitive for external electromagnetic interferences, the SIR
chip is shielded by two concentric cylinders: the outer cylinder is made of cryo-
perm and the inner one of copper with a 100?m coating of superconducting lead.
All SIR channel components (including input optical elements) are mounted on a
single plate inside a 240 ? 180 ? 80mm box cooled by the thermo-straps to the
temperature of about 4.2K.
10.3.3 TELIS-SIR Channel Performance
The TELIS-SIR channel has been characterized in eight micro-windows that have
been selected for the flight in (Sweden). These micro-windows have the following
• 495.04GHz for H218O
• 496.88GHz for HDO
• 505.60GHz for BrO
• 507.27GHz for ClO
• 515.25GHz for O2, pointing, and temperature
10 Integrated Submm Wave Receiver: Development and Applications285
Fig. 10.19 Measured DSB receiver noise temperature of the SIR device selected for flight at
8GHz IF (solid line) and integrated in the 4–8GHz IF range (dashed line)
• 519.25GHz for BrO and NO2
• 607.70GHz for ozone isotopes
• 619.10GHz for HCl, ClO and HOCl
Initial flight values for the parameters for the FFO, SIS, and HM mixers have
been determined for each micro-window. Dedicated algorithms allowing for fast
switching between LO frequencies and for in-flight optimization of the SIR have
been developed (see below). It takes about 1min of stabilization and optimization
to switch between two LO settings. All experimental results discussed here have
been obtained with the SIR flight device.
The measured double sideband (DSB) receiver noise temperature TR, uncor-
rected for any loss, is presented in Fig.10.19 as a function of LO frequency. As
can be seen, the noise is well below 200K at all frequencies of interest, with a
minimum of 120K at 500 and 600GHz. The noise peak around 540–575GHz
is partially spurious, caused by absorption of water vapor in the path between
calibration sources and the cryostat, and partially real – due to properties of the
SIS-mixer tuning circuitry. The relatively high noise in this band is of no concern
for science observations, since this part of the atmospheric spectrum is obscured
by a highly saturated water-vapor line rendering it virtually useless for atmospheric
science. The noise as a function of IF is fairly flat in the frequency range 4–8GHz,
as can be seen in Fig.10.20, where (DSB) receiver noise temperature is plotted as
a function of IF. The dependence of the receiver noise temperature on the SIS bias
voltage is shown in Fig.10.21; one can see that for Nb–AlN/NbN circuits there is
very wide range of SIS bias voltages where TRis almost constant.
286 V.P. Koshelets et al.
Fig. 10.20 DSB receiver noise temperature as a function of the IF, taken at two FFO frequencies:
497 and 601GHz
Fig. 10.21 DSB receiver noise temperature as a function of the SIS bias voltage measured at the
FFO frequency 497GHz
10Integrated Submm Wave Receiver: Development and Applications 287
Fig. 10.22 Spectra of the FFO operating at 515.2GHz (blue dashed line – frequency locked; red
solid line – phase-locked). Linewidth .LW/ D 1:5MHz; signal-to-noise ratio .SNR/ D 36dB;
spectral ration .SR/ D 93:5%. Spectra measured with RBW D 1MHz; span D 100MHz
After optimization of the FFO design, the free-running LW between 7 and
0.5MHz has been measured in the frequency range 350–750GHz (see Fig.10.10),
which allows to phase lock from 35% to 95% of the emitted FFO. Example of the
free-running (frequency-locked)and phase locked spectra of the FFO measured for
flight SIR at one of the frequencies selected for first TELIS flight are presented in
Data for five important TELIS frequencies are summarized in Table10.1. It
should be mentioned that the noise of the digital electronics at frequencies of about
1MHz slightly increases the measured LW value, while the PLL is able to suppress
the interference (that results in larger SR than can be expected from measured LW).
Note also the dependence of the SR and LW on the FFO bias current related to
variation of the differential resistance along FS.
For the TELIS measurement strategy, it is important to know whether the timing
of limb sounding should depend on the stability of the complete receiver chain.
The stability determines the optimum achievable measurement time for a single
integration, and thus the required frequency of the calibration cycle. The stability
of the complete TELIS-SIR system has been determined with a noise-fluctuation
bandwidth of 17MHz, and the results  are presented in Fig.10.23. For the two
IF channels that are used to determine the Allan variance, it is found that the Allan
stability time is about 13.5s. When the difference of the two channels is taken to
determine the Allan variance (this is the so-called spectroscopic, or differential,
288V.P. Koshelets et al.
Table 10.1 Data for the flight SIR at selected TELIS frequencies
FFO frequency (GHz)LW (MHz)
FFO Ib (mA)
Fig. 10.23 System stability of the SIR channel. FFO is phase locked at 600GHz. The two lines
at the top (red and green) represent individual channel variances, the blue line is representative of
the spectroscopic variance and the straight black line corresponds to the radiometer equation
mode), the Allan stability time of 20s is found. This is comparable to stabilities
measured for astronomical receivers.
Within TELIS, a 1.5s integration time per tangent height is used. This is mainly
driven by the required integrated signal levels at the autocorrelator input. The
stability of the SIR channel thereforeposes no constraints on the observingstrategy.
The SIR is a complicated device as it contains multiple interactive supercon-
ducting elements: an SIS mixer, an FFO, and an HM for the FFO phase locking.
Special algorithms and procedures have been developed and tested to facilitate
characterization of the SIR at reasonable timescales and for the SIR control during
the flight. These routines include:
• Fast definition of the FFO operational conditions (both on the FS and in the
• Measurements of the free-running FFO LW.
10Integrated Submm Wave Receiver: Development and Applications 289
• Optimization of the LSU and HM parameters.
• Optimization of the PLL operation.
• Minimization of the SIR noise temperature.
• Setting all predefined SIR parameters in the exact sequence for control during
• Continuous monitoring of the main SIR parameters.
• Adjustment (or recovering) of the SIR operational state.
10.3.4 Kiruna Campaigns and Preliminary Science Results
TELIS had two successful scientific campaigns from Kiruna, North-Sweden, in
March 2009 and in January 2010. The instrument was launched together with the
MIPAS instrument on the MIPAS-B2 gondola (see Fig.10.24). The launch of both
flights took place around midnight. During the ascents, the SIR channel behaved
nominally and already after 30min the first spectra were recorded. In the 2009
flight, the first flight ceiling of 35km was reached after 3h and 1h later the flight
continued at 28km altitude. In the 2010 flight, the ascent took more than 4h to
reach a flight ceiling of 34km where the balloon stayed for the remainder of the
for species with a diurnal cycle and for instrument calibration. The instrument
proved to be stable against the strong temperature variations of the atmosphere
duringascent (with ambienttemperaturesas lowas ?90ıC) andduringsunrise.The
south eastern wind allowed for long flights of about 12h over Finland during both
Fig. 10.24 TELIS-MIPAS launch at Esrange, Sweden; March 2009. Balloon size: 400;000m3;
payload weight: 1,200kg
290 V.P. Koshelets et al.
campaigns. After sunrise, the diurnal cycle of various species was monitored and in
total severalhundredlimb sequenceshavebeenrecordedin eachflight. TheMIPAS-
landing and recovery, the instruments were found to be undamaged, allowing for
post-flight checks and calibration measurements.
The science goals of the campaign from Kiruna, North Sweden, were threefold:
investigation of the stratospheric hydrological cycle by measurements of isotopic
water, catalytic ozone destruction by chlorine chemistry, and the bromine content
of the stratosphere. In addition, measurements were performed for space-borne
instruments (ENVISAT satellite and in 2010 also SMILES aboard the International
SpaceStation).Data presentedin Fig.10.25provethe capabilities ofthe TELIS-SIR
channel for high-resolution spectroscopy. In this case, the FFO frequency is tuned
to 505.600 GHz, the telescope is 6 degrees up-looking, and the gondola altitude is
35.780km.The widthof the ozonelines arealmost fullydeterminedby atmospheric
conditions (Doppler and pressure broadening) and are about 10MHz, as expected.
Chlorine ozone destruction peaks in the arctic winter and/or spring when the so-
called polar vortex breaks up. During this event, the ClO radical, responsible for
catalytic ozone destruction, becomes available in huge amounts. However, chlorine
is also stored in nonreactivereservoirspecies of which HCl is an importantmember.
The amount of HCl in the stratosphere is a measure of the total nonactiveCl content
and is as such an important species to monitor in ozone chemistry studies.
InFig.10.26measuredspectraareshowninwhichtheHCl lineat IF D 6:81GHz
is well pronounced for line of sights. A line of sight is determined by the flight
5.05.5 6.06.5 7.0
Fig. 10.25 Ozone spectrum recorded during the 2009 campaign with the SIR channel (FFO
frequency D 505:6GHz) from an altitudeof 35.780kmand withthetelescope pointing6ıupward.
The width of the lines is ca. 10MHz and is fully determined by atmospheric conditions. The
intensity of the received signal in Kelvin is plotted as ordinate of the graphs
10 Integrated Submm Wave Receiver: Development and Applications291
Fig. 10.26 HCl spectra
recorded during the 2009
campaign with the SIR
frequency D 619:1GHz)
from an altitude of ca. 28km.
The spectra correspond to
line of sights with tangent
heights in the range of 10.5
(top) to 25.5km (bottom),
which is also the altitude for
which a particular spectrum is
most sensitive. The intensity
of the received signal in
Kelvin is plotted as ordinate
of the graphs
altitude of the balloon platform and the tilt of the telescope. Each line of sight
results in different altitude sensitivities and effectively probes different parts of
the atmosphere. The altitudes mentioned in Fig.10.26 refer to the lowest probed
altitude, the tangent height, for a certain line of sight and generally corresponds to
the altitude for which the measurement is mostly sensitive.
Bromine depletes ozone even more aggressively than chlorine on a per molecule
basis, but its abundances are much lower. In fact, the total amount of stratospheric
bromine is still not settled and is currently one of the main uncertainties in the
importance of bromine in ozone depletion.
In 2009, the flight took place in nonvortex conditions whereas in 2010 the flight
was in the polar vortex. The diurnal cycle of ClO has been observed in both flights,
albeit with a much higher time resolution in the 2010 flight (ca. 1min). In 2009,
BrO has been detected, although barely as the line was superimposed on another
spectral feature. However, in 2010 the BrO line, with a level of only ca. 0.3K, was
isolated and clearly detected. The data reduction is on-going but the first spectra for
HCl, ClO, and BrO are presented in Figs.10.26, 10.27, and 10.28, respectively.
10.3.5SIR for Noninvasive Medical Diagnostics
High sensitivity and spectral resolution of the integrated spectrometer enables the
analysis of multicomponent gas mixtures. Exhaled air of human includes about
400 gases, of which some can be indicators of various diseases and pathology. For
example,nitric oxide,NO, was detected in the exhaled air of patients sufferingfrom
bronchial asthma, pneumonia, and other chronic inflammatory diseases of upper
airways. Besides, nitric oxide may have an effect on the reaction of tumors and
292 V.P. Koshelets et al.
Fig. 10.27 ClO spectra
recorded during the 2010
campaign with the SIR
frequency D 507:3GHz)
from an altitude of ca. 34km.
The two sets of spectra
correspond, respectively, to
25 and 19km tangent heights,
i.e. the altitudes for which
these spectra are most
sensitive. The increase of ClO
over time is clearly visible.
The intensity of the received
signal in Kelvin is plotted as
ordinate of the graphs
6.06.1 184.108.40.206 6.5
IF Frequency [GHz]
ClO diurnal cycle
Fig. 10.28 BrO spectra
recorded during the 2010
campaign with the SIR
frequency D 519:3GHz)
from an altitude of ca. 34km.
In the daytime measurement
(orange), the BrO line is
visible at 5.6GHz. During the
(black), BrO is absent, as
expected. The shown tangent
heights are from top to
bottom 26, 28, 30, and 32km,
i.e. the altitudes for which
these spectra are most
sensitive. The intensity of the
received signal in Kelvin is
plotted as ordinate of the
5.5 5.6 5.7 5.8 5.96.0
IF Frequency [GHz]
30 times averaged
10 Integrated Submm Wave Receiver: Development and Applications293
healthy tissues on radiation therapy. Another example may concern the opportunity
of noninvasive diagnostics of gastritis or peptic ulcer of the stomach by measuring
the concentration of ammonia in exhaled air. Nowadays urease respiratory tests
(application of urea with C13) are mainly used to detect the diseases. However, the
method is quite expensive and its sensitivity is restricted by natural variations of
C13in exhaled air during the procedure.Natural concentrations of ammonia instead
are quite low, so the measurement of the ammonia concentration could be a good
alternative. Another important application concerns the noninvasive diagnostics of
diabetes, where exhaled acetone is an indicator.
A laboratory setup for spectral analysis of the exhalted air has been developed at
range, spectral resolution below 1MHz), based on the integrated spectrometer for
atmospheremonitoring.The instrument parameters allow us to measure the spectral
lines of the rotational transitions for most of the substances in the exhalted air. The
laboratory setup has been developed and demonstrated using the gases OCS and
NH3in the laboratory gas cell. Clear and well-defined response has been measured
at the expected frequencies of spectral lines for pressures down to 10?3mBar.
Examples of the NH3spectra recorded by the SIR with the Fast Fourier Transform
of the additional oscillator are presented in Fig.10.29 and 10.30, respectively. The
possibility to measure the spectral response in a few seconds has been demonstrated
experimentally. This allows to carry on the real-time medical survey. First spectral
measurements by the integrated receiver of exhalted air in the sub-THz range have
demonstrated good selectivity and speed of the analysis as well as high sensitivity.
Fig. 10.29 NH3spectra measured by the SIR with FFTS back-end at different pressures
294 V.P. Koshelets et al.
Fig. 10.30 Response (derivative of the spectral line), measured for NH3gas by the integrated
receiver with implementation of novel technique (details will be published elsewhere)
For example, we have measured ammonia concentration with sensitivity on the
order of 10?9(1ppb).
Thecapabilityofthe SIR forhighresolutionatmosphericspectroscopyhas beensuc-
cessfully proven with scientific balloon flights from Kiruna, North Sweden. During
the two 12-h missions, phase locked SIR operation and frequency switching in the
480–650GHz frequency range has been realized. An intrinsic spectral resolution
of the SIR well below 1MHz has been confirmed by CW signal measurements
in the laboratory. An uncorrected DSB noise temperature below 120K has been
measured for the SIR when operated with a phase locked FFO at an IF bandwidth
of 4–8GHz. To ensure remote operation of the phase locked SIR several software
procedures for automatic control have been developed and tested. The first tentative
HCl profile has been presented and its quality looks promising for future data
reduction. Diurnal cycles of ClO and BrO have been observed at different viewing
configurations (altitude), with BrO line level of only about 0.5K. Possibilities to
use the SIR devices for analysis of the breathed out air at medical survey have been
demonstrated. The SIR can be considered as an operational device, ready for many
Acknowledgements The authors thank colleagues at DLR, IPM, IREE, and SRON for help
and assistance in the SIR channel design and characterization: J Barkhof, A Baryshev, J Kooi,
10Integrated Submm Wave Receiver: Development and Applications295
O Koryukin, A Pankratov, D Paveliev O Pylypenko, M Romanini, and S Shitov; as well as T de
Graauw and W Wild are acknowledged for their support of this work.
The work was supported in parts by RFBR projects 09–02–00246, 09–02–12172-ofi-m, Grant
for Leading Scientific School 5423.2010.2 and State contract No. 02.740.11.0795.
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