High-Gain Current Amplifiers for Low-Power MOSFET-C Filters
Department ofElectrical Engineering
Faculty ofEngineering, Prince of Songkhla University
Hat-Yai, Thailand e-mail:phanumas.kgpsu.ac.th
Mahanakorn Microelectronics Research Centre
Mahanakorn University ofTechnology
Bangkok, Thailand e-mail:apisakgmut.ac.th
alternatives for low-power MOSFET-C or active-RC filters.
High gain of the proposed current amplifiers relies on internal
high impedance nodes. Such high impedance is realised using
positive feedback cross-coupling configuration to cancel its
negative feedback counterpart. Two different output stages,
namely class-A and class-AB are also introduced. Bias circuits
suitable for the proposed amplifiers are also introduced where
they further help minimise power consumption and silicon
area in a high-order
employed in a simple second-order filter were compared with a
conventional folded-cascode OTA amplifier under 1-V supply.
Simulation results with 0.18gm CMOS process indicate correct
functionalities and superior linearity performance offered by
the proposed amplifiers.
MOSFET-C INTEGRATOR EMPLOYING CURRENT
employing an ideal current amplifier (CA) with current gain
ofAi as an active building block in Fig. 1, a simple analysis
filter design. The proposed circuits
Y + Y
whereY1=1/R, Y2=sCand YL= load admittance. IfAi>> Y2+
YL, an ideal integrator transfer function can be realised
accordingly, that is
The rapid growth of today's wireless communication
market has placed an ever-increasing demand on integrated
filter performance particularly in terms of linearity, dynamic
range and low power consumption under a low voltage
supply constraint. The MOSFET-C filter design technique
has portrayed itself as a simple and effective method for
integrating high complexity continuous-time
single CMOS chip with large dynamic range and accurate
frequency response -.
filters on a
A key building block in MOSFET-C filters (and also as in a
classical active-RC filter) is an operational amplifier (OA).
operational transconductance amplifier (OTA)
transconductance, can also be employed as an active building
block. Achieving a large transconductance OTA under a low
voltage supply might not be straightforward. In the past
decade, OTA have been extensively researched for various
circuit applications -. This work employs a very high-
gain current amplifier (CA) as an alternative to OA and OTA
in MOSFET-C filter (active-RC) while still maintains a
amplifiers have also been used in voltage amplifier design
(, , ) and recently in gmC-Opamp high frequency
filters () in order to achieve high-bandwidth and low-
Fig.1 MOSFET-C or active-RC integrators employing high-
it has been demonstrated in ,
 that an
Similar to OA-RC and OTA-RC cases, a linear integrator
can be realised from Fig. 1 as long as the current amplifier
possesses a high gain without extra linearization circuitry.
Fig.2 shows a possible architecture of a very high-gain
current amplifier. The high current gain of this current
amplifier comes from an intermediate high-impedance node
X and it is equals to Ai=Zx Gm.
less power than OA, with
Previously, high-gain current
Fig.2 High-gain current amplifier concept
0-7803-9390-2/06/$20.00 ©)2006 IEEE457
PROPOSED HIGH-GAIN CURRENT AMPLIFIERS FOR
MOSFET-C OR ACTIVE-RC FILTERS
A balanced MOSFET-C integrator employing a high-
gain current amplifier based upon the structure in Fig.2 is
depicted in Fig.3a and Fig.3b. A common-gate NMOS
structure M1-M2 is utilised as a current buffer allowing input
current to flow into high impedance nodes X, X'. At these
recognised by differential signals. In contrast, these same
nodes are seen by common-mode signals as low impedance,
hence DC bias voltage can be established without requiring
additional common-mode feedback (CMFB) amplifier as in
the case ofthe circuit in . This gain enhancement and DC
voltage setting-up technique thus consumes no extra current
An output stage of the amplifier utilises a
PMOS source-coupled pair with current source 21p where it
provides capability of common-mode rejection
amplifier. However, such structure resembles class-A output
configuration, therefore the output current swing is limited
by a current source (21p).
A more attractive current amplifier structure is shown in
Fig.3b where PMOS M3, M4 are connected in parallel to
MI1, M13 and M12, M14 respectively in order to provide
output current in a class-AB manner allowing large current
signal excursion because it is not limited by fixed current
source. This circuit thus should render better linearity than its
counterpart inFig.3a. However, this output structurerequires
M5-M6 and current mirrors M7-M1O with an appropriate
cross-coupling to provide (i) common-mode rejection (ii)
differential signal addition (combining drain signal currents
ofM5 with M4 and M6 with M3). Fig.4 illustrates how this
particular output stage handle signals differently. Such
circuit arrangement provides feed-forward common-mode
rejection, but in a different manner to that proposed in ,
because the diferential signals (and transconductances) of
M3, M6 and M4, M5 are added constructively. Therefore it
can effectively provide common-mode rejecting without
incurring any penalty in the overall transconductance/bias-
current efficiency, i.e., bias current consumed by M5-M8 is
not wastedjust for common-mode rejection.
is combined with the cross-coupling
(a) Class-A output
(b) Class-AB output
(c) Symbolic representations
Fig.3 Proposed high-gaincurrentamplifiers
Additional simple tuneable positive or negative resistance
networks shown in Fig.5 could be attached to node X, X' of
the circuits in Fig.3 so that the gain can be electronically
adjusted. Note also that the amplifier bias setting voltages
VCG and VNO are supplied from bias circuitries as described
inthenext section.O,, ,
(a) Comon-mode: rejection
IV.SECOND-ORDERFILTERAND BIASING TECHNIQUE
M7LlF lM 9
MO 1 IM8
applications in a high-order filter design, here a resonator as
shown in Fig.6 whose structure is widely known , . The
Fig.4 Signal handling ofthe output stage in Fig. 3b
Fig.5 Tuneable negative andpositive resistance network
resonator employs a bias network that sets all necessary DC
voltages for the whole filter without requiring any additional
CMFB amplifiers for each current amplifier.
Fig.6 MOSFET-Cfilter employing theproposed amplifiers.
Such bias set-up network (Fig.7) employs a feedback servo
loop to set VCG for every current amplifier so that the DC
voltage at amplifier's input terminals (sources of MNI and
MN2 in Fig.3) is equal to VREFi=VSET. In the filter structure
with cascade configuration, it is common to bias amplifiers'
input and output terminals at the same DC level to allow
optimal signal swing. Therefore, the DC output voltage of
each current amplifier should also be set at VSET.
* IN~~~2/0.6 M5
In most cases of MOSFET-C or active-RC filters (cascade
between amplifiers' output and input terminals via resistors
(or triode-MOS),and ifthese resistorscarryinfinitesimal DC
currents, the DC output terminal voltages of the amplifiers
will be automatically equal to those of input terminals. The
second servo loop in Fig.7a with VREFO=VSET is thus needed
to make sure that those interconnecting resistors carry no DC
currents, where it sets voltage VNO for every single current
amplifier in the filter to make sure that M3 and M5 (M4 and
M6) carry the same DC bias current. The bias network for
the amplifier in Fig.3b (Fig.7b) does not require a second
servo-loop because the output bias current ofM3 and M4 is
equal to M9 and M1O thanks to a common-mode rejection
network as described in the previous section. Therefore the
output bias voltage is set via resistor R and
To bias filter
-, there are always interconnections
(a) For Fig.3a
To bias filter
(b) For Fig.3b
Fig.7 Bias circuitriesforMOSFET-Cfilters in Fig. 6.
Therefore in a high-order filter design (such as in , ),
only one of this bias network is needed for the whole filters
and it thus helps minimise power consumption and silicon
area since no extra CMFB amplifiers are required for DC
voltage setting up. The proposed current amplifiers in Fig.3
can be readily modified into two-stage OTAs by injecting
input voltage signal at gates of Ml, M2 with their sources
joined together. However, each OTA necessitates high-
performance CMFB amplifier at its output to set up DC bias
voltage and maintain filter stability. Such OTAs are thus not
suitable for low-power filter design.
amplifier was employed and compared with a conventional
balanced folded-cascode OTA  (simulated using the same
process under the same supply of IV). Note that the negative
resistance network in Fig.5 is also attached at intermediate
node of such
enhancement and Q control.
currents between the proposed CAs and the OTA were
selected such that these amplifiers consume same bias
current and silicon area so that a
recognised. However, in the case of the OTA, CMFB
amplifiers were employed to set up DC voltage, i.e., such
OTA-RC filterrequiresmorepower consumption by design.
availablein the process, the proposed current
sizes and bias
V. SIMULATIONRESULTSWith IN-IP-2.OgA (implemented from a simple
The resonator of Fig.6 was simulated under 1-V supply
voltage with VSET=0.5V employing 0.18gm CMOS process.
Using low threshold-voltage MOS (IVTN
simple triode NMOS transistors), C1=C2=2pF, the filter ac
responses are illustrated in Fig.8. At Ql10, the total noise of
these three filters (lowpass output2 were found to be fairly Download full-text
similar and it was equal to
lHz to 100MHz). The frequency responses of the filter are
depicted in Fig.8.
same limited input range of21pR=2x2.0gAx50kQ=0±2V.
Moreover, without a negative resistance network connected
to the intermediate low impedance node of the folded-
cascode OTA, high-Q cannot be attained and linearity is
severely degraded because the transconductance of the OTA
is not sufficiently high.
1 x 10- v2 (integrated noise from
CONCLUSION AND FUTURE WORKS
transcon uctance amplifier in MOSFET-C and active-RC
filters. The second-order filters employing the proposed
current amplifiers render better linearity performance over
the conventional folded-cascode OTA. Future works include
utilisation of theproposed amplifiersinhigh-orderfilters
similar to those in ,  for practical applications.
X\ 4= ... 0 % 0 0 High-gam current amplifiers have been
I I I I
iM 10 fl
(b) Q tuning(a) Lowpass andbandpass responses
O: CA class-A
This work is financially supported by Thailand Research
Fund (TRF) under grant number MRG4780085.
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Fig.9 THD comparison atfn= 1OOkHz
0: CA class-A
E]:o CA class-AB
0: CA class-A
Distortionperformance comparison of thelowpass output
single-tone and two-tone linearity tests are demonstrated,
where THD = total harmonic distortion and IMD3 = third-
order intermodulation distortion. Fig. l0b indicates IMD3
comparison with input frequency sweeps from 100kHz to
-26dBVp for each tone). It can be seen that the proposed
current amplifiers render superior performance over the
covetina OTA. Imrvmn over
10dB (for IMD3) has been noticed. Note that due to the fixed
current source Ip, the lowpass filters implementedwith CA mn
Fig.3a and the conventional folded-cascode OTA have the
0) is shown in Fig.9, Fig.10 in which both
(all witeQad0)iS shown in Fig.9,Fige10inwhcth bote
G. Palmisano, G. Palumbo and S. Pennisi, "High-Drive CMOS
15d(fr TI-ID an