High-gain current amplifiers for low-power MOSFET-C filters
ABSTRACT Current amplifiers have been proposed as alternatives for low-power MOSFET-C or active-RC filters. High gain of the proposed current amplifiers relies on internal high impedance nodes. Such high impedance is realised using positive feedback cross-coupling configuration to cancel its negative feedback counterpart. Two different output stages, namely class-A and class-AB are also introduced. Bias circuits suitable for the proposed amplifiers are also introduced where they further help minimise power consumption and silicon area in a high-order filter design. The proposed circuits employed in a simple second-order filter were compared with a conventional folded-cascode OTA amplifier under 1-V supply. Simulation results with 0.18mum CMOS process indicate correct functionalities and superior linearity performance offered by the proposed amplifiers
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ABSTRACT: The state of the art of continuous-time filter design is reviewed. Several techniques are discussed and compared in terms of performance and implementation feasibility in different fabrication technologies. This review does not aim at historical completeness, but rather emphasizes techniques that have proven their worth in commercial applications. Brief mention is also made of experimental work which, in the opinion of the author, shows promise for the futureIEEE Journal of Solid-State Circuits 04/1994; · 3.06 Impact Factor
Conference Proceeding: Fully integrated active RC filters in MOS technology[show abstract] [hide abstract]
ABSTRACT: Fully integrated voiceband continous-time active RC filters in CMOS technology will be discussed. With ±5V supplies, 6V p-p signals can be handled in less than 1% THD and 96dB dynamic range.Solid-State Circuits Conference. Digest of Technical Papers. 1983 IEEE International; 03/1983 · 3.06 Impact Factor
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ABSTRACT: A voice-band continuous-time filter is described which was designed based on the technique of fully balanced networks and was fabrication in a 3.5-/spl mu/ CMOS technology. The filter implements a fifth-order elliptic low-pass transfer function with 0.05-dB passband ripple and 3.4 kHz cutoff frequency. A phase-locked loop control system fabricated on the same chip automatically references the frequency response of the filter to an external fixed clock frequency. The cutoff frequency was found to vary by less than 0.1% for an operating temperature range of 0-85/spl deg/C. The absolute value accuracy of the cutoff frequency was 0.5% (standard deviation). With /spl plusmn/5-V power supplies the measured dynamic range of the filter was approximately 100 dB.IEEE Journal of Solid-State Circuits 01/1986; · 3.06 Impact Factor
High-Gain Current Amplifiers for Low-Power MOSFET-C Filters
Department of Electrical Engineering
Faculty of Engineering, Prince of Songkhla University
Hat-Yai, Thailand e-mail: email@example.com
Mahanakorn Microelectronics Research Centre
Mahanakorn University of Technology
Bangkok, Thailand e-mail: firstname.lastname@example.org
MOSFET-C INTEGRATOR EMPLOYING CURRENT
Abstract— Current amplifiers have been proposed as
alternatives for low-power MOSFET-C or active-RC filters.
High gain of the proposed current amplifiers relies on internal
high impedance nodes. Such high impedance is realised using
positive feedback cross-coupling configuration to cancel its
negative feedback counterpart. Two different output stages,
namely class-A and class-AB are also introduced. Bias circuits
suitable for the proposed amplifiers are also introduced where
they further help minimise power consumption and silicon
area in a high-order filter design. The proposed circuits
employed in a simple second-order filter were compared with a
conventional folded-cascode OTA amplifier under 1-V supply.
Simulation results with 0.18µ µm CMOS process indicate correct
functionalities and superior linearity performance offered by
the proposed amplifiers.
The rapid growth of today’s wireless communication
market has placed an ever-increasing demand on integrated
filter performance particularly in terms of linearity, dynamic
range and low power consumption under a low voltage
supply constraint. The MOSFET-C filter design technique
has portrayed itself as a simple and effective method for
integrating high complexity continuous-time filters on a
single CMOS chip with large dynamic range and accurate
frequency response -.
A key building block in MOSFET-C filters (and also as in a
classical active-RC filter) is an operational amplifier (OA).
However, it has been demonstrated in ,  that an
operational transconductance amplifier (OTA) typically
consume less power than OA, with sufficiently large
transconductance, can also be employed as an active building
block. Achieving a large transconductance OTA under a low
voltage supply might not be straightforward. In the past
decade, OTA have been extensively researched for various
circuit applications -. This work employs a very high-
gain current amplifier (CA) as an alternative to OA and OTA
in MOSFET-C filter (active-RC) while still maintains a
required filter realization. Previously, high-gain current
amplifiers have also been used in voltage amplifier design
(, , ) and recently in gmC-Opamp high frequency
filters () in order to achieve high-bandwidth and low-
Consider a MOSFET-C or active-RC integrator
employing an ideal current amplifier (CA) with current gain
of Ai as an active building block in Fig.1, a simple analysis
yield voltage output/input relationship (in terms of
admittance Y and Ai)
where Y1=1/R, Y2=sC and YL = load admittance. If Ai >> Y2 +
YL, an ideal integrator transfer function can be realised
accordingly, that is
Fig.1 MOSFET-C or active-RC integrators employing high-
gain current amplifiers
Similar to OA-RC and OTA-RC cases, a linear integrator
can be realised from Fig.1 as long as the current amplifier
possesses a high gain without extra linearization circuitry.
Fig.2 shows a possible architecture of a very high-gain
current amplifier. The high current gain of this current
amplifier comes from an intermediate high-impedance node
X and it is equals to Ai = Zx⋅Gm.
Fig.2 High-gain current amplifier concept
ISCAS 20060-7803-9390-2/06/$20.00 ©2006 IEEE
III. PROPOSED HIGH-GAIN CURRENT AMPLIFIERS FOR
MOSFET-C OR ACTIVE-RC FILTERS
A balanced MOSFET-C integrator employing a high-
gain current amplifier based upon the structure in Fig.2 is
depicted in Fig.3a and Fig.3b. A common-gate NMOS
structure M1-M2 is utilised as a current buffer allowing input
current to flow into high impedance nodes X, X´. At these
nodes in Fig.3a, a diode-connected PMOS (negative
feedback, M7-M8) is combined with the cross-coupling
PMOS (positive feedback, M9-M10) to realise high
impedance. However, such high impedance is only
recognised by differential signals. In contrast, these same
nodes are seen by common-mode signals as low impedance,
hence DC bias voltage can be established without requiring
additional common-mode feedback (CMFB) amplifier as in
the case of the circuit in . This gain enhancement and DC
voltage setting-up technique thus consumes no extra current
consumption. An output stage of the amplifier utilises a
PMOS source-coupled pair with current source 2IP where it
provides capability of common-mode rejection for the
amplifier. However, such structure resembles class-A output
configuration, therefore the output current swing is limited
by a current source (2IP).
A more attractive current amplifier structure is shown in
Fig.3b where PMOS M3, M4 are connected in parallel to
M11, M13 and M12, M14 respectively in order to provide
output current in a class-AB manner allowing large current
signal excursion because it is not limited by fixed current
source. This circuit thus should render better linearity than its
counterpart in Fig.3a. However, this output structure requires
M5-M6 and current mirrors M7-M10 with an appropriate
cross-coupling to provide (i) common-mode rejection (ii)
differential signal addition (combining drain signal currents
of M5 with M4 and M6 with M3). Fig.4 illustrates how this
particular output stage handle signals differently. Such
circuit arrangement provides feed-forward common-mode
rejection, but in a different manner to that proposed in ,
because the differential signals (and transconductances) of
M3, M6 and M4, M5 are added constructively. Therefore it
can effectively provide common-mode rejecting without
incurring any penalty in the overall transconductance/bias-
current efficiency, i.e., bias current consumed by M5-M8 is
not wasted just for common-mode rejection.
Additional simple tuneable positive or negative resistance
networks shown in Fig.5 could be attached to node X, X´ of
the circuits in Fig.3 so that the gain can be electronically
adjusted. Note also that the amplifier bias setting voltages
VCG and VNO are supplied from bias circuitries as described
in the next section.
IV. SECOND-ORDER FILTER AND BIASING TECHNIQUE
Amplifiers of Fig.3 are employed to demonstrate
applications in a high-order filter design, here a resonator as
shown in Fig.6 whose structure is widely known , . The
(a) Class-A output
(b) Class-AB output
(c) Symbolic representations
Fig.3 Proposed high-gain current amplifiers
M7M9 M10 M8
(a) Common-mode: rejection
(b) Differential: addition
Fig.4 Signal handling of the output stage in Fig.3b
Fig.5 Tuneable negative and positive resistance network
resonator employs a bias network that sets all necessary DC
voltages for the whole filter without requiring any additional
CMFB amplifiers for each current amplifier.
Such bias set-up network (Fig.7) employs a feedback servo
loop to set VCG for every current amplifier so that the DC
voltage at amplifier’s input terminals (sources of MN1 and
MN2 in Fig.3) is equal to VREFi=VSET. In the filter structure
with cascade configuration, it is common to bias amplifiers’
input and output terminals at the same DC level to allow
optimal signal swing. Therefore, the DC output voltage of
each current amplifier should also be set at VSET.
In most cases of MOSFET-C or active-RC filters (cascade
configuration) -, there are always interconnections
between amplifiers’ output and input terminals via resistors
(or triode-MOS), and if these resistors carry infinitesimal DC
currents, the DC output terminal voltages of the amplifiers
will be automatically equal to those of input terminals. The
second servo loop in Fig.7a with VREFo=VSET is thus needed
to make sure that those interconnecting resistors carry no DC
currents, where it sets voltage VNO for every single current
amplifier in the filter to make sure that M3 and M5 (M4 and
M6) carry the same DC bias current. The bias network for
the amplifier in Fig.3b (Fig.7b) does not require a second
servo-loop because the output bias current of M3 and M4 is
equal to M9 and M10 thanks to a common-mode rejection
network as described in the previous section. Therefore the
output bias voltage is set via resistor R and it equals
Therefore in a high-order filter design (such as in , ),
only one of this bias network is needed for the whole filters
and it thus helps minimise power consumption and silicon
area since no extra CMFB amplifiers are required for DC
voltage setting up. The proposed current amplifiers in Fig.3
can be readily modified into two-stage OTAs by injecting
input voltage signal at gates of M1, M2 with their sources
joined together. However, each OTA necessitates high-
performance CMFB amplifier at its output to set up DC bias
voltage and maintain filter stability. Such OTAs are thus not
suitable for low-power filter design.
The resonator of Fig.6 was simulated under 1-V supply
voltage with VSET=0.5V employing 0.18µm CMOS process.
Using low threshold-voltage MOS (|VTN| = 0.25V, |VTP| =
Fig.6 MOSFET-C filter employing the proposed amplifiers.
To bias filter
To bias filter
(a) For Fig.3a
To bias filter
(b) For Fig.3b
Fig.7 Bias circuitries for MOSFET-C filters in Fig.6.
0.25V) available in the process, the proposed current
amplifier was employed and compared with a conventional
balanced folded-cascode OTA  (simulated using the same
process under the same supply of 1V). Note that the negative
resistance network in Fig.5 is also attached at intermediate
low-impedance node of such OTA to allow gain
enhancement and Q control. Transistor sizes and bias
currents between the proposed CAs and the OTA were
selected such that these amplifiers consume same bias
current and silicon area so that a fair comparison is
recognised. However, in the case of the OTA, CMFB
amplifiers were employed to set up DC voltage, i.e., such
OTA-RC filter requires more power consumption by design.
With IN=IP=2.0µA (implemented from a simple single-
transistor current source), R1=R2=R3=50kΩ (utilising
simple triode NMOS transistors), C1=C2=2pF, the filter ac
responses are illustrated in Fig.8. At Q~10, the total noise of
these three filters (lowpass output) were found to be fairly
similar and it was equal to ~1×10-6V2 (integrated noise from
1Hz to 100MHz). The frequency responses of the filter are
depicted in Fig.8.
(a) Lowpass and bandpass responses (b) Q tuning
Fig.8 Second-order filter ac response
: CA class-A
: CA class-AB
Fig.9 THD comparison at fin = 100kHz
: CA class-A
: CA class-AB
: CA class-A
: CA class-AB
(a) fin=100kHz+110kHz (b) IMD3 vs frequency
Fig.10 IMD3 comparison
Distortion performance comparison of the lowpass output
(all with Q~10) is shown in Fig.9, Fig.10 in which both
single-tone and two-tone linearity tests are demonstrated,
where THD = total harmonic distortion and IMD3 = third-
order intermodulation distortion. Fig.10b indicates IMD3
comparison with input frequency sweeps from 100kHz to
1500kHz (for roughly constant output magnitude of
-26dBVp for each tone). It can be seen that the proposed
current amplifiers render superior performance over the
conventional OTA. Improvement over 15dB (for THD) and
10dB (for IMD3) has been noticed. Note that due to the fixed
current source IP, the lowpass filters implemented with CA in
Fig.3a and the conventional folded-cascode OTA have the
same limited input range of 2IP·R = 2×2.0µA×50kΩ=±0.2V.
Moreover, without a negative resistance network connected
to the intermediate low impedance node of the folded-
cascode OTA, high-Q cannot be attained and linearity is
severely degraded because the transconductance of the OTA
is not sufficiently high.
VI. CONCLUSION AND FUTURE WORKS
High-gain current amplifiers have been successfully
employed as an alternative active building block to
transconductance amplifier in MOSFET-C and active-RC
filters. The second-order filters employing the proposed
current amplifiers render better linearity performance over
the conventional folded-cascode OTA. Future works include
utilisation of the proposed amplifiers in high-order filters
similar to those in ,  for practical applications.
amplifiers or operational
This work is financially supported by Thailand Research
Fund (TRF) under grant number MRG4780085.
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