A tunable EBG absorber for radio-frequency power imaging
ABSTRACT Absorption characteristics of a tunable electromagnetic band-gap (EBG) absorber are analyzed, which is designed to capture 2d radio-frequency (RF) power distributions incident on the absorber surface. The EBG absorber has lumped resistors interconnecting the mushroom-type surface patches to absorb the incident RF power at the resonance frequency where the EBG structure exhibits a high-impedance feature. The absorbed RF power distribution is measured by directly detecting the amounts of RF power consumed by the individual resistors. Varactor diodes are inserted in parallel with the resistors for tuning the resonance frequency of narrowband absorption. The absorption characteristics at normal incidence are evaluated in detail based on an equivalent circuit model which exactly explains the frequency behavior of the surface impedance of the tunable EBG absorber observed in EM simulation. The small resistance existing in the varactor diode makes it difficult for the surface impedance to be matched with the incident wave impedance (i.e., for a high absorption to be achieved) over a wide range of resonance frequency. A means to improve the absorption performance of the tunable EBG absorber is examined.
- SourceAvailable from: Agostino Monorchio[show abstract] [hide abstract]
ABSTRACT: High-impedance surfaces (HIS) comprising lossy frequency selective surfaces (FSS) are employed to design thin electromagnetic absorbers. The structure, despite its typical resonant behavior, is able to perform a very wideband absorption in a reduced thickness. Losses in the frequency selective surface are introduced by printing the periodic pattern through resistive inks and hence avoiding the typical soldering of a large number of lumped resistors. The effect of the surface resistance of the FSS and dielectric substrate characteristics on the input impedance of the absorber is discussed by means of a circuital model. It is shown that the optimum value of surface resistance is affected both by substrate parameters (thickness and permittivity) and by FSS element shape. The equivalent circuit model is then used to introduce the working principles of the narrowband and the wideband absorbing structure and to derive the best-suited element for wideband absorption.IEEE Transactions on Antennas and Propagation 06/2010; · 2.33 Impact Factor
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ABSTRACT: A navel ultra-thin radar-absorbing material (RAM) using metanwterials is presented and the absorption performance is examined. Due to the high-impedance property of the metamalerials. the thickness of the RAM is about several tenths of the center wavelength of the absorption band, which is considerably thinner than conventional absorbers. The absorption bandwidth of the RAM is about several hundred megahertz01/2005;
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ABSTRACT: By covering a metal ground plane with a periodic surface texture, we can alter its electromagnetic properties. The impedance of this metasurface can be modeled as a parallel resonant circuit, with sheet inductance L, and sheet capacitance C. The reflection phase varies with frequency from +π to -π, and crosses through 0 at the LC resonance frequency, where the surface behaves as an artificial magnetic conductor. By incorporating varactor diodes into the texture, we have built a tunable impedance surface, in which an applied bias voltage controls the resonance frequency, and the reflection phase. We can program the surface to create a tunable phase gradient, which can electronically steer a reflected beam over +/- 40° in two dimensions, for both polarizations. We have also found that this type of resonant surface texture can provide greater bandwidth than conventional reflectarray structures. This new electronically steerable reflector offers a low-cost alternative to a conventional phased array.IEEE Transactions on Antennas and Propagation 11/2003; · 2.33 Impact Factor
978-1-4244-5118-0/11/$26.00 ©2011 IEEE
A Tunable EBG Absorber for Radio-Frequency Power Imaging
Satoshi Yagitani1, Keigo Katsuda1, Ryo Tanaka1, Masayuki Nojima1,
Yoshiyuki Yoshimura2, and Hirokazu Sugiura2
1Graduate School of Natural Science and Technology, Kanazawa University,
Kakuma-machi, Kanazawa 920-1192, Japan
2Industrial Research Institute of Ishikawa, 2-1 Kuratsuki, Kanazawa 920-8203, Japan
designed to capture 2d radio-frequency (RF) power distributions incident on the absorber surface. The EBG absorber
has lumped resistors interconnecting the mushroom-type surface patches to absorb the incident RF power at the
resonance frequency where the EBG structure exhibits a high-impedance feature. The absorbed RF power distribution
is measured by directly detecting the amounts of RF power consumed by the individual resistors. Varactor diodes are
inserted in parallel with the resistors for tuning the resonance frequency of narrowband absorption. The absorption
characteristics at normal incidence are evaluated in detail based on an equivalent circuit model which exactly explains
the frequency behavior of the surface impedance of the tunable EBG absorber observed in EM simulation. The small
resistance existing in the varactor diode makes it difficult for the surface impedance to be matched with the incident
wave impedance (i.e., for a high absorption to be achieved) over a wide range of resonance frequency. A means to
improve the absorption performance of the tunable EBG absorber is examined.
A variety of thin electromagnetic absorbers have been designed based on the artificial high-impedance surfaces
such as frequency-selective surfaces and metamaterial surfaces (e.g.,  and references therein). At the resonance
frequency where these surfaces exhibit the high-impedance feature, an incident wave is absorbed by the additional
resistive components which are matched with the incident wave impedance. Recently it was proposed that a thin
absorber can be used for monitoring 2d radio-frequency (RF) power distributions incident on the absorber surface .
A mushroom-type electromagnetic band-gap (EBG) structure is used as the high-impedance surface, where the
absorption is achieved by “lumped resistors” connecting between the adjacent patches on the mushroom layer . With
this configuration the power of an RF wave incident on the mushroom surface is absorbed (or consumed) by the lumped
resistors. By directly measuring the power consumption in each of the lumped resistors arranged in a 2-d matrix, the 2-
d distribution of the RF power incident and absorbed on the mushroom surface is obtained. Such an “RF power
imager” has inherently a narrowband feature around the resonance frequency fixed by the geometrical and
constitutional structure of the EBG surface. To extend the measurable frequency range, the resonance frequency is
made electronically tunable by additional varactor diodes (varactors), as in [4-5]. A 347-mm square EBG absorber was
designed and fabricated to cover the absorbing frequency range from 700 MHz up to 2.7 GHz. Power distributions
were detected at 8 × 8 locations on the absorber, at each of which two RF power detectors with the sensitivity of -70
dBm were placed to measure two orthogonal polarizations. The measured power distributions were transferred to a PC
and displayed as real-time 2-d power images at a rate of 30 images/second. The RF power distributions radiated from a
dipole antenna were measured to be consistent with those expected theoretically, which validated the proposed
technique to measure the RF power distributions. Using such an RF power imager, the power distributions of even
impulsive RF signals and/or noises can be captured and visualized in situ and in real-time, while the electromagnetic
environment is almost undisturbed by the EBG absorber.
In the present study, the absorption characteristics of the tunable EBG absorber designed for RF power
imaging is evaluated in detail based on equivalent circuit analysis and EM simulation.
2. A Tunable EBG Absorber for Detecting RF Power Distribution
Figure 1 shows the geometrical structure of a square unit cell of the tunable EBG absorber designed for RF
Absorption characteristics of a tunable electromagnetic band-gap (EBG) absorber are analyzed, which is
power imaging, which has lumped resistors and varactors inserted between the adjacent patches on the surface . The
gap g between the adjacent patches is much smaller than the patch size w, and the periodicity of the unit cells, a = w + g,
is set much smaller than the wavelength. For an electromagnetic wave at normal incidence, the surface impedance of
the mushroom structure itself is represented as a parallel connection of the effective inductance and capacitance. The
high-impedance feature is achieved at the LC resonance frequency, where the mushroom layer behaves like an artificial
magnetic conductor. The incident wave power is absorbed by the lumped resistors interconnecting the surface patches;
if we take the value of the resistors as R = 377 Ω matched with the incident wave impedance, the incident wave should
be completely absorbed at the resonance frequency . On the other hand, tunability is achieved by the varactors
inserted in parallel with the resistors, by altering the capacitance component of the EBG structure which specifies the
resonance frequency. As in the same manner designed by , the varactors are oriented in opposite directions in each
alternate row as well as in each alternate column of the matrix of mushrooms. Reverse biases are supplied to all the
varactors by alternately biasing half of the cells, and grounding the other half in a checkerboard pattern (see Fig. 3 of
). A separate biasing circuit is placed on the backside of the ground plane. Thus, by applying appropriate bias
voltages to the varactors, we can control the frequency of RF power absorption. The locations of the varactor and the
resistor, as well as their separation distance d, on each side of a patch has an effect on the surface impedance, as
discussed in Sec. 3.
η0= 377 Ω
η0= 377 Ω
Fig. 1: Structure of a unit cell of the EBG absorber Fig. 2: Equivalent circuit model
On the EBG absorber in Fig. 1, the incident wave power is absorbed and dissipated in the surface resistors,
when there are no losses in the varactors and in the substrate. The amount of power absorbed by each resistor depends
on the incident polarization; the resistors connecting the adjacent patches in the x- and y-directions absorb the amounts
of RF power with the electric field polarized in the x- and y-directions, respectively . In either case, the power
absorbed by each resistor is considered to be the Poynting flux of the incident wave multiplied by the area of a unit cell.
By detecting directly the amounts of power consumed in the 2-d matrix of surface resistors, the 2-d power distribution
of the RF wave illuminating the EBG surface is measured with polarization discrimination. Power detectors are put on
the backside of the EBG absorber, to detect the amounts of power consumed by the individual surface resistors .
3. Equivalent Circuit Analysis
Here the characteristics of the tunable EBG absorber shown in Fig. 1 are evaluated. The geometrical and
constitutional parameters are similar to those of the EBG absorber designed in , which had 33 × 33 square unit cells
formed on an FR-4 substrate of 347 mm square and 1.6 mm thick. The size of a patch is w = 10 mm and the gap
between the adjacent patches is g = 0.5 mm, so that the cell periodicity is a = 10.5 mm. The via diameter is 0.6 mm.
The relative permittivity of the FR-4 substrate is taken as 4.56 with no loss (tanδ = 0). The varactor is modeled as a
series RLC circuit; the series resistance RD = 1 Ω, the parasitic inductance LD = 1.8 nH and the capacitance CD is
variable from 0.67 to 12 pF (which makes the abosorber tunable from 700 MHz to 2.7 GHz). It is noted that the resistor
was chosen here as R = 845 Ω instead of 377 Ω, to have maximum absorption at 2 GHz (see the discussion in the next
paragraph). Using these parameters, the absorption characteristics of the EBG absorber were computed by an EM
simulator (CST Microwave Studio). A linearly polarized plane wave was incident normally on the EBG surface. A
square area containing four unit cells of the absorber was modeled by defining the periodic boundary condition, which
corresponds to simulating infinitely extending periodic unit cells.
shown in Fig.2 (a). The effective capacitance and inductance of the mushroom structure are CS = 0.628 pF and LS =
1.86 nH, respectively. The effects of the surface currents on a patch flowing toward the varactor (CD , LD and RD) and
the resistor (R) are represented by a transformer, LV and LR coupled with the coefficient k, which are dependent of the
geometrical configuration of the varactor and the resistor on each side of the patch (i.e., their locations as well as their
separation distance d). The parameters of the transformer can be replaced with LV’, LR’ and their mutual inductance M
as in Fig. 2 (b); for the case of d = 1 mm, LV’ = 0.371 nH, LR’ = 0.41 nH and M = 0.235 nH. From this circuit, two
resonance frequencies are derived by solving Eq. (1) for ω .
The surface impedance Z of the EBG absorber becomes purely resistive RP at each resonance frequency, and the
reflection coefficient is given as Γ = (RP −η0) / (RP +η0), where η0 = 377 Ω is the incident wave impedance. When the
capacitance of the varactor CD is varied, the resonance frequencies are changed accordingly. The value of the resistance
RP at the resonance frequency ω r is calculated as
where // means the parallel connection of the impedance. The typical frequency variation of the surface impedance
under the circuit parameters mentioned above is shown in Fig. 3. The absolute value of surface impedance |Z| is shown
by the solid line for the case of CD = 1.35 pF. In this case the first and second resonance frequencies appear at 2.0 GHz
and 6.7 GHz, respectively, which are observed as the two peaks on |Z|. When CD is varied from 12 pF down to 0.67 pF,
the first resonance frequency is changed from 700 MHz up to 2.7 GHz, whereas the second one is from 6.23 GHz up to
6.85 GHz. Over each resonance frequency range, the value of the resistance moves on the broken curve specified by Eq.
(2). From the viewpoint of achieving frequency tunability, the first resonance should be taken, as the variable range of
the second resonance frequency is unpractically narrow. To achieve a high absorption at each resonance frequency, the
resistance should be as close as possible to the incident wave impedance, η0 = 377 Ω (the dotted line). From Eq. (2), RP
becomes equal to R at ω r = ω S , and RP ~ (ZS
becomes smaller as the resonance frequency becomes lower. Thus the existence of the small resistance RD in the
varactor has a considerable effect on the frequency behavior of the surface impedance . Here ω S /2π = 4.66 GHz and
becomes larger and smaller than 377 Ω, respectively, leading to impedance mismatch in either case.
The equivalent circuit which exactly reproduces the absorption characteristics obtained in the simulation is
L CLL M CL C
/RD) (ω r /ω S)2 // [R / (1+M /LS)2] for ω r << ω S ; the surface resistance
/RD = 2.96 kΩ, and R was chosen as 845 Ω so that RP becomes 377 Ω at 2.0 GHz. Above and below 2.0 GHz, RP
Impedance |Z| [Ω]
(CD= 1.35 pF)
CD= 5.81 pF
Fig. 3: Surface impedance of the EBG absorber Fig. 4: Reflection magnitude of the EBG absorber
analysis and the simulation, which are plotted by gray and dotted lines, respectively. In each case four representative
profiles are plotted for the varactor capacitances fixed as CD = 5.81 pF, 2.51 pF, 1.35 pF and 0.81 pF, which correspond
to the resonance frequencies of 1.0 GHz, 1.5 GHz, 2.0 GHz and 2.5 GHz, respectively. For each of the capacitance
values, the reflection profile calculated by the equivalent circuit analysis agrees well to that observed in the simulation,
Figure 4 shows the reflection magnitude profiles for the first resonance obtained by the equivalent circuit
which validates the accuracy of the equivalent circuit in Fig. 2. Also for the second resonance, though not shown here,
a good agreement was observed between the reflection profiles obtained by the equivalent circuit analysis and in the
simulation. On each curve, the reflection becomes minimum (the absorption becomes maximum) at the resonance
frequency determined by the value of CD. As discussed above, the amount of absorption becomes highest (S11 ~ −45
dB) at the resonance frequency of 2.0 GHz for the case of CD = 1.35 pF, where the surface impedance of the EBG
absorber RS becomes 377 Ω. When the resonance frequency goes above or below 2.0 GHz, the maximum absorption is
degraded due to the increase in mismatch between RS and 377 Ω.
As discussed above, Eq. (2) is approximated as RP ~ (ZS
frequencies ω r << ω S . If the three conditions, (ZS
Ω so that a high absorption would be achieved over a certain range of resonance frequency. Since the first condition is
rewritten as ω r
The other is to increase the effective inductance LS by increasing the thickness of the substrate. The gray line in Fig. 3
shows the surface resistance profile expected for the case of RD = 0.5 Ω, R = 377 Ω, LS = 3.72 nH (corresponding to the
substrate thickness of 3.2 mm) and M = 0 nH (which is achievable by adjusting the distance d between the resistor and
the varactor). In this case RP ~ 377 Ω is achieved over a wide range of resonance frequency above 1 GHz.
Absorption characteristics of the tunable EBG absorber used for imaging the RF power distributions were
evaluated by the equivalent circuit analysis and the simulation. A realistic equivalent circuit model was derived which
exactly explains the frequency behavior of the surface impedance of the EBG absorber for normal incidence observed in
the simulation. It was shown that the small resistance existing in the varactor diode makes it difficult for the surface
impedance to be matched with the incident wave impedance over a wide range of resonance frequency. A means to
improve the absorption performance of the tunable EBG absorber was examined. Evaluation for oblique incidence will
be the future work.
The authors would like to thank Dr. M. Ozaki for his valuable suggestions and discussions. This study was
supported by KAKENHI (21560444).
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/RD) (ω r /ω S)2 // [R / (1+M /LS)2] at first resonance
/RD) (ω r /ω S)2 >> R, R = 377 Ω and M << LS , are satisfied, RP ~ 377
2 / RD >> R, one way to make it so is to use the varactor with the resistance RD as small as possible.